Whitaker, Jerry C. "

Frontmatter"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

ELECTRONICS HANDBOOK SERIES

Series Editor:

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Jerry C. Whitaker

POWER

VACUUM

TUBES

S e c o n d E d i t i o n

Boca Raton London New York Washington, D.C.

CRC Press

Jerry C. Whitaker

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Whitaker, Jerry C.

Power vacuum tubes handbook / Jerry C. Whitaker

p. cm. -- (Electronics handbook series)

Includes bibliographical references and index.
ISBN 0-8493-1345-7 (alk. paper)
1. Vacuum tubes.
2. Power electronics. I. Title. II. Series.

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99-21062

CIP

Preface

The phrase "high technology" is perhaps one of the more overused descriptions in our
technical vocabulary. It is a phrase generally reserved for discussion of integrated cir-
cuits, fiber optics, satellite systems, and computers. Few people would associate high
technology with vacuum tubes. The notion that vacuum tube construction is more art
than science may have been true 10 or 20 years ago, but today it's a different story.

The demand on the part of industry for tubes capable of higher operating power and

frequency, and the economic necessity for tubes that provide greater efficiency and re-
liability, have moved power tube manufacturers into the high-tech arena. Advance-
ments in tube design and construction have given end users new transmitters and RF
generators that allow industry to grow and prosper.

If you bring up the subject of vacuum tubes to someone who has never worked on a

transmitter or high-power RF generator, you are likely to get a blank stare and a ques-
tion: "Do they make those anymore?" Although receiving tubes have more-or-less dis-
appeared from the scene, power tubes are alive and well and are performing vital func-
tions in thousands of divergent applications. Solid-state and tube technologies each
have their place, each with its strengths and weaknesses. It should be noted that even re-
ceiving tubes are staging somewhat of a comeback in high-end audio applications.

Tube design and development, although accompanied by less fanfare, is advancing

as are developments in solid-state technology. Power tubes today are designed with an
eye toward high operating efficiency and high gain/bandwidth properties. Above all, a
tube must be reliable and provide long operating life. The design of a new power tube is
a lengthy process that involves computer-aided calculations and advanced modeling.

Despite the inroads made by solid-state technology, the power vacuum tube occu-

pies--and will continue to occupy--an important role in the generation of high-power
radio frequency energy in the high-frequency regions and above. No other device can
do the job as well. Certainly, solid-state cannot, especially if cost, size, and weight are
important considerations.

The field of science encompassed by power vacuum tubes is broad and exciting. It is

an area of growing importance to military and industrial customers, and a discipline in
which significant research is now being conducted.

Power vacuum tubes include a wide range of devices, each for specific applications.

Devices include power grid tubes (triodes, tetrodes, and pentodes) and microwave
power tubes (klystrons, traveling wave tubes, gyrotrons, and numerous other high-fre-
quency devices). Research is being conducted for both tube classes to extend output
power and maximum frequency, and to improve operating efficiency.

This book examines the underlying technology of each type of power vacuum tube

device in common use today and provides examples of typical applications. New devel-
opment efforts also are reported, and the benefits of the work explained.

This Second Edition of Power Vacuum Tubes is directed toward engineering person-

nel involved in the design, specification, installation, and maintenance of high-power
equipment utilizing vacuum tubes. Basic principles are discussed, with emphasis on
how the underlying technology dictates the applications to which each device is dedi-

cated. Supporting mathematics are included where appropriate to explain the material
being discussed. Extensive use of technical illustrations and schematic diagrams aid
the reader in understanding the fundamental principles of the subject.

Today's modern power tube is unlike the power tubes in use a decade ago. And with

the trend in industry toward operation at higher power levels and higher frequencies,
the vacuum tube is certain to remain on the scene for a long time to come.

Jerry C. Whitaker

For updated information on this and other engineering books, visit the web site

www.technicalpress.com

About the Author

Jerry Whitaker is a technical writer based in Morgan Hill, California, where he oper-
ates the consulting firm Technical Press. Mr. Whitaker has been involved in various
aspects of the communications industry for more than 20 years. He is a Fellow of the
Society of Broadcast Engineers and an SBE-certified Professional Broadcast Engi-
neer. He is also a member and Fellow of the Society of Motion Picture and Television
Engineers, and a member of the Institute of Electrical and Electronics Engineers. Mr.
Whitaker has written and lectured extensively on the topic of electronic systems in-
stallation and maintenance.

Mr. Whitaker is the former editorial director and associate publisher of Broadcast

Engineering and Video Systems magazines. He is also a former radio station chief engi-
neer and TV news producer.

Mr. Whitaker is the author of a number of books, including:

·

AC Power Systems, 2nd Edition, CRC Press, 1998.

·

DTV: The Revolution in Electronic Imaging, McGraw-Hill, 1998.

·

Editor-in-Chief, NAB Engineering Handbook, National Association of Broad-
casters, 1998.

·

Editor-in-Chief, The Electronics Handbook, CRC Press, 1996.

·

Coauthor, Communications Receivers: Principles and Design, McGraw-Hill,
1996.

·

Electronic Displays: Technology, Design, and Applications, McGraw-Hill,
1994.

·

Coauthor, Interconnecting Electronic Systems, CRC Press, 1992.

·

Coeditor, Television Engineering Handbook, revised edition, McGraw-Hill,
1992.

·

Coeditor, Information Age Dictionary, Intertec/Bellcore, 1992.

·

Maintaining Electronic Systems, CRC Press, 1991.

·

Radio Frequency Transmission Systems: Design and Operation, McGraw-Hill,
1990.

·

Coauthor, Television and Audio Handbook for Technicians and Engineers,
McGraw-Hill, 1990.

Mr. Whitaker has twice received a Jesse H. Neal Award Certificate of Merit from the
Association of Business Publishers for editorial excellence. He also has been recog-
nized as Educator of the Year by the Society of Broadcast Engineers.

Acknowledgement

The author wishes to express appreciation to Varian Associates, whose support was
invaluable in the preparation of this book.

Contents

Preface

About the Author

Chapter 1: Power Vacuum Tube Applications

1.1 Introduction

1.1.1 Vacuum Tube Development

1.1.2 Standardization

1.1.3 Transmission Systems

1.2 Vacuum Tube Applications

1.2.1 Market Overview

1.2.2 AM Radio Broadcasting

1.2.3 Shortwave Broadcasting

1.2.4 FM Radio Broadcasting

1.2.5 TV Broadcasting

1.2.6 Satellite Transmission

1.2.7 Radar

1.2.8 Electronic Navigation

1.2.9 Microwave Radio

1.2.10 Induction Heating

1.2.11 Electromagnetic Radiation Spectrum

1.3 Bibliography

Chapter 2: Modulation Systems and Characteristics

2.1 Introduction

2.1.1 Modulation Systems

2.1.2 Principles of Resonance

2.1.3 Frequency Source

2.1.4 Operating Class

2.1.5 Broadband Amplifier Design

2.1.6 Thermal and Circuit Noise

2.2 Amplitude Modulation

2.2.1 High-Level AM Modulation

2.2.2 Vestigial-Sideband Amplitude Modulation

2.2.3 Single-Sideband Amplitude Modulation

2.2.4 Quadrature Amplitude Modulation (QAM)

2.3 Frequency Modulation

2.3.1 Modulation Index

2.3.2 Phase Modulation

2.3.3 Modifying FM Waves

2.3.4 Preemphasis and Deemphasis

2.3.5 Modulation Circuits

2.4 Pulse Modulation

2.4.1 Digital Modulation Systems

2.4.2 Pulse Amplitude Modulation

2.4.3 Pulse Time Modulation (PTM)

2.4.4 Pulse Code Modulation

2.4.5 Delta Modulation

2.4.6 Digital Coding Systems

2.4.7 Baseband Digital Pulse Modulation

2.4.8 Spread Spectrum Systems

2.5 References

2.6 Bibliography

Chapter 3: Vacuum Tube Principles

3.1 Introduction

3.2 Characteristics of Electrons

3.2.1 Electron Optics

3.2.2 Thermal Emission From Metals

3.2.3 Secondary Emission

3.2.4 Diode

3.2.5 Triode

3.2.6 Tetrode

3.2.7 Pentode

3.2.8 High-Frequency Operating Limits

3.3 Vacuum Tube Design

3.3.1 Device Cooling

3.3.2 Cathode Assembly

3.3.3 Grid Structures

3.3.4 Plate Assembly

3.3.5 Ceramic Elements

3.3.6 Tube Construction

3.3.7 Connection Points

3.3.8 Tube Sockets

3.4 Neutralization

3.4.1 Circuit Analysis

3.4.2 Circuit Design

3.4.3 Grounded-Grid Amplifier Neutralization

3.4.4 Self-Neutralizing Frequency

3.4.5 Neutralization Adjustment

3.5 References

3.6 Bibliography

Chapter 4: Designing Vacuum Tube Circuits

4.1 Introduction

4.1.1 Class A Amplifier

4.1.2 Class B and AB Amplifiers

4.1.3 Class C Amplifier

4.2 Principles of RF Power Amplification

4.2.1 Drive Power Requirements

4.2.2 Mechanical and Electrical Considerations

4.2.3 Bypassing Tube Elements

4.2.4 Parasitic Oscillations

4.2.5 Shielding

4.2.6 Protection Measures

4.3 Cavity Amplifier Systems

4.3.1 Bandwidth and Efficiency

4.3.2 Current Paths

4.3.3 The 1/4-Wavelength Cavity

4.3.4 The 1/2-Wavelength Cavity

4.3.5 Folded 1/2-Wavelength Cavity

4.3.6 Wideband Cavity

4.3.7 Output Coupling

4.3.8 Mechanical Design

4.4 High-Voltage Power Supplies

4.4.1 Silicon Rectifiers

4.4.2 Operating Rectifiers in Series

4.4.3 Operating Rectifiers in Parallel

4.4.4 Silicon Avalanche Rectifiers

4.4.5 Thyristor Servo Systems

4.4.6 Polyphase Rectifier Circuits

4.4.7 Power Supply Filter Circuits

4.5 Parameter Sampling Circuits

4.6 References

4.7 Bibliography

Chapter 5: Applying Vacuum Tube Devices

5.1 Introduction

5.2 AM Power Amplification Systems

5.2.1 Control Grid Modulation

5.2.2 Suppressor Grid Modulation

5.2.3 Cathode Modulation

5.2.4 High-Level AM Amplification

5.2.5 Pulse Width Modulation

5.3 Linear Amplification

5.3.1 Device Selection

5.3.2 Grid-Driven Linear Amplifier

5.3.3 Cathode-Driven Linear Amplifier

5.3.4 Intermodulation Distortion

5.4 High-Efficiency Linear Amplification

5.4.1 Chireix Outphasing Modulated Amplifier

5.4.2 Doherty Amplifier

5.4.3 Screen-Modulated Doherty-Type Amplifier

5.4.4 Terman-Woodyard Modulated Amplifier

5.4.5 Dome Modulated Amplifier

5.5 Television Power Amplifier Systems

5.5.1 System Considerations

5.5.2 Power Amplifier

5.6 FM Power Amplifier Systems

5.6.1 Cathode-Driven Triode Amplifier

5.6.2 Grounded-Grid vs. Grid-Driven Tetrode

5.6.3 Grid-Driven Tetrode/Pentode Amplifiers

5.6.4 Impedance Matching into the Grid

5.6.5 Neutralization

5.7 Special-Application Amplifiers

5.7.1 Distributed Amplification

5.7.2 Radar

5.8 References

5.9 Bibliography

Chapter 6: Microwave Power Tubes

6.1 Introduction

6.1.1 Linear-Beam Tubes

6.1.2 Crossed-Field Tubes

6.2 Grid Vacuum Tubes

6.2.1 Planar Triode

6.2.2 High-Power UHF Tetrode

6.2.3 Diacrode

6.3 Klystron

6.3.1 Reflex Klystron

6.3.2 The Two-Cavity Klystron

6.3.3 The Multicavity Klystron

6.3.4 Beam Pulsing

6.3.5 Integral vs. External Cavity

6.3.6 MSDC Klystron

6.4 Klystrode/Inductive Output Tube (IOT)

6.4.1 Theory of Operation

6.4.2 Electron Gun

6.4.3 Grid Structure

6.4.4 Input Cavity

6.4.5 Output Cavity

6.4.6 Application Considerations

6.4.7 Continuing Research Efforts

6.5 Constant Efficiency Amplifier

6.5.1 Theory of Operation

6.6 Traveling Wave Tube

6.6.1 Theory of Operation

6.6.2 Operating Efficiency

6.6.3 Operational Considerations

6.7 Crossed-Field Tubes

6.7.1 Magnetron

6.7.2 Backward Wave Oscillator

6.7.3 Strap-Fed Devices

6.7.4 Gyrotron

6.8 Other Microwave Devices

6.8.1 Quasiquantum Devices

6.8.2 Variations on the Klystron

6.9 Microwave Tube Life

6.9.1 Life-Support System

6.9.2 Protection Measures

6.9.3 Filament Voltage Control

6.9.4 Cooling System

6.9.5 Reliability Statistics

6.10 References

6.11 Bibliography

Chapter 7: RF Interconnection and Switching

7.1 Introduction

7.1.1 Skin Effect

7.2 Coaxial Transmission Line

7.2.1 Electrical Parameters

7.2.2 Electrical Considerations

7.2.3 Coaxial Cable Ratings

7.2.4 Mechanical Parameters

7.3 Waveguide

7.3.1 Propagation Modes

7.3.2 Ridged Waveguide

7.3.3 Circular Waveguide

7.3.4 Doubly Truncated Waveguide

7.3.5 Impedance Matching

7.3.6 Installation Considerations

7.3.7 Cavity Resonators

7.4 RF Combiner and Diplexer Systems

7.4.1 Passive Filters

7.4.2 Four-Port Hybrid Combiner

7.4.3 Non-Constant-Impedance Diplexer

7.4.4 Constant-Impedance Diplexer

7.4.5 Microwave Combiners

7.4.6 Hot Switching Combiners

7.4.7 Phased-Array Antenna Systems

7.5 High-Power Isolators

7.5.1 Theory of Operation

7.5.2 Applications

7.6 References

7.7 Bibliography

Chapter 8: Cooling Considerations

8.1 Introduction

8.1.1 Thermal Properties

8.1.2 Heal Transfer Mechanisms

8.1.3 The Physics of Boiling Water

8.2 Application of Cooling Principles

8.2.1 Forced-Air Cooling Systems

8.2.2 Water Cooling

8.2.3 Vapor-Phase Cooling

8.2.4 Temperature Measurements

8.2.5 Air-Handling System

8.3 Operating Environment

8.3.1 Air-Handling System

8.3.2 Air Cooling System Design

8.3.3 Site Design Guidelines

8.3.4 Water/Vapor Cooling System Maintenance

8.4 References

8.5 Bibliography

Chapter 9: Reliability Considerations

9.1 Introduction

9.1.1 Terminology

9.2 Quality Assurance

9.2.1 Inspection Process

9.2.2 Reliability Evaluation

9.2.3 Failure Analysis

9.2.4 Standardization

9.3 Reliability Analysis

9.3.1 Statistical Reliability

9.3.2 Environmental Stress Screening

9.3.3 Latent Defects

9.3.4 Operating Environment

9.3.5 Failure Modes

9.3.6 Maintenance Considerations

9.4 Vacuum Tube Reliability

9.4.1 Thermal Cycling

9.4.2 Tube-Changing Procedure

9.4.3 Power Tube Conditioning

9.4.4 Filament Voltage
9.4.5 Filament Voltage Management

9.4.6 PA Stage Tuning

9.4.7 Fault Protection

9.4.8 Vacuum Tube Life

9.4.9 Examining Tube Performance

9.4.10 Shipping and Handling Vacuum Tubes

9.5 Klystron Reliability

9.5.1 Cleaning and Flushing the Cooling System

9.5.2 Cleaning Ceramic Elements

9.5.3 Reconditioning Klystron Gun Elements

9.5.4 Focusing Electromagnet Maintenance

9.5.5 Power Control Considerations

9.6 References

9.7 Bibliography

Chapter 10: Device Performance Criteria

10.1 Introduction

10.2 Measurement Parameters

10.2.1 Power Measurements

10.2.2 Decibel Measurement

10.2.3 Noise Measurement

10.2.4 Phase Measurement

10.2.5 Nonlinear Distortion

10.3 Vacuum Tube Operating Parameters

10.3.1 Stage Tuning

10.3.2 Amplifier Balance

10.3.3 Parallel Tube Amplifiers

10.3.4 Harmonic Energy

10.3.5 Klystron Tuning Considerations

10.3.6 Intermodulation Distortion

10.3.7 VSWR

10.4 RF System Performance

10.4.1 Key System Measurements

10.4.2 Synchronous AM in FM Systems

10.4.3 Incidental Phase Modulation

10.4.4 CarrierAmplitude Regulation

10.4.5 Site-Related Intermodulation Products

10.5 References

10.6 Bibliography

Chapter 11: Safe Handling of Vacuum Tube Devices

11.1 Introduction

11.2 Electric Shock

11.2.1 Effects on the Human Body

11.2.2 Circuit Protection Hardware

11.2.3 Working with High Voltage

11.2.4 First Aid Procedures

11.3 Operating Hazards

11.3.1 OSHA Safety Considerations

11.3.2 Beryllium Oxide Ceramics

11.3.3 Corrosive and Poisonous Compounds

11.3.4 FC-75 Toxic Vapor

11.3.5 Nonionizing Radiation

11.3.6 X-Ray Radiation Hazard

11.3.7 Implosion Hazard

11.3.8 Hot Coolant and Surfaces

11.3.9 Polychlorinated Biphenyls

11.4 References

11.5 Bibliography

Chapter 12: Reference Data

Chapter 13: Glossary

Index of Figures

Index of Tables

Cited References

This book is dedicated to my daughter

Alexis Ann Whitaker

My greatest joy is watching you grow up

Whitaker, Jerry C. "Chapter

1 Power Vacuum Tube Applications"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

1

Power Vacuum Tube Applications

1.1

Introduction

The continuing demand for energy control devices capable of higher operating power,
higher maximum frequency, greater efficiency, and extended reliability have pushed
tube manufacturers to break established performance barriers. Advancements in tube
design and construction have given engineers new RF generating systems that allow
industry to grow and prosper. Power grid vacuum tubes have been the mainstay of
transmitters and other RF generation systems since the beginning of radio. Today, the
need for new gridded and microwave power tubes is being met with new processes
and materials.

Although low-power vacuum tubes have been largely replaced by solid-state de-

vices, vacuum tubes continue to perform valuable service at high-power levels and, par-
ticularly, at high frequencies. The high-power capability of a vacuum device results
from the ability of electron/vacuum systems to support high-power densities. Values
run typically at several kilowatts per square centimeter, but may exceed 10 MW/cm

2

.

No known dielectric material can equal these values. For the foreseeable future, if high
power is required, electron/vacuum devices will remain the best solution.

It is worthwhile to point out that certain devices within the realm of receiving tubes

still continue to find application within high-end audio systems. A select group of tubes
never went out of style because of their intrinsic benefits, at least as perceived by ele-
ments of the audiophile community. These "golden devices" include the 12AT7,
12AU7, 12AX7, 6L6, 6V6, and even the 5U4. These components are used today not
just in classic 1960s-era audio amplifiers, but also in new microphone preamps and
power amplifiers manufactured for sale to discriminating customers.

1.1.1 Vacuum Tube Development

Receiving tubes have more or less disappeared from the scene (the foregoing notwith-
standing) because of the development of transistors and integrated circuits. Power
grid and microwave tubes, however, continue to push the limits of technology. Power
tubes are an important part of RF technology today.

© 1999 CRC Press LLC

From 1887--when Heinrich Hertz first sent and received radio waves--to the pres-

ent, an amazing amount of progress has been made by engineers and scientists. The
public takes for granted today what was considered science fiction just a decade or two
ago. The route from the primitive spark-gap transmitters to the present state of the art
has been charted by the pioneering efforts of many. It is appropriate to review some of
the milestones in electron tube development. Much of the fundamental work on power
vacuum devices can be traced to early radio broadcasting, which--along with tele-
phone technology--has brought the nations of the world closer together than the early
pioneers of the art could have imagined. More than 80 years have passed since Charles
D. (Doc) Herrold founded a voice station (as it then was known) at San Jose, CA. Devel-
opments since have been the result of many inspired breakthroughs and years of plain
hard work.

Pioneer Developers

In 1895, 21-year-old Guglielmo Marconi and his brother Alfonso first transmitted ra-
dio signals across the hills behind their home in Bologna, Italy. Born in 1874 to an
Italian merchant and a Scotch-Irish mother, young Marconi had learned of Hertzian
waves from August Righi, a professor at the University of Bologna. Convinced that
such waves could be used for wireless communication, Marconi conducted prelimi-
nary experiments using a spark-gap source and a coherer detector. Unable to interest
the Italian government in his invention, Marconi took his crude transmitter and re-
ceiver to England, where he demonstrated his wireless system to officials of the Brit-
ish Post Office. Marconi received a patent for the device in July 1897. With the finan-
cial support of his mother's relatives, Marconi organized the Wireless Telegraph and
Signal Company that same year to develop the system commercially. Regular transat-
lantic communications commenced in 1903 when a Marconi station at Cape Cod,
MA, sent a short message from President Theodore Roosevelt to King Edward VII in
England.

The invention of the vacuum tube diode by J. Ambrose Fleming in 1904 and the tri-

ode vacuum tube amplifier by Lee De Forest in 1906 launched the electronics industry
as we know it. The De Forest invention was pivotal. It marked the transition of the vac-
uum tube from a passive to an active device. The new "control" electrode took the form
of a perforated metal plate of the same size and shape as the existing anode, positioned
between the filament and the anode. Encouraged by the early test results, De Forest
worked to perfect his invention, trying various mechanical arrangements for the new
grid.

With the invention of the control grid, De Forest had set in motion a chain of events

that led the vacuum tube to become the key element in the emerging discipline of elec-
tronics. Early experimenters and radio stations took this new technology and began de-
veloping their own tubes using in-house capabilities, including glassblowing. As the
young electronics industry began to grow, vacuum tubes were produced in great quan-
tity and standardized (to a point), making it possible to share new developments and ap-
plications. A major impetus for standardization was the U.S. military, which required
vacuum tubes in great quantities during World War I. Pushed by the navy, a standard-

© 1999 CRC Press LLC

ized design, including base pins and operating parameters, was forged. The economic
benefits to both the tube producers and tube consumers were quickly realized.

Most radio stations from 1910 through 1920 built their own gear. For example, at the

University of Wisconsin, Madison, special transmitting tubes were built by hand as
needed to keep radio station 9XM, which later became WHA, on the air. The tubes were
designed, constructed, and tested by Professor E. M. Terry and a group of his students in
the university laboratories. Some of the tubes also were used in wireless telephonic ex-
periments carried on with the Great Lakes Naval Training Station during 1918, when a
wartime ban was imposed on wireless broadcasts.

It took many hours to make each tube. The air was extracted by means of a mercury

vapor vacuum pump while the filaments were lighted and the plate voltage was on. As
the vacuum increased, the plate current was raised until the plate became red-hot. This
out-gassing process was primitive, but it worked. The students frequently worked
through the night to get a tube ready for the next day's broadcast. When completed, the
device might last only a few hours before burning out.

Plate dissipation on Professor Terry's early tubes, designated #1, #2, and so on, was

about 25 W. Tube #5 had a power output of about 50 W. Tubes #6 to #8 were capable of
approximately 75 W. Tube #8 was one of the earliest handmade commercial products.

The addition of a "screen" grid marked the next major advancement for the vacuum

tube. First patented in 1916 by Dr. Walter Schottky of the Siemens & Halske Company
(Germany), the device garnered only limited interest until after World War I when the
Dutch firm, Philips, produced a commercial product. The "double-grid" Philips type Q
was introduced in May 1923. Later variations on the initial design were made by Philips
and other manufacturers.

Early in the use of the tetrode it was determined that the tube was unsuited for use as

an audio frequency power amplifier. Under certain operating conditions encountered in
this class of service, the tetrode exhibited a negative-resistance characteristic caused by
secondary emission from the anode being attracted to the positively charged screen. Al-
though this peculiarity did not affect the performance of the tetrode as an RF amplifier,
it did prevent use of the device for AF power amplification. The pentode tube, utilizing
a third ("suppressor") grid was designed to overcome this problem. The suppressor grid
provided a means to prevent the secondary emissions from reaching the screen grid.
This allowed the full capabilities of the tube to be realized.

Philips researchers Drs. G. Holst and B. Tellegen are credited for the invention, re-

ceiving a patent for the new device in 1926.

Radio Central

The first major project that the young Radio Corporation of America tackled was the
construction of a huge radio transmitting station at Rocky Point, NY. The facility,
completed in 1921, was hailed by President Harding as a milestone in wireless prog-
ress. The president, in fact, put the station into operation by throwing a switch that had
been rigged up at the White House. Wireless stations around the globe had been
alerted to tune in for a congratulatory statement by the president.

© 1999 CRC Press LLC

For a decade, this station--known as Radio Central--was the only means of direct

communications with Europe. It was also the "hopping off" point for messages trans-
mitted by RCA to Central and South America.

The Rocky Point site was famous not only for its role in communications, but also

for the pioneers of the radio age who regularly visited there. The guest book lists such
pioneers as Guglielmo Marconi, Lee De Forest, Charles Steinmetz, Nikola Tesla, and
David Sarnoff. Radio Central was a milestone in transatlantic communications.

Originally, two antenna structures stood at the Rocky Point site, each with six 410 ft

towers. The towers stretched over a 3-mile area on the eastern end of Long Island.

The facility long outlived its usefulness. RCA demolished a group of six towers in

the 1950s; five more were destroyed in early 1960. The last tower of the once mighty
Radio Central was taken down on December 13, 1977.

WLW: The Nation's Station

Radio station WLW has a history as colorful and varied as any in the United States. It
is unique in that it was the only station ever granted authority to broadcast with 500
kW. This accomplishment pushed further the limits of vacuum tube technology.

The station actually began with 20 W of power as a hobby of Powell Crosley, Jr. The

first license for WLW was granted by the Department of Commerce in 1922. Crosley
was authorized to broadcast on a wavelength of 360 meters with a power of 50 W, three
evenings a week. Growth of the station was continuous. WLW operated at various fre-
quencies and power levels until, in 1927, it was assigned to 700 kHz at 50 kW and re-
mained there. Operation at 50 kW began on October 4, 1928. The transmitter was lo-
cated in Mason, OH. The station could be heard as far away as Jacksonville, FL, and
Washington, DC.

The superpower era of WLW began in 1934. The contract for construction of the

enormous transmitter was awarded to RCA in February 1933. Tests on the unit began
on January 15, 1934. The cost of the transmitter and associated equipment was approxi-
mately $400,000--not much today, but a staggering sum in the middle of the Great De-
pression.

At 9:02 p.m. on May 2, 1934, programming was commenced with 500 kW of power.

The superpower operation was designed to be experimental, but Crosley managed to
renew the license every 6 months until 1939. The call sign W8XO occasionally was
used during test periods, but the regular call sign of WLW was used for programming.

"Immense" is the only way to describe the WLW facility. The antenna (including the

flagpole at the top) reached a height of 831 ft. The antenna rested on a single ceramic in-
sulator that supported the combined force of 135 tons of steel and 400 tons exerted by
the guys. The tower was guyed with eight 1-7/8-in cables anchored 375 ft from the base
of the antenna.

The main antenna was augmented by a directional tower designed to protect CFRB,

Toronto, when the station was using 500 kW at night. The directional system was
unique in that it was the first designed to achieve both horizontal directivity and verti-
cal-angle suppression.

© 1999 CRC Press LLC

A spray pond in front of the building provided cooling for the system, moving 512

gallons of water per minute. Through a heat exchanger, the water then cooled 200 gal-
lons of distilled water in a closed system that cooled the transmitting tubes.

The transmitter consumed an entire building. Modulation transformers, weighing

37,000 lb each, were installed in the basement. Three plate transformers, a rectifier fil-
ter reactor, and a modulation reactor were installed outside the building. The exciter for
the transmitter produced 50 kW of RF power. A motor-generator was used to provide
125 V dc for control circuits.

The station had its own power substation. While operating at 500 kW, the transmitter

consumed 15,450,000 kWh per year. The facility was equipped with a complete ma-
chine shop because station personnel had to build much of the ancillary hardware
needed. Equipment included gas, arc and spot welders, a metal lathe, milling machine,
engraving machine, sander, drill press, metal brake, and a table saw. A wide variety of
electric components were also on hand.

WLW operated at 500 kW until March 1, 1939, when the FCC ordered the station to

reduce power to 50 kW. The station returned to superpower operation a few times dur-
ing World War II for government research. The days when WLW could boast of being
"the nation's station," however, were in the past.

UHF: A New Technical Challenge

The early planners of the U.S. television system thought that 13 channels would more
than suffice. The original channel 1 was from 44 MHz to 50 MHz, but because of pos-
sible interference with other services, it was dropped before any active use. There re-
mained 12 channels for normal broadcasting. Bowing to pressure from various
groups, the FCC revised its allocation table in 1952 to permit UHF-TV broadcasting
for the first time. The new band was not, however, a bed of roses. Many people went
bankrupt, building UHF stations only to find few receivers were available to the pub-
lic. UHF converters soon became popular. The first converters were so-called match-
box types that were good for one channel only. More expensive models mounted on
top of the TV receiver and were tunable. Finally, the commission issued an edict that
all TV set manufacturers had to include UHF tuning in their receivers. This move
opened the doors for significant market penetration for UHF broadcasters.

The klystron has been the primary means of generating high-power UHF-TV sig-

nals since the introduction of UHF broadcasting.

Birth of the Klystron

Quietly developed in 1937, the klystron truly revolutionized the modern world. In-
deed, the klystron may have helped save the world as we know it. More than 50 years
after it was first operated in a Stanford University laboratory by Russell Varian and
his brother Sigurd, the klystron and its offspring remain irreplaceable, even in the age
of solid-state microelectronics.

The Varian brothers were unusually bright and extremely active. Mechanically

minded, they produced one invention after another. Generally, Sigurd would think up an

© 1999 CRC Press LLC

idea, Russell would devise a method for making it work, then Sigurd would build the
device.

Through the influence of William Hansen, a former roommate of Russell and a

physics professor at Stanford University, the Varians managed to get nonpaying jobs as
research associates in the Stanford physics lab. They had the right to consult with mem-
bers of the faculty and were given the use of a small room in the physics building.

Hansen's role, apparently, was to shoot down ideas as fast as the Varians could

dream them up. As the story goes, the Varians came up with 36 inventions of varying
impracticality. Then they came up with idea number 37. This time Hansen's eyes wid-
ened. On June 5, 1937, Russell proposed the concept that eventually became the kly-
stron tube. The device was supposed to amplify microwave signals. With $100 for sup-
plies granted by Stanford, Sigurd built it.

The device was simple: A filament heated by an electric current in turn heated a

cathode. A special coating on the cathode gave off electrons when it reached a suffi-
ciently high temperature. Negatively charged electrons attracted by a positively
charged anode passed through the first cavity of the klystron tube. Microwaves in the
cavity interacted with the electrons and passed through a narrow passage called a drift
tube. In the drift tube, the electrons tended to bunch up; some speeded up, some slowed
down. At the place in the drift tube where the bunching was most pronounced, the elec-
trons entered a second cavity, where the stronger microwaves were excited and ampli-
fied in the process.

The first klystron device was lit up on the evening of August 19, 1937. Performance

was marginal, but confirmed the theory of the device. An improved klystron was com-
pleted and tested on August 30. The name for the klystron came from the Greek verb
klyzo, which refers to the breaking of a wave--a process much like the overtaking of
slow electrons by fast ones.

The Varians published the results of their discovery in the Journal of Applied Phys-

ics. For reasons that have never been clear, their announcement immediately impressed
British scientists working in the same field, but was almost entirely ignored by the Ger-
mans. The development of the klystron allowed British and American researchers to
build smaller, more reliable radar systems. Klystron development paralleled work be-
ing done in England on the magnetron.

The successful deployment of microwave radar was accomplished by the invention

of the cavity magnetron at Manchester University in the late 1930s [10]. It was one of
Britain's "Top Secrets" handed over to the Americans early in the war. The cavity mag-
netron was delivered to the Radiation Laboratory at MIT, where it was incorporated
into later wartime radar systems. During the Battle of Britain in May 1940, British de-
fenses depended upon longer wavelength radar (approximately 5 m), which worked but
with insufficient resolution. The magnetron provided high-power microwave energy at
10 cm wavelengths, which improved detection resolution enormously.

Armed with the magnetron and klystron, British and American scientists perfected

radar, a key element in winning the Battle of Britain. So valuable were the secret de-
vices that the British decided not to put radar in planes that flew over occupied Europe
lest one of them crash, and the details of the components be discovered.

© 1999 CRC Press LLC

After the war, the Varians--convinced of the potential for commercial value of the

klystron and other devices they had conceived--established their own company. For
Stanford University, the klystron represents one of its best investments: $100 in seed
money and use of a small laboratory room were turned into $2.56 million in licensing
fees before the patents expired in the 1970s, three major campus buildings and hun-
dreds of thousands of dollars in research funding.

At about the same time the Varian brothers were working to perfect the klystron, An-

drew Haeff of RCA Laboratories was developing what would come to be known as the
inductive output tube (IOT). The crucial feature in both devices was that the electron
beam, having given up a large part of its energy in a small, low-capacitance gap, tended
to spread because of space-charge effects (the mutual repulsion of electrons). The char-
acteristics of electron flow make it possible to use multiple collectors with differing po-
tentials in tubes with large energy spread in the spent electron beam, such as the kly-
stron and IOT. This multistage depressed-collector (MSDC) technique results in a con-
siderable increase in operating efficiency. One of the earliest patents on the MSDC was
issued in 1943 to Charles Litton, who later formed a company that would become
Litton Industries.

Figure 1.1

shows the basic MSDC structure, developed from the orig-

inal Litton patent.

Figure 1.1 Simplified drawing of a multistage depressed-collector klystron using six
collector elements.

© 1999 CRC Press LLC

Nuclear Magnetic Resonance (NMR)

NMR, a technique that has revolutionized chemistry throughout the world, had its ori-
gins in experiments carried out on opposite sides of the North American continent in
the late 1940s. As is often the case with basic discoveries in science, the original in-
tent of the work was far removed from the ultimate practical application. NMR was
first demonstrated during the winter of 1945 by professors Felix Bloch and William
Hansen and their associates at Stanford University. At the same time another group
was working independently at Harvard University, directed by Dr. E. M. Purcell. The
Nobel Prize for physics was awarded to Bloch and Purcell in 1952 in recognition of
their pioneering work in NMR, which allows chemists to make structural determina-
tions of substances, and thereby see how a molecule is put together.

The Transistor Is Born

In December 1947, Dr. William Shockley of Bell Laboratories changed the course of
history by demonstrating to his colleagues a newly discovered device that exhibited
what he called the transistor effect. From this demonstration, and a later one at the
Bell Labs in New York City on June 30, 1948, sprang one of the most important in-
ventions of the 20th century--the working transistor. (The phrase "transistor" came
about from a contraction of "transfer resistor.") For their development efforts, Bell
Telephone scientists John Bardeen, Walter H. Brattain, and William Shockley re-
ceived the Nobel Prize for physics in 1956.

The first experimental transistor made its debut as a 3/16-in-diameter, 5/8-in-long

metal cylinder. Today, transistors are fabricated in dimensions finer than the wave-
length of light. The first integrated circuit, made more than 30 years ago, had two tran-
sistors and measured 7/16-in-long and 1/16-in-thick. Today, a VHSIC (very high speed
integrated circuit) die with more than 10 million transistors is in production for use in
desktop computer systems.

The transistor was an economical and durable alternative to receiving vacuum tubes.

The tubes were big, fragile, and had a relatively short operating life. They consumed
lots of power and, as a result, got very hot. Transistors offered a way to make a product
that was compact, efficient, and reliable.

What made transistors distinctive was that they were fabricated from a single solid

material that either insulated or conducted, depending on the purity of the base mate-
rial. A solitary transistor could be scribed on nothing more than a chip of germanium
(subsequently supplanted by silicon). Eager to exploit the riches of solid-state technol-
ogy, engineers drew up complicated circuits, cramming boards of discrete devices into
tight little islands.

Satellite Technology

The phrase "live via satellite" is commonplace today. Communications satellites have
meant not only live coverage of world events, but also more service to more people at
lower cost. The potential for use of satellites for communications was first demon-
strated in 1960 with the launching by the United States of Echo 1 and Echo 2. These

© 1999 CRC Press LLC

passive reflector satellites bounced radio signals across the Atlantic. This type of sat-
ellite, however, left a lot to be desired for communications purposes. Echo was super-
seded by Telstar 1, an active repeater satellite, launched 2 years later. Telstar demon-
strated that color video signals could be reliably broadcast across the oceans and, in
doing so, captured the interest and imagination of the public.

The first live transatlantic telecast was relayed by Telstar on July 10, 1962. The pic-

ture was of the American flag fluttering in front of the sending station at Andover, ME.
More panoramic telecasts, showing life in widely distant places, were exchanged be-
tween the United States and Europe 13 days later.

With the potential of international communications becoming increasingly appar-

ent, Congress passed the Communications Satellite Act in 1962. The legislation cre-
ated, among other things, Comsat, which was to establish, in cooperation with organi-
zations in other countries, a global commercial communications satellite system as
quickly as possible.

U.S. initiative under the Communications Satellite Act, combined with growing in-

ternational interest in the new technology, led to the formation of Intelsat in 1964.
Acting as a technical manager of Intelsat during its initial growth period, Comsat devel-
oped Intelsat's first geosynchronous commercial communications satellite, Early Bird.
The project brought to reality the concept envisaged some 20 years earlier by Arthur C.
Clarke, the noted British science fiction writer.

Early Bird (also known as Intelsat 1) was launched from Cape Kennedy on April 2,

1965 and placed into synchronous orbit 22,300 miles above the coast of Brazil. The
launch marked the first step toward a worldwide network of satellites linking the peo-
ples of many nations. Early Bird, the only mode of live transatlantic television, provided
in July 1965 the first live telecast (via Intelsat satellite) to the United States, an Ameri-
can-vs.-Soviet track meet. All of these breakthroughs relied on new, compact, efficient,
and lightweight microwave power tubes.

Although a dramatic improvement over the transatlantic telecommunications facili-

ties at the time, Early Bird was nonetheless limited in capacity and capability. For ex-
ample, in order for the only TV channel to be operative, all 240 voice channels had to be
shut down. Furthermore, the cost of Early Bird time was high.

Following Early Bird's introduction to service in the Atlantic Ocean region, the

challenge remained to develop a global network. The next step toward that goal was
taken on July 11, 1967, with the successful launch of Intelsat 2, which established satel-
lite communications between the U.S. mainland and Hawaii. Two years later Intelsat III
was launched for Indian Ocean region service, thereby completing the provisions of
global coverage.

Fortunately, global coverage capacity was in place just in time for the TV audi-

ence--estimated to be the largest in world history--to see man set foot on the moon.
The satellite system that had been a vision by Arthur C. Clarke two decades earlier, and
a formidable legislative mandate more than a decade earlier, had emerged as a reality.

© 1999 CRC Press LLC

1.1.2 Standardization

Throughout the history of product development, design standardization has been crit-
ically important. To most engineers, the term "standards" connotes a means of pro-
moting an atmosphere of interchangeability of basic hardware. To others, it evokes
thoughts of a slowdown of progress, of maintaining a status quo--perhaps for the
benefit of a particular group. Both camps can cite examples to support their view-
points, but no one can seriously contend that we would be better off without stan-
dards. The standardization process has been an important element in the advancement
of power vacuum tube technology.

In 1836 the U.S. Congress authorized establishment of the Office of Weights and

Measures (OWM) for the primary purpose of ensuring uniformity in customs house
dealings. The Treasury Department was charged with its operation. As advancements
in science and technology fueled the industrial revolution, it was apparent that stan-
dardization of hardware and test methods was necessary to promote commercial devel-
opment and to compete successfully in the world market. The industrial revolution in
the 1830s introduced the need for interchangeable parts and hardware. Wide use of
steam railways and the cotton gin, for example, were possible only with mechanical
standardization.

By the late 1800s, professional organizations of mechanical, electrical, and chemi-

cal engineers were founded with this aim in mind. The American Institute of Electrical
Engineers developed standards based on the practices of the major electrical manufac-
turers between 1890 and 1910. Because such activities were not within the purview of
the OWM, there was no government involvement during this period. It took the pres-
sures of war production in 1918 to cause the formation of the American Engineering
Standards Committee (AESC) to coordinate the activities of various industry and engi-
neering societies. This group became the American Standards Association (ASA) in
1928.

Parallel development occurred worldwide. The International Bureau of Weights and

Measures was founded in 1875, the International Electrotechnical Commission (IEC)
in 1904, and the International Federation of Standardizing Bodies in 1926. Following
World War II, this group was reorganized as the International Standards Organization
(ISO). Today, representatives of approximately 54 countries serve on the ISO's 145
technical committees.

The International Telecommunications Union (ITU) was founded in 1865 for the

purpose of coordinating and interfacing telegraphic communications worldwide. To-
day, its 164 member countries develop regulations and voluntary recommendations re-
lating to telecommunications systems.

Many of the early standards relating to vacuum tubes in the United States were de-

veloped by equipment manufacturers, first under the banner of the Radio Manufac-
turers Association (RMA), then the Radio, Electronic, and Television Manufacturers
Association (RETMA), and now the Electronic Industries Association (EIA). The In-
stitute of Radio Engineers (the forerunner of the IEEE) was responsible for measure-
ment standards and techniques.

© 1999 CRC Press LLC

Standards usually are changed only through natural obsolescence. Changes in basic

quantities, such as units of length and volume, are extremely difficult for the general
public to accept. In 1900 nearly all members of the scientific, commercial, and engi-
neering communities supported the change to the metric system. But the general public
finds the idea of changing ingrained yardsticks for weight and measure as unpalatable
as learning to speak a new language in a native country. The effort to convert to the met-
ric system is, in fact, still under way, with various degrees of success.

1.1.3 Transmission Systems

Radio communication is the grandfather of vacuum tube development. Amplitude
modulation (AM) was the first modulation system that permitted voice communica-
tions to take place. This simple scheme was predominant throughout the 1920s and
1930s. Frequency modulation (FM) came into regular broadcast service during the
1940s. TV broadcasting, which uses amplitude modulation for the visual portion of
the signal and frequency modulation for the aural portion of the signal, became avail-
able to the public during the mid-1940s. These two basic approaches to modulating a
carrier have served the communications industry well for many decades. Although
the basic schemes still are used today, numerous enhancements have been made.

Technology also has changed the rules by which the communications systems devel-

oped. AM radio, as a technical system, offered limited audio fidelity but provided de-
sign engineers with a system that allowed uncomplicated transmitters and simple, inex-
pensive receivers. FM radio, on the other hand, offered excellent audio fidelity but re-
quired a complex and unstable transmitter (in the early days) and complex, expensive
receivers. It is no wonder that AM radio flourished while FM remained relatively stag-
nant for at least 20 years after being introduced to the public. It was not until transistors
and later integrated circuits became commonly available that FM receivers gained con-
sumer acceptance.

TV broadcasting evolved slowly during the late 1940s and early 1950s. Color trans-

missions were authorized as early as 1952. Color receivers were not purchased in large
numbers by consumers, however, until the mid-1960s. Early color sets were notorious
for poor reliability and unstable performance. They also were expensive. As with FM,
all that changed with the introduction of transistors and, later, integrated circuits. Most
color TV receivers produced today consist of an integrated circuit (IC) chip set number-
ing 8 to 15 devices.

1.2

Vacuum Tube Applications

The range of uses for power grid and microwave tubes is wide and varied. Some of the
more common applications include the following:

·

AM radio broadcasting, with power levels up to 50 kW

·

Shortwave radio, with power levels of 50 kW to several megawatts

·

FM radio broadcasting, with power levels up to 100 kW

© 1999 CRC Press LLC

·

Television, with operating power levels up to 5 MW

·

Radar, with widely varying applications ranging from airborne systems to
over-the-horizon networks

·

Satellite communications, incorporating the ground segment and space segment

·

Industrial heating and other commercial processes

Figure 1.2

charts the major device types in relation to power and frequency.

1.2.1 Market Overview

There is a huge installed base of power vacuum tubes. Users cover a wide range of
disciplines and facilities. The power tube market can be divided into the following
general segments:

·

Medical. Applications include diagnostics, such as X ray and CAT scanners, and
treatment, such as radio therapy. Devices used include X ray tubes and klystrons.
This important area of power tube application has seen continuous growth over
the past two decades.

·

Scientific. Applications include various types of research, including particle ac-
celerators for nuclear research. A wide variety of devices are used in this area.
Specialized units can be found in "big science" projects, such as the huge Stan-
ford linear accelerator. The worldwide scientific market for microwave power
tubes continues to grow as researchers reach for higher power levels and higher
frequencies of operation.

·

Electronic Warfare. Applications include ship- and air-based radar systems,
eavesdropping systems, and radio-signal jamming equipment. Electronic war-
fare applications are a driving force in power microwave tube development. For
example, the Aegis weapons system ship uses hundreds of microwave power
tubes.

·

Communications. Applications include space- and ground-based satellite sys-
tems, microwave relay, and specialized point-to-point communications systems.
Power tubes play an important role in all types of high-power communications
systems.

·

Radar. Applications include ground-, ship-, and air-based radar for both com-
mercial and military applications.

·

Industrial Heating. Applications center on material heating and chemical pro-
cessing, and environmental uses such as smokestack scrubbing.

·

Broadcasting. Applications are TV and radio broadcasting, including shortwave.
High-efficiency UHF-TV is an area of rapid development.

Despite the inroads made by solid-state technology, power vacuum tube applications
continue to span a broad range of disciplines around the world.

© 1999 CRC Press LLC

Figure 1.2 Vacuum tube operating parameters based on power and frequency.

©

1
999
CRC Press LLC

1.2.2 AM Radio Broadcasting

AM radio stations operate on 10 kHz channels spaced evenly from 540 to 1600 kHz.
Various classes of stations have been established by the Federal Communications
Commission (FCC) and agencies in other countries to allocate the available spectrum
to given regions and communities. In the United States, the basic classes are clear, re-
gional, and local. Current practice uses the CCIR (international) designations as class
A, B, and C, respectively. Operating power levels range from 50 kW for a clear chan-
nel station to as little as 250 W for a local station. AM stations choosing to do so may
operate in stereo using the C-QUAM system. (C-QUAM is a registered trademark of
Motorola.) To receive a stereo AM broadcast, consumers must purchase a new stereo
radio. The C-QUAM system transmits the stereo sum signal (in others words, the
monophonic signal) in the usual manner and places the stereo difference signal on a
phase-modulated subchannel. Decoder circuits in the receiver reconstruct the stereo
signals.

1.2.3 Shortwave Broadcasting

The technologies used in commercial and government-sponsored shortwave broad-
casting are closely allied with those used in AM radio. However, shortwave stations
usually operate at significantly higher powers than AM stations.

International broadcast stations use frequencies ranging from 5.95 to 26.1 MHz.

The transmissions are intended for reception by the general public in foreign countries.

Figure 1.3

shows the frequencies assigned by the FCC for international broadcast

shortwave service. The minimum output power is 50 kW. Assignments are made for
specific hours of operation at specific frequencies.

Figure 1.3 Operating frequency bands for shortwave broadcasting.

© 1999 CRC Press LLC

High-power shortwave transmitters have been installed to serve large geographical

areas and to overcome jamming efforts by foreign governments. Systems rated for
power outputs of 500 kW and more are common. RF circuits and vacuum tube devices
designed specifically for high-power operation are utilized.

Most shortwave transmitters have the unique requirement for automatic tuning to

one of several preset operating frequencies. A variety of schemes exist to accomplish
this task, including multiple exciters (each set to the desired operating frequency) and
motor-controlled variable inductors and capacitors. Tune-up at each frequency is per-
formed by the transmitter manufacturer. The settings of all tuning controls are stored in
a memory device or as a set of trim potentiometer adjustments. Automatic retuning of a
high-power shortwave transmitter can be accomplished in less than 30 s in most cases.

1.2.4 FM Radio Broadcasting

FM radio stations operate on 200 kHz channels spaced evenly from 88.1 to 107.9
MHz. In the United States, channels below 92.1 MHz are reserved for noncommer-
cial, educational stations. The FCC has established three classifications for FM sta-
tions operating east of the Mississippi River and four classifications for stations west
of the Mississippi. Power levels range from a high of 100 kW effective radiated power
(ERP) to 3 kW or less for a lower classification. The ERP of a station is a function of
transmitter power output (TPO) and antenna gain. ERP is determined by multiplying
these two quantities together and allowing for line loss.

A transmitting antenna is said to have "gain" if, by design, it concentrates useful en-

ergy at low radiation angles, rather than allowing a substantial amount of energy to be
radiated above the horizon (and be lost in space). FM and TV transmitting antennas are
designed to provide gain through vertically stacking individual radiating elements.

Stereo broadcasting is used almost universally in FM radio today. Introduced in the

mid-1960s, stereo has contributed in large part to the success of FM radio. The left and
right sum (monophonic) information is transmitted as a standard frequency-modulated
signal. Filters restrict this main channel signal to a maximum bandwidth of approxi-
mately 17 kHz. A pilot signal is transmitted at low amplitude at 19 kHz to enable decod-
ing at the receiver. The left and right difference signal is transmitted as an ampli-
tude-modulated subcarrier that frequency-modulates the main FM carrier. The center
frequency of the subcarrier is 38 kHz. Decoder circuits in the FM receiver matrix the
sum and difference signals to reproduce the left and right audio channels.

Figure 1.4

il-

lustrates the baseband signal of a stereo FM station.

Auxiliary Services

Modern FM broadcast stations are capable of broadcasting not only stereo program-
ming, but one or more subsidiary channels as well. These signals, referred to by the
FCC as subsidiary communications authorization (SCA) services, are used for the
transmission of stock market data, background music, control signals, and other in-
formation not normally part of the station's main programming. Although these ser-
vices do not provide the same range of coverage or audio fidelity as the main stereo

© 1999 CRC Press LLC

program, they perform a public service and can be a valuable source of income for the
broadcaster.

SCA systems provide efficient use of the available spectrum. The most common

subcarrier frequency is 67 kHz, although higher subcarrier frequencies may be used.
Stations that operate subcarrier systems are permitted by the FCC to exceed (by a small
amount) the maximum 75 kHz deviation limit, under certain conditions. The
subcarriers utilize low modulation levels, and the energy produced is maintained essen-
tially within the 200 kHz bandwidth limitation of FM channel radiation.

1.2.5 TV Broadcasting

TV transmitters in the United States operate in three frequency bands:

·

Low-band VHF--channels 2 through 6 (54 to 72 MHz and 76 to 88 MHz).

·

High-band VHF--channels 7 through 13 (174 to 216 MHz).

·

UHF--channels 14 through 69 (470 to 806 MHz). UHF channels 70 through 83
(806 to 890 MHz) currently are assigned to land mobile radio services. Certain
TV translators may continue to operate on these frequencies on a secondary basis.

Because of the wide variety of operating parameters for TV stations outside the

United States, this section will focus primarily on TV transmission as it relates to the
United States.

Table 1.1

lists the frequencies used by TV broadcasting. Maximum

power output limits are specified by the FCC for each type of service. The maximum ef-

Figure 1.4 Composite baseband stereo FM signal. A full left-only or right-only signal will
modulate the main (L+R) channel to a maximum of 45 percent. The stereophonic
subchannel is composed of upper sideband (USB) and lower sideband (LSB) compo-
nents.

© 1999 CRC Press LLC

fective radiated power for low-band VHF is 100 kW; for high-band VHF it is 316 kW;
and for UHF it is 5 MW.

The second major factor that affects the coverage area of a TV station is antenna

height, known in the industry as height above average terrain (HAAT). HAAT takes
into consideration the effects of the geography in the vicinity of the transmitting tower.
The maximum HAAT permitted by the FCC for a low- or high-band VHF station is
1000 ft (305 m) east of the Mississippi River and 2000 ft (610 m) west of the Missis-
sippi. UHF stations are permitted to operate with a maximum HAAT of 2000 ft (610 m)
anywhere in the United States (including Alaska and Hawaii).

Table 1.1 Channel Designations for VHF- and UHF-TV Stations in the United States

Channel

Frequency

(MHz)

Channel

Frequency

(MHz)

Channel

Frequency

(MHz)

2

54 60

30

566 572

58

734 740

3

60 66

31

572 578

59

740 746

4

66 72

32

578 584

60

746 752

5

76 82

33

584 590

61

752 758

6

82 88

34

590 596

62

758 764

7

174 180

35

596 602

63

764 770

8

180 186

36

602 608

64

770 776

9

186 192

37

608 614

65

776 782

10

192 198

38

614 620

66

782 788

11

198 204

39

620 626

67

788 794

12

204 210

40

626 632

68

794 800

13

210 216

41

632 638

69

800 806

14

470 476

42

638 644

70

806 812

15

476 482

43

644 650

71

812 818

16

482 488

44

650 656

72

818 824

17

488 494

45

656 662

73

824 830

18

494 500

46

662 668

74

830 836

19

500 506

47

668 674

75

836 842

20

506 512

48

674 680

76

842 848

21

512 518

49

680 686

77

848 854

22

518 524

50

686 692

78

854 860

23

524 530

51

692 698

79

860 866

24

530 536

52

698 704

80

866 872

25

536 542

53

704 710

81

872 878

26

542 548

54

710 716

82

878 884

27

548 554

55

716 722

83

884 890

28

554 560

56

722 728

29

560 566

57

728 734

© 1999 CRC Press LLC

The ratio of visual output power to aural power may vary from one installation to an-

other, but the aural typically is operated at between 10 to 20 percent of the visual power.
This difference is the result of the reception characteristics of the two signals. Much
greater signal strength is required at the consumer's receiver to recover the visual por-
tion of the transmission than the aural portion. The aural power output is intended to be
sufficient for good reception at the fringe of the station's coverage area, but not beyond.
It is pointless for a consumer to be able to receive a TV station's audio signal, but not the
video.

1.2.6 Satellite Transmission

Commercial satellite communication began on July 10, 1962, when TV pictures were
first beamed across the Atlantic Ocean through the Telstar 1 satellite. Three years
later, the Intelsat system of geostationary relay satellites saw its initial craft, Early
Bird 1, launched into a rapidly growing communications industry. In the same year,
the U.S.S.R. inaugurated the Molnya series of satellites, traveling in an elliptical orbit
to better meet the needs of that nation. The Molnya satellites were placed in an orbit
inclined about 64°, relative to the equator, with an orbital period half that of the earth.

All commercial satellites in use today operate in a geostationary orbit. A geostation-

ary satellite is one that maintains a fixed position in space relative to earth because of its
altitude, roughly 22,300 miles above the earth. Two primary frequency bands are used:
the C-band (4 to 6 GHz) and the Ku-band (11 to 14 GHz). Any satellite relay system in-
volves three basic sections:

·

An uplink transmitting station, which beams signals toward the satellite in its
equatorial geostationary orbit

·

The satellite (the space segment of the system), which receives, amplifies, and re-
transmits the signals back to earth

·

The downlink receiving station, which completes the relay path

Because of the frequencies involved, satellite communications is designated as a mi-

crowave radio service. As such, certain requirements are placed upon the system. As
with terrestrial microwave, the path between transmitter and receiver must be line of
sight. Meteorological conditions, such as rain and fog, result in detrimental attenuation
of the signal. Arrangements must be made to shield satellite receive antennas from ter-
restrial interference. Because received signal strength is based upon the inverse square
law, highly directional transmit and receive parabolic antennas are used, in turn requir-
ing a high degree of aiming accuracy. To counteract the effects of galactic and thermal
noise sources on low-level signals, amplifiers are designed for exceptional low noise
characteristics.

Figure 1.5

shows the primary elements of a satellite relay system.

Satellite Link

Like other relay stations, the communications spacecraft contains antennas for receiv-
ing and retransmission. From the receive antenna, signals pass through a low-noise

© 1999 CRC Press LLC

amplifier before frequency conversion to the transmit band. A high-power amplifier
(HPA) feeds the received signal to a directional antenna, which beams the information
to a predetermined area of the earth to be served by the satellite, as illustrated in

Fig-

ure 1.6

.

Power to operate the electronics hardware is generated by solar cells. Inside the sat-

ellite, storage batteries, kept recharged by the solar cell arrays, carry the electronic load,
particularly when the satellite is eclipsed by the earth. Power to the electronics on the
craft requires protective regulation to maintain consistent signal levels. Most of the
equipment operates at low voltages, but the final stage of each transponder chain ends
in a high-power amplifier. The HPA of C-band satellite channels may include a travel-
ing wave tube (TWT) or a solid-state power amplifier (SSPA). Ku-band systems rely
primarily on TWT devices at this writing. Klystrons and TWTs require multiple volt-
age levels. The filaments operate at low voltages, but beam focus and electron collec-
tion electrodes require potentials in the hundreds and thousands of volts. To develop
such a range of voltages, the satellite power supply includes voltage converters.

From these potentials, the klystron or TWT produces output powers in the range of

8.5 to 20 W. Most systems are operated at the lower end of the range to increase reliabil-
ity and life expectancy. In general, the lifetime of the spacecraft is assumed to be 7
years.

A guidance system is included to stabilize the attitude of the craft as it rotates around

the earth. Small rocket engines are provided for maintaining an exact position in the as-
signed geostationary arc. This work is known as station-keeping.

Figure 1.5 Principal elements of a satellite communications link.

© 1999 CRC Press LLC

1.2.7 Radar

The word radar is an acronym for radio detection and ranging. The name accurately
spells out the basic function of a radar system. Measurement of target angles is an ad-
ditional function of most radar equipment. Doppler velocity also may be measured as
an important parameter. A block diagram of a typical pulsed radar system is shown in

Figure 1.7

. Any system can be divided into six basic subsections:

·

Exciter and synchronizer: controls the sequence of transmission and reception
functions.

·

Transmitter: generates a high-power RF pulse of specified frequency and shape.

·

Microwave network: couples the transmitter and receiver sections to the antenna.

·

Antenna system: consists of a radiating/receiving structure mounted on a me-
chanically steered servo-driven pedestal. A stationary array, which uses electri-

Figure 1.6 The power levels in transmission of a video signal via satellite.

© 1999 CRC Press LLC

cal steering of the antenna system, may be used in place of the mechanical system
shown in the figure.

·

Receiver: selects and amplifies the return pulse picked up by the antenna.

·

Signal processor and display: integrates the detected echo pulse, synchronizer
data, and antenna pointing data for presentation to an operator.

Radar technology is used for countless applications.

Table 1.2

lists some of the more

common uses.

Operating Parameters

Because radar systems have many diverse applications, the parameters of frequency,
power, and transmission format also vary widely. There are no fundamental bounds
on the operating frequencies of radar. In fact, any system that locates objects by de-
tecting echoes scattered from a target that has been illuminated with electromagnetic
energy can be considered radar. Although the principles of operation are similar re-
gardless of the frequency, the functions and circuit parameters of most radar systems
can be divided into specific operating bands.

Table 1.3

shows the primary bands in

use today. As shown in the table, letter designations have been developed for most of
the operating bands.

Radar frequencies have been selected to minimize atmospheric attenuation by rain

and snow, clouds and fog, and (at some frequencies) electrons in the air. The frequency
bands must also support wide bandwidth radiation and high antenna gain.

Figure 1.7 Simplified block diagram of a pulsed radar system.

© 1999 CRC Press LLC

Transmission Equipment

The operating parameters of a radar transmitter are entirely different from those of the
other transmitters discussed so far. Broadcast and satellite systems are characterized
by medium-power, continuous-duty applications. Radar, on the other hand, is charac-

Table 1.3 Standard Radar Frequency Bands

Table 1.2 Typical Radar Applications

Air surveillance

Long-range early warning
Ground-controlled intercept
Target acquisition
Height finding and three-dimensional
analysis
Airport and air-route management

Space and missile surveillance

Ballistic missile warning
Missile acquisition
Satellite surveillance

Surface-search and military surveillance

Sea search and navigation
Ground mapping
Artillery location
Airport taxiway control

Weather forecasting/tracking

Observation and prediction
Aircraft weather avoidance
Cloud-visibility determination

Tracking and guidance

Antiaircraft fire control
Surface fire control
Missile guidance
Satellite instrumentation
Aircraft approach and landing

Frequency Band

Frequency Range Radiolocation Bands Based on ITU As-

signments in Region II

VHF

30 300 MHz

137 134 MHz

UHF

300 1000 MHz

216 255 MHz

L-band

1.0 2.0 GHz

1.215 1.4 GHz

S-band

2.0 4.0 GHz

2.3 2.55 GHz, 2.7 3.7 GHz

C-band

4.0 8.0 GHz

5.255 5.925 GHz

X-band

8.0 12.5 GHz

8.5 10.7 GHz

Ku-band

12.5 18.0 GHz

13.4 14.4 GHz, 15.7 17.7 GHz

K-band

18.0 26.5 GHz

23 24.25 GHz

Ka-band

26.5 40 GHz

33.4 36 GHz

Millimeter

Above 40 GHz

© 1999 CRC Press LLC

terized by high-power pulsed transmissions of relatively low duty cycle. The unique
requirements of radar have led to the development of technology that is foreign to
most communications systems.

Improvements in semiconductor design and fabrication have made solid-state radar

sets possible. Systems producing several hundred watts of output power at frequencies
up to 2 GHz have been installed. Higher operating powers are achieved by using paral-
lel amplification.

Despite inroads made by solid-state devices, vacuum tubes continue to be the main-

stay of radar technology. Tube-based systems consist of the following stages:

·

Exciter: generates the necessary RF and local-oscillator frequencies for the sys-
tem.

·

Power supply: provides the needed operating voltages for the system.

·

Modulator: triggers the power output tube into operation. Pulse-shaping of the
transmitted signal is performed in the modulator stage.

·

RF amplifier: converts the dc input from the power supply and the trigger signals
from the modulator into a high-energy, short-duration pulse.

1.2.8 Electronic Navigation

Navigation systems based on radio transmissions are used every day by commercial
airlines, general aviation aircraft, ships, and the military. Electronic position-fixing
systems also are used in surveying work. Although the known speed of propagation of
radio waves allows good accuracies to be obtained in free space, multipath effects
along the surface of the earth are the primary enemies of practical airborne and
shipborne systems. A number of different navigation tools, therefore, have evolved to
obtain the needed accuracy and coverage area.

Electronic navigation systems can be divided into three primary categories:

·

Long-range, useful for distances of greater than 200 miles. Long-range systems
are used primarily for transoceanic navigation.

·

Medium-range, useful for distances of 20 to 200 miles. Medium-range systems
are used mainly in coastal areas and above populated land masses.

·

Short-range, useful for distances of less than 20 miles. Short-range systems are
used for approach, docking, or landing applications.

Electronic navigation systems can be divided further into cooperative or self-con-

tained systems. Cooperative systems depend on one- or two-way transmission between
one or more ground stations and the vehicle. Such systems are capable of providing the
vehicle with a location fix, independent of its previous position. Self-contained sys-
tems, housed entirely within the vehicle, may be radiating or nonradiating. They typi-
cally are used to measure the distance traveled, but have errors that increase with dis-
tance and/or time. The type of system chosen for a particular application depends upon

© 1999 CRC Press LLC

a number of considerations, including how often the location of the vehicle must be de-
termined and how much accuracy is required.

Because aircraft and ships may travel to any part of the world, many electronic navi-

gation systems have received standardization on an international scale.

Virtually all radio frequencies have been used in navigation at one point or another.

Systems operating at low frequencies typically use high-power transmitters with mas-
sive antenna systems. With a few exceptions, frequencies and technologies have been
chosen to avoid dependence on ionospheric reflection. Such reflections can be valuable
in communications systems, but are usually unpredictable.

Table 1.4

lists the principal

frequency bands used for radionavigation.

Direction Finding

Direction finding (DF) is a simple and widely used navigation aid. The position of a
moving transmitter may be determined by comparing the arrival coordinates of the ra-
diated energy at two or more fixed (known) points. Conversely, the position of a re-
ceiving point may be determined by comparing the direction coordinates from two or
more known transmitters.

The weakness of this system is its susceptibility to site errors. This shortcoming may

be reduced through the use of a large DF antenna aperture. In many cases, a multiplicity
of antennas, suitably combined, can be made to favor the direct path and discriminate
against indirect paths, as illustrated in

Figure 1.8

.

Ship navigation is a common application of DF. Coastal beacons operate in the 285

to 325 kHz band specifically for ship navigation. This low frequency provides ground
wave coverage over seawater to about 1000 miles. Operating powers vary from 100 W
to 10 kW. A well-designed shipboard DF system can provide accuracies of about ±2°
under typical conditions.

Two-Way Distance Ranging

Automatic distance measuring can be accomplished through placement of a tran-
sponder on a given target, as illustrated in

Figure 1.9

. The system receives an interro-

gator pulse and replies to it with another pulse, usually on a different frequency. Vari-
ous codes can be employed to limit responses to a single target or class of target.

Distance-measuring equipment (DME) systems are one application of two-way dis-

tance ranging. An airborne interrogator transmits 1 kW pulses at a 30 Hz rate on one of
126 channels spaced 1 MHz apart. (The operating band is 1.025 to 1.150 GHz.) A
ground transponder responds with similar pulses on a different channel (63 MHz above
or below the interrogating channel).

In the airborne set, the received signal is compared with the transmitted signal. By

determining the time difference of the two pulses, a direct reading of miles may be
found. Typical accuracy of this system is ±0.2 miles.

Ground transponders usually are configured to handle interrogation from up to 100

aircraft simultaneously.

© 1999 CRC Press LLC

Figure 1.8 Direction-finding error resulting from beacon reflections.

Table 1.4 Radio Frequencies Used for Electronic Navigation

System

Frequency Band

Omega

10 13 kHz

VLF

16 24 kHz

Decca

70 130 kHz

Loran C/D

100 kHz

ADF/NDB

200 1600 kHz

Coastal DF

285 325 kHz

Consol

250 350 kHz

Marker beacon

75 MHz

ILS localizer

108 112 MHz

VOR

108 118 MHz

ILS glide slope

329 335 MHz

DME, Tacan

960 1210 MHz

ATCRBS

1.03 1.09 GHz

GPS

1.227 1.575 GHz

Altimeter

4.2 GHz

Talking beacon

9.0 GHz

MLS

5.0 GHz

© 1999 CRC Press LLC

Differential Distance Ranging

The differential distance ranging system requires two widely-spaced transmitters, one
at each end of a predefined link. By placing two transmitters on the ground, it is un-
necessary to carry a transmitter in the vehicle. One transmitter is the master and the
other is a slave repeating the master (see

Figure 1.10

). The mobile receiver measures

the time difference in arrival of the two signals. For each time difference, there is a hy-
perbolic line of position that defines the target location. (Such systems are known as
hyperbolic systems.) The transmissions may be either pulsed or continuous wave us-
ing different carrier frequencies. A minimum of two stations is required to produce a
fix.

If both stations in a differential distance ranging system are provided with stable,

synchronized clocks, distance measurements can be accomplished through one-way
transmissions whose elapsed time is measured with reference to the clocks. This mode
of operation is referred to as one-way distance ranging and is illustrated in

Figure 1.11

.

Loran C

Hyperbolic positioning is used in the Loran C navigation system. (Loran is named af-
ter its intended application, long-range navigation.) Chains of transmitters, located
along coastal waters, radiate pulses at a carrier frequency of 100 kHz. Because all sta-
tions operate on the same frequency, discrimination among chains is accomplished by
different pulse-repetition frequencies. A typical chain consists of a master station and
two slaves, about 600 miles from the master. Each antenna is approximately 1300 ft
high and is fed 5 MW pulses of defined rise and decay characteristics. Shaping of the
rise and decay times is necessary to keep the radiated spectrum within the assigned
band limits of 90 to 100 kHz.

Figure 1.9 The concept of two-way distance ranging.

© 1999 CRC Press LLC

Pulsed transmissions are used to obtain greater average power at the receiver with-

out resorting to higher peak power at the transmitters. The master station transmits
groups of nine pulses 1 ms apart, and the slaves transmit groups of eight pulses 1 ms
apart. These groups are repeated at a rate of 10 to 25 per second. Within each pulse, the
phase of the RF carrier can be varied for communications purposes.

Coverage of Loran C extends to all U.S. coastal areas, plus certain areas of the North

Pacific, North Atlantic, and Mediterranean. Currently, more than 17 chains employ
about 50 transmitters.

Figure 1.11 The concept of one-way distance ranging.

Figure 1.10 The concept of differential distance ranging (hyperbolic).

© 1999 CRC Press LLC

Omega

Omega is another navigation system based on the hyperbolic concept. The system is
designed to provide worldwide coverage from just eight stations. Omega operates on
the VLF band, from 10 to 13 kHz. Skywave propagation is relatively stable for these
frequencies, providing an overall system accuracy on the order of 1 mile, even at
ranges of 5000 miles.

There are no masters or slaves in the Omega system; stations transmit according to

their own standard. Each station has its particular operating code and transmits on one
frequency at a time for a minimum of about 1 s. The cycle is repeated every 10 s. These
slow rates are necessary because of the high Qs of the transmitting antennas.

A simple Omega receiver monitors for signals at 10.2 kHz and compares emissions

from one station to those of another by use of an internal oscillator. The phase differ-
ence data is transferred to a map with hyperbolic coordinates.

Most Omega receivers also are able to use VLF communications stations for naviga-

tion. There are about 10 such facilities operating from 16 to 24 kHz. Output powers
range from 50 kW to 1 MW. Frequency stability is maintained to 1 part in 10

12

. This al-

lows one-way DME to be accomplished with a high degree of accuracy.

GPS

The Global Positioning System (NAVSTAR GPS) is a U.S. Department of Defense
(DoD) developed, worldwide, satellite-based radionavigation system. The constella-
tion consists of 24 operational satellites. GPS full operational capability was declared
on July 17, 1995.

GPS provides two levels of service--a Standard Positioning Service (SPS) and a

Precise Positioning Service (PPS). SPS is a positioning and timing service that is avail-
able to all GPS users on a continuous, worldwide basis. SPS is provided on the GPS L1
frequency, which contains a coarse acquisition (C/A) code and a navigation data mes-
sage. SPS provides, on a daily basis, the capability to obtain horizontal positioning ac-
curacy within 100 meters (95 percent probability) and 300 meters (99.99 percent prob-
ability), vertical positioning accuracy within 140 meters (95 percent probability), and
timing accuracy within 340 ns (95 percent probability).

The GPS L1 frequency also contains a precision (P) code that is not a part of the SPS.

PPS is a highly accurate military positioning, velocity, and timing service that is avail-
able on a continuous, worldwide basis to users authorized by the DoD. PPS is the data
transmitted on GPS L1 and L2 frequencies. PPS is designed primarily for U.S. military
use. P-code-capable military user equipment provides a predictable positioning accu-
racy of at least 22 meters horizontally and 27.7 meters vertically, and timing/time inter-
val accuracy within 90 ns (95 percent probability).

The GPS satellites transmit on two L-band frequencies: L1 = 1575.42 MHz and L2 =

1227.6 MHz. Three pseudo-random noise (PRN) ranging codes are used. The
coarse/acquisition (C/A) code has a 1.023 MHz chip rate, a period of 1 ms, and is used
by civilian users for ranging and by military users to acquire the P-code. Bipolar-phase
shift key (BPSK) modulation is utilized. The transmitted PRN code sequence is actu-
ally the modulo-2 addition of a 50 Hz navigation message and the C/A code. The SPS

© 1999 CRC Press LLC

receiver demodulates the received code from the L1 carrier, and detects the differences
between the transmitted and the receiver-generated code. The SPS receiver uses an ex-
clusive-OR truth table to reconstruct the navigation data, based upon the detected dif-
ferences in the two codes.

The precision (P) code has a 10.23 MHz rate, a period of seven days and is the princi-

ple navigation ranging code for military users. The Y-code is used in place of the P-code
whenever the anti-spoofing (A-S) mode of operation is activated. The C/A code is
available on the L1 frequency and the P-code is available on both L1 and L2. The vari-
ous satellites all transmit on the same frequencies, L1 and L2, but with individual code
assignments.

Each satellite transmits a navigation message containing its orbital elements, clock

behavior, system time, and status messages. In addition, an almanac is provided that
gives the approximate data for each active satellite. This allows the user set to find all
satellites after the first has been acquired.

1.2.9 Microwave Radio

Microwave radio relay systems carry a significant portion of long-haul telecommuni-
cations in the United States and other countries.

Table 1.5

lists the major com-

mon-carrier bands and their typical uses. The goal of microwave relay technology has
been to increase channel capacity and lower costs. Solid-state devices have provided
the means to accomplish this goal. Current efforts focus on the use of fiber optic land-
lines for terrestrial long-haul communications systems. Satellite circuits also have
been used extensively for long-distance common-carrier applications.

Single-sideband amplitude modulation is used for microwave systems because of its

spectrum efficiency. Single-sideband systems, however, require a high degree of lin-
earity in amplifying circuits. Several techniques have been used to provide the needed
channel linearity. The most popular is amplitude predistortion to cancel the inherent
nonlinearity of the power amplifier.

Microwave relay systems typically use frequency-division multiplexing (FDM) to

combine signals for more efficient transmission. For example, three 600-channel mas-
ter groups may be multiplexed into a single baseband signal. This multiplexing often is
done some distance (a mile or more) from the transmitter site. A coaxial wireline is

Table 1.5 Common-Carrier Microwave Frequencies Used in the United States

Band (GHz)

Allotted frequencies

Bandwidth

Applications

2

2.11 2.13 Hz
2.16 2.18 GHz

20 MHz

General purpose, limited
use

4

3.7 4.2 GHz

20 MHz

Long-haul point-to-point re-
lay

6

5.925 6.425 GHz

500 MHz

Long and short haul

11

10.7 11.7 GHz

500 MHz

Short haul

18

17.7 19.7 GHz

1.0 GHz

Short haul, limited use

© 1999 CRC Press LLC

used to bring the baseband signal to the microwave equipment. Depending on the dis-
tance, intermediate repeaters may be used. The baseband signal is applied to an FM ter-
minal (FMT) that frequency-modulates a carrier of typically 70 MHz. This IF signal
then modulates a 20-MHz-wide channel in the 4 GHz band.

1.2.10 Induction Heating

Induction heating is achieved by placing a coil carrying alternating current adjacent
to a metal workpiece so that the magnetic flux produced induces a voltage in the
workpiece. This causes current flow and heats the workpiece. Power sources for in-
duction heating include:

·

Motor-generator sets, which operate at low frequencies and provide outputs from
1 kW to more than 1 MW.

·

Vacuum tube oscillators, which operate at 3 kHz to several hundred megahertz at
power levels of 1 kW to several hundred kilowatts.

Figure 1.12

shows a 20 kW in-

duction heater using a vacuum tube as the power generating device.

·

Inverters, which operate at 10 kHz or more at power levels of as much as several
megawatts. Inverters using thyristors (silicon controlled rectifiers) are replacing
motor-generator sets in most high-power applications.

Figure 1.12 Basic schematic of a 20 kW induction heater circuit.

© 1999 CRC Press LLC

Dielectric Heating

Dielectric heating is a related application for RF technology. Instead of heating a con-
ductor, as in induction heating, dielectric heating relies on the capacitor principle to
heat an insulating material. The material to be heated forms the dielectric of a capaci-
tor, to which power is applied. The heat generated is proportional to the loss factor
(the product of the dielectric constant and the power factor) of the material. Because
the power factor of most dielectrics is low at low frequencies, the range of frequencies
employed for dielectric heating is higher than for induction heating. Frequencies of a
few megahertz to several gigahertz are common.

1.2.11 Electromagnetic Radiation Spectrum

The usable spectrum of electromagnetic radiation frequencies extends over a range
from below 100 Hz for power distribution to 10

20

Hz for the shortest X rays (see

Fig-

ure

1.13

). The lower frequencies are used primarily for terrestrial broadcasting and

communications. The higher frequencies include visible and near-visible infrared and
ultraviolet light, and X rays. The frequencies of interest to RF engineers range from
30 kHz to 30 GHz. (See

Table 1.6

.)

Figure 1.13 The electromagnetic spectrum.

© 1999 CRC Press LLC

Low Frequency (LF): 30 to 300 kHz

The LF band is used for around-the-clock communications services over long dis-
tances and where adequate power is available to overcome high levels of atmospheric
noise. Applications include:

·

Radionavigation

·

Fixed/maritime communications and navigation

·

Aeronautical radionavigation

·

Low-frequency broadcasting (Europe)

·

Underwater submarine communications (10 to 30 kHz)

Medium Frequency (MF): 300 kHz to 3 MHz

The low-frequency portion of this band is used for around-the-clock communication
services over moderately long distances. The upper portion of the MF band is used
principally for moderate-distance voice communications. Applications in this band
include:

·

AM radio broadcasting (535.5 to 1605.5 kHz)

·

Radionavigation

·

Fixed/maritime communications

·

Aeronautical radionavigation

·

Fixed and mobile commercial communications

Table 1.6 Frequency Band Designations

Description

Designation

Frequency

Extremely Low Frequency

ELF (1) Band

3 Hz up to 30 Hz

Super Low Frequency

SLF (2) Band

30 Hz up to 300 Hz

Ultra Low Frequency

ULF (3) Band

300 Hz up to 3 kHz

Very Low Frequency

VLF (4) Band

3 kHz up to 30 kHz

Low Frequency

LF (5) Band

30 kHz up to 300 kHz

Medium Frequency

MF (6) Band

300 kHz up to 3 MHz

High Frequency

HF (7) Band

3 MHz up to 30 MHz

Very High Frequency

VHF (8) Band

30 MHz up to 300 MHz

Ultra High Frequency

UHF (9) Band

300 MHz up to 3 GHz

Super High Frequency

SHF (10) Band

3 GHz up to 30 GHz

Extremely High Frequency

EHF (11) Band

30 GHz up to 300 GHz

--

-- (12) Band

300 GHz up to 3 THz

© 1999 CRC Press LLC

·

Amateur radio

·

Standard time and frequency services

High Frequency (HF): 3 to 30 MHz

This band provides reliable medium-range coverage during daylight and, when the
transmission path is in total darkness, worldwide long-distance service. The reliabil-
ity and signal quality of long-distance service depends to a large degree upon iono-
spheric conditions and related long-term variations in sunspot activity affecting
skywave propagation. Applications include:

·

Shortwave broadcasting

·

Fixed and mobile service

·

Telemetry

·

Amateur radio

·

Fixed/maritime mobile

·

Standard time and frequency services

·

Radio astronomy

·

Aeronautical fixed and mobile

Very High Frequency (VHF): 30 to 300 MHz

The VHF band is characterized by reliable transmission over medium distances. At
the higher portion of the VHF band, communication is limited by the horizon. Appli-
cations include:

·

FM radio broadcasting (88 to 108 MHz)

·

Low-band VHF-TV broadcasting (54 to 72 MHz and 76 to 88 MHz)

·

High-band VHF-TV broadcasting (174 to 216 MHz)

·

Commercial fixed and mobile radio

·

Aeronautical radionavigation

·

Space research

·

Fixed/maritime mobile

·

Amateur radio

·

Radiolocation

© 1999 CRC Press LLC

Ultrahigh Frequency (UHF): 300 MHz to 3 GHz

Transmissions in this band are typically line of sight. Short wavelengths at the upper
end of the band permit the use of highly directional parabolic or multielement anten-
nas. Applications include:

·

UHF terrestrial television (470 to 806 MHz)

·

Fixed and mobile communications

·

Telemetry

·

Meteorological aids

·

Space operations

·

Radio astronomy

·

Radionavigation

·

Satellite communications

·

Point-to-point microwave relay

Superhigh Frequency (SHF): 3 to 30 GHz

Communication in this band is strictly line of sight. Very short wavelengths permit
the use of parabolic transmit and receive antennas of exceptional gain. Applications
include:

·

Satellite communications

·

Point-to-point wideband relay

·

Radar

·

Specialized wideband communications

·

Developmental research

·

Military support systems

·

Radiolocation

·

Radionavigation

·

Space research

1.3

Bibliography

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land Park, KS, June 1986.

Benson, K. B., and J. C. Whitaker (eds.), Television Engineering Handbook, revised

ed., McGraw-Hill, New York, 1991.

© 1999 CRC Press LLC

Benson, K. B., and J. C. Whitaker, Television and Audio Handbook for Technicians

and Engineers, McGraw-Hill, New York, 1989.

Brittain, James E., "Scanning the Past: Guglielmo Marconi," Proceedings of the IEEE,

Vol. 80, no. 8, August 1992.

Burke, William, "WLW: The Nation's Station," Broadcast Engineering, Intertec Pub-

lishing, Overland Park, KS, November 1967.

Clerc, Guy, and William R. House, "The Case for Tubes," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, pp. 67 - 82, May 1991.

Dorschug, Harold, "The Good Old Days of Radio," Broadcast Engineering, Intertec

Publishing, Overland Park, KS, May 1971.

Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 3rd ed.,

McGraw-Hill, New York, 1989.

Fink, D., and D. Christiansen (eds.), Electronics Engineer's Handbook, 2nd ed.,

McGraw-Hill, New York, 1982.

Goldstein, Irving, "Communications Satellites: A Revolution in International Broad-

casting," Broadcast Engineering, Intertec Publishing, Overland Park, KS, May
1979.

Jordan, Edward C. (ed.), Reference Data for Engineers: Radio, Electronics, Computer

and Communications, 7th Ed., Howard W. Sams Company, Indianapolis, IN, 1985.

McCroskey, Donald, "Setting Standards for the Future," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, May 1989.

Nelson, Cindy, "RCA Demolishes Last Antenna Tower at Historic Radio Central,"

Broadcast Engineering, Intertec Publishing, Overland Park, KS, February 1978.

Paulson, Robert, "The House That Radio Built," Broadcast Engineering, Intertec Pub-

lishing, Overland Park, KS, April 1989.

Peterson, Benjamin B., "Electronic Navigation Systems," in The Electronics Hand-

book, Jerry C. Whitaker (ed.), CRC Press, Boca Raton, Fla., pp. 17101733, 1996.

Pond, N. H., and C. G. Lob, "Fifty Years Ago Today or On Choosing a Microwave

Tube," Microwave Journal, September 1988.

Riggins, George, "The Real Story on WLW's Long History," Radio World, Falls

Church, VA, March 8, 1989.

Schow, Edison, "A Review of Television Systems and the Systems for Recording

Television," Sound and Video Contractor, Intertec Publishing, Overland Park, KS,
May 1989.

Schubin, Mark, "From Tiny Tubes to Giant Screens," Video Review, April 1989.
Stokes, John W., Seventy Years of Radio Tubes and Valves, Vestal Press, New York,

1982.

Symons, Robert S., "Tubes--Still Vital After All These Years," IEEE Spectrum,

IEEE, New York, pp. 5363, April 1998.

Terman, Frederick E., Radio Engineering, 3rd ed., McGraw-Hill, New York, 1947.
Varian Associates, "An Early History," Varian, Palo Alto, CA.
Whitaker, J. C., Maintaining Electronic Systems, CRC Press, Boca Raton, FL, 1992.
Whitaker, J. C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1991.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

2 Modulation Systems and Characterisitics"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

2

Modulation Systems and

Characteristics

2.1

Introduction

Radio frequency (RF) power amplifiers are used in countless applications at tens of
thousands of facilities around the world. The wide variety of applications, however,
stem from a few basic concepts of conveying energy and information by means of an
RF signal. Furthermore, the devices used to produce RF energy have many similari-
ties, regardless of the final application. Although communications systems represent
the most obvious use of high-power RF generators, numerous other common applica-
tions exist, including:

·

Induction heating and process control systems

·

Radar (ground, air, and shipboard)

·

Satellite communications

·

Atomic science research

·

Medical research, diagnosis, and treatment

The process of generating high-power RF signals has been refined over the years to

an exact science. Advancements in devices and circuit design continue to be made each
year, pushing ahead the barriers of efficiency and maximum operating frequency. Al-
though different applications place unique demands on the RF design engineer, the fun-
damental concepts of RF amplification are applicable to virtually any system.

2.1.1 Modulation Systems

The primary purpose of most communications and signaling systems is to transfer in-
formation from one location to another. The message signals used in communication
and control systems usually must be limited in frequency to provide for efficient
transfer. This frequency may range from a few hertz for control systems to a few
megahertz for video signals to many megahertz for multiplexed data signals. To facil-

© 1999 CRC Press LLC

itate efficient and controlled distribution of these components, an encoder generally
is required between the source and the transmission channel. The encoder acts to
modulate the signal, producing at its output the modulated waveform. Modulation is a
process whereby the characteristics of a wave (the carrier) are varied in accordance
with a message signal, the modulating waveform. Frequency translation is usually a
by-product of this process. Modulation may be continuous, where the modulated
wave is always present, or pulsed, where no signal is present between pulses.

There are a number of reasons for producing modulated waves, including:

·

Frequency translation. The modulation process provides a vehicle to perform the
necessary frequency translation required for distribution of information. An in-
put signal may be translated to its assigned frequency band for transmission or ra-
diation.

·

Signal processing. It is often easier to amplify or process a signal in one fre-
quency range as opposed to another.

·

Antenna efficiency. Generally speaking, for an antenna to be efficient, it must be
large compared with the signal wavelength. Frequency translation provided by
modulation allows antenna gain and beamwidth to become part of the system de-
sign considerations. The use of higher frequencies permits antenna structures of
reasonable size and cost.

·

Bandwidth modification. The modulation process permits the bandwidth of the
input signal to be increased or decreased as required by the application. Band-
width reduction permits more efficient use of the spectrum, at the cost of signal fi-
delity. Increased bandwidth, on the other hand, provides increased immunity to
transmission channel disturbances.

·

Signal multiplexing. In a given transmission system, it may be necessary or desir-
able to combine several different signals into one baseband waveform for distri-
bution. Modulation provides the vehicle for such multiplexing. Various modula-
tion schemes allow separate signals to be combined at the transmission end and
separated (demultiplexed) at the receiving end. Multiplexing may be accom-
plished by using, among other systems, frequency-domain multiplexing (FDM)
or time-domain multiplexing (TDM).

Modulation of a signal does not come without the possible introduction of undesir-

able attributes. Bandwidth restriction or the addition of noise or other disturbances are
the two primary problems faced by the transmission system designer.

2.1.2 Principles of Resonance

All RF generations rely on the principles of resonance for operation. Three basic sys-
tems exist:

·

Series resonance circuits

·

Parallel resonance circuits

© 1999 CRC Press LLC

·

Cavity resonators

Series Resonant Circuits

When a constant voltage of varying frequency is applied to a circuit consisting of an
inductance, capacitance, and resistance (all in series), the current that flows depends
upon frequency in the manner shown in

Figure 2.1

. At low frequencies, the capacitive

reactance of the circuit is large and the inductive reactance is small, so that most of the
voltage drop is across the capacitor, while the current is small and leads the applied
voltage by nearly 90°. At high frequencies, the inductive reactance is large and the ca-
pacitive reactance is low, resulting in a small current that lags nearly 90° behind the
applied voltage; most of the voltage drop is across the inductance. Between these two
extremes is the resonant frequency, at which the capacitive and inductive reactances
are equal and, consequently, neutralize each other, leaving only the resistance of the
circuit to oppose the flow of current. The current at this resonant frequency is, accord-
ingly, equal to the applied voltage divided by the circuit resistance, and it is very large
if the resistance is low.

The characteristics of a series resonant circuit depend primarily upon the ratio of in-

ductive reactance

L to circuit resistance R, known as the circuit Q:

Q

L

R

=

(2.1)

The circuit Q also may be defined by:

Q

E

E

s

d

=

2

(2.2)

Where:
E

s

= energy stored in the circuit

E

d

= energy dissipated in the circuit during one cycle

Most of the loss in a resonant circuit is the result of coil resistance; the losses in a

properly constructed capacitor are usually small in comparison with those of the coil.

The general effect of different circuit resistances (different values of Q) is shown in

Figure 2.1

. As illustrated, when the frequency differs appreciably from the resonant fre-

quency, the actual current is practically independent of circuit resistance and is nearly
the current that would be obtained with no losses. On the other hand, the current at the
resonant frequency is determined solely by the resistance. The effect of increasing the
resistance of a series circuit is, accordingly, to flatten the resonance curve by reducing
the current at resonance. This broadens the top of the curve, giving a more uniform cur-
rent over a band of frequencies near the resonant point. This broadening is achieved,
however, by reducing the selectivity of the tuned circuit.

© 1999 CRC Press LLC

Parallel Resonant Circuits

A parallel circuit consisting of an inductance branch in parallel with a capacitance
branch offers an impedance of the character shown in

Figure 2.2

. At low frequencies,

the inductive branch draws a large lagging current while the leading current of the ca-
pacitive branch is small, resulting in a large lagging line current and a low lagging cir-
cuit impedance. At high frequencies, the inductance has a high reactance compared
with the capacitance, resulting in a large leading line current and a corresponding low
circuit impedance that is leading in phase. Between these two extremes is a frequency
at which the lagging current taken by the inductive branch and the leading current en-
tering the capacitive branch are equal. Being 180° out of phase, they neutralize, leav-
ing only a small resultant in-phase current flowing in the line; the impedance of the
parallel circuit is, therefore, high.

Figure 2.1 Characteristics of a series resonant circuit as a function of frequency for a
constant applied voltage and different circuit

Qs: (a) magnitude, (b) phase angle.

(

a)

(

b)

© 1999 CRC Press LLC

The effect of circuit resistance on the impedance of the parallel circuit is similar to

the influence that resistance has on the current flowing in a series resonant circuit, as is
evident when

Figures 2.1

and

2.2

are compared. Increasing the resistance of a parallel

circuit lowers and flattens the peak of the impedance curve without appreciably altering
the sides, which are relatively independent of the circuit resistance.

The resonant frequency F

o

of a parallel circuit can be taken as the same frequency at

which the same circuit is in series resonance:

F

LC

0

1

2

=

(2.3)

Where:
L = inductance in the circuit
C = capacitance in the circuit

Figure 2.2 Characteristics of a parallel resonant circuit as a function of frequency for dif-
ferent circuit

Qs: (a) magnitude, (b) phase angle.

(

a)

(

b)

© 1999 CRC Press LLC

When the circuit Q is large, the frequencies corresponding to the maximum imped-

ance of the circuit and to unity power factor of this impedance coincide, for all practical
purposes, with the resonant frequency defined in this way. When the circuit Q is low,
however, this rule does not necessarily apply.

Cavity Resonators

Any space completely enclosed with conducting walls may contain oscillating elec-
tromagnetic fields. Such a space also exhibits certain resonant frequencies when ex-
cited by electrical oscillations. Resonators of this type, commonly termed cavity reso-
nators, find extensive use as resonant circuits at very high frequencies and above. For
such applications, cavity resonators have a number of advantages, including:

·

Mechanical and electrical simplicity

·

High Q

·

Stable parameters over a wide range of operating conditions

If desired, a cavity resonator can be configured to develop high shunt impedance.

The simplest cavity resonators are sections of waveguides short-circuited at each

end and

g

/2 wavelengths long, where is the guide wavelength. This arrangement re-

sults in a resonance analogous to that of a 1/2-wavelength transmission line short-cir-
cuited at the receiving end.

Any particular cavity is resonant at a number of frequencies, corresponding to dif-

ferent possible field configurations that exist within the enclosure. The resonance hav-
ing the longest wavelength (lowest frequency) is termed the dominant or fundamental
resonance. In the case of cavities that are resonant sections of cylindrical or rectangular
waveguides, most of the possible resonances correspond to various modes that exist in
the corresponding waveguides.

The resonant wavelength is proportional in all cases to the size of the resonator. If

the dimensions are doubled, the wavelength corresponding to resonance will likewise
be doubled. This fact simplifies the construction of resonators of unusual shapes whose
proper dimensions cannot be calculated easily.

The resonant frequency of a cavity can be changed through one or more of the fol-

lowing actions:

·

Altering the mechanical dimensions of the cavity

·

Coupling reactance into the cavity

·

Inserting a copper paddle inside the cavity and moving it to achieve the desired
resonant frequency

Small changes in mechanical dimensions can be achieved by flexing walls, but large
changes typically require some type of sliding member. Reactance can be coupled
into the resonator through a coupling loop, thus affecting the resonant frequency. A
copper paddle placed inside the resonator affects the normal distribution of flux and
tends to raise the resonant frequency by an amount determined by the orientation of

© 1999 CRC Press LLC

the paddle. This technique is similar to the way in which copper can be used to pro-
duce small variations in the inductance of a coil.

Coupling to a cavity resonator can be achieved by means of a coupling loop or a cou-

pling electrode. Magnetic coupling is achieved by means of a small coupling loop ori-
ented so as to enclose magnetic flux lines existing in the desired mode of operation. A
current passed through such a loop then excites oscillations of this mode. Conversely,
oscillations existing in the resonator induce a voltage in such a coupling loop. The com-
bination of a coupling loop and cavity resonator is equivalent to the ordinary coupled
circuit shown in

Figure 2.3

. The magnitude of magnetic coupling can be readily con-

trolled by rotating the loop. The coupling is reduced to zero when the plane of the loop
is parallel to the magnetic flux.

The practical application of cavity resonators for VHF frequencies is discussed in

Section 4.3; resonators for microwave devices are discussed in Section 7.3.6.

2.1.3 Frequency Source

Every RF amplifier requires a stable frequency reference. At the heart of most sys-
tems is a quartz crystal. Quartz acts as a stable high Q mechanical resonator. Crystal
resonators are available for operation at frequencies ranging from 1 kHz to 300 MHz
and beyond.

The operating characteristics of a crystal are determined by the cut of the device

from a bulk "mother" crystal. The behavior of the device depends heavily on the size
and shape of the crystal, and the angle of the cut. To provide for operation at a wide
range of frequencies, different cuts, vibrating in one or more selected modes, are used.

The properties of a piezoelectric quartz crystal can be expressed in terms of three

sets of axes. (See

Figure 2.4

.) The axis joining the points at the ends of the crystal is

known as the optical axis, and electrical stresses applied in this direction exhibit no pi-
ezoelectric properties. The three axes X

, X , and X , passing through the corners

of the hexagon that forms the section perpendicular to the optical axes, are known as the
electrical axes. The three axes Y

, Y , and Y , which are perpendicular to the faces

of the crystal, are the mechanical axes.

If a flat section is cut from a quartz crystal in such a way that the flat sides are perpen-

dicular to an electrical axis, as shown in

Figure 2.4

(b) (X-cut), the mechanical stresses

along the Y-axis of such a section produce electric charges on the flat sides of the crys-
tal. If the direction of these stresses is changed from tension to compression, or vice
versa, the polarity of the charges on the crystal surfaces is reversed. Conversely, if elec-
tric charges are placed on the flat sides of the crystal by applying a voltage across the
faces, a mechanical stress is produced in the direction of the Y-axis. This property by
which mechanical and electrical properties are interconnected in a crystal is known as
the piezoelectric effect and is exhibited by all sections cut from a piezoelectric crystal.
Thus, if mechanical forces are applied across the faces of a crystal section having its flat
sides perpendicular to a Y-axis, as shown in

Figure 2.4

(a), piezoelectric charges will be

produced because forces and potentials developed in such a crystal have components
across at least one of the Y- or X-axes, respectively.

© 1999 CRC Press LLC

An alternating voltage applied across a quartz crystal will cause the crystal to vi-

brate and, if the frequency of the applied alternating voltage approximates a frequency
at which mechanical resonance can exist in the crystal, the amplitude vibrations will be
large. Any crystal has a number of such resonant frequencies that depend upon the crys-
tal dimensions, the type of mechanical oscillation involved, and the orientation of the
plate cut from the natural crystal.

Crystals are temperature-sensitive, as shown in

Figure 2.5

. The extent to which a de-

vice is affected by changes in temperature is determined by its cut and packaging. Crys-
tals also exhibit changes in frequency with time. Such aging is caused by one or both of
the following:

Figure 2.4 Cross section of a quartz crystal taken in the plane perpendicular to the opti-
cal axis: (

a) Y-cut plate, (b) X-cut plate.

Figure 2.3 Cylindrical resonator incorporating a coupling loop: (

a) orientation of loop

with respect to cavity, (

b) equivalent coupled circuit.

(

a)

(

b)

(

a)

(

b)

© 1999 CRC Press LLC

·

Mass transfer to or from the resonator surface

·

Stress relief within the device itself

·

Crystal aging is most pronounced when the device is new. As stress within the in-
ternal structure is relieved, the aging process slows.

Frequency Stabilization

The stability of a quartz crystal is inadequate for most commercial and industrial ap-
plications. Two common methods are used to provide the required long-term fre-
quency stability:

·

Oven-controlled crystal oscillator--The crystal is installed in a tempera-
ture-controlled box. Because the temperature in the box is constant, controlled by
a thermostat, the crystal remains on-frequency. The temperature of the enclosure
usually is set to the turnover temperature of the crystal. (The turnover point is il-
lustrated in

Figure 2.5

.)

·

Temperature-compensated crystal oscillator (TCXO)--The frequency vs. tem-
perature changes of the crystal are compensated by varying a load capacitor. A
thermistor network typically is used to generate a correction voltage that feeds a

Figure 2.5 The effects of temperature on two types of AT-cut crystals.

© 1999 CRC Press LLC

voltage-controlled capacitor (varactor) to retune the crystal to the desired on-fre-
quency value.

Equivalent Circuit of a Quartz Resonator

Because of the piezoelectric effect, the quartz resonator behaves physically as a vi-
brating object when driven by a periodic electric signal near a resonant frequency of
the cavity [1,2]. This resonant behavior may be characterized by equivalent electrical
characteristics, which may then be used to accurately determine the response and per-
formance of the electromechanical system. Dye and Butterworth (1926) [3] devel-
oped the equivalent lump electrical network that describes an electrostatic signal driv-
ing a mechanically vibrating system (an electromechanical transducer). The equiva-
lent electric network that models the piezo resonator was later developed by Van Dyke
(1928), Butterworth, and others, and is shown in

Figure 2.6

. In the equivalent net-

work, the circuit parameters R

1

, L

1

, and C

1

represent parameters associated with the

quartz crystal, while C

0

represents the shunt capacitance of the resonator electrodes in

parallel with the container that packages the crystal.

The inductance L

1

is associated with the mass of the quartz slab, and the capacitance

C

1

is associated with the stiffness of the slab. Finally, the resistance, R

1

is determined by

the loss processes associated with the operation of the crystal. Each of the equivalent
parameters can be measured using standard network measurement techniques. Typical
parameters are shown in

Table 2.1

.

Application of an electric field across the electrodes of the crystal results in both

face and thickness shear waves developing within the crystal resonator. For a Y-cut
crystal, face shear modes occur in the X-Z plane, while thickness shear modes develop
in the X-Y plane.

The vibration is enhanced for electric signals near a resonant frequency of the reso-

nator. Low frequency resonances tend to occur in the face shear modes, while high fre-
quency resonances occur in thickness shear modes. In either case, when the applied
electric signal is near a resonant frequency, the energy required to maintain the vibra-
tional motion is small.

Figure 2.6 Equivalent net-
work of a piezoelectric reso-
nator. (

From [1]. Used with

permission.)

© 1999 CRC Press LLC

Temperature Compensation

Compensation for temperature-related frequency shifts can be accomplished by con-
necting a reactance, such as a variable voltage capacitor (a varactor) in series with the
resonator [1]. The effect of this reactance is to pull the frequency to cancel the effect
of frequency drift caused by temperature changes. An example of a frequency versus
temperature characteristic both before and after compensation is shown in

Figure 2.7

.

Several techniques for temperature compensation are possible. They include:

·

Analog temperature compensation

·

Hybrid analog-digital compensation

·

Digital temperature compensation

For analog compensation, a varactor is controlled by a voltage divider network that

contains both thermistors and resistors to manage the compensation. Stability values
are in the range 0.5 to 10 ppm. The lowest value requires that individual components in
the voltage divider network be adjusted under controlled-temperature conditions. This
is difficult over large temperature ranges for analog networks because of the tight com-
ponent value tolerances required and because of interactions between different analog
segments. Analog networks can be segmented in order to increase the independence of
the component adjustments required to improve stability.

These problems can be alleviated through the use of some form of hybrid or digital

compensation. In this scheme, the crystal oscillator is connected to a varactor in series,
as in the analog case. This capacitor provides the coarse compensation (about ±4 parts
in 10

7

). The fine compensation is provided by the digital network to further improve the

stability. The hybrid analog-digital method is then capable of producing stability values
of ±5 parts in 10

8

over the range 40°C to +80°C.

Parameter

200 kHz Fundamental

2 MHz

30 Mhz 3rd
Overtone

90 Mhz 5th
Overtone

R

1

2 kS

100

20

40

L

1

27 H

520 mH

11 mH

6 mH

C

1

0.024 pF

0.012 pF

0.0026 pF

0.0005 pF

C

0

9 pF

4 pF

6 pF

4 pF

Q

18 x 10

3

18 x 10

3

18 x 10

3

18 x 10

3

Table 2.1 Typical Equivalent Circuit Parameter Values for Quartz Crystals (

After [1].)

© 1999 CRC Press LLC

Stress Compensation

Stress compensation is employed in order to stabilize the crystal resonator frequency
against shifts caused by mechanical stresses [1]. These result from temperature
changes or from electrode changes, due to differences in the expansion coefficients,
which can occur over time. The frequency variations are caused by alteration of the
crystal elastic coefficients with stress bias. The method developed for stress compen-
sation is a special crystal cut called the stress-compensated or SC-cut. This cut was
developed to obtain a resonator whose frequency would be independent of specified
stresses.

The SC-cut has some advantages over the more common AT- and BT-cuts. These in-

clude a flatter frequency-temperature response, especially at the turning points, and
less susceptibility to aging because of changes in the mounting stresses. The lower sen-
sitivity to frequency changes means that SC-cut resonators can be frequency-tempera-
ture tested more rapidly than their AT- and BT-cut counterparts. A disadvantage of the
SC-cut is that it is sensitive to air-damping and must be operated in a vacuum environ-
ment to obtain optimum Q factors. The frequency-temperature curve is also roughly
70° higher for the SC-cut over the more common AT-cut, which could also be a disad-
vantage in some applications.

Aging Effects

Aging in crystal oscillators generally refers to any change over time that will affect
the frequency characteristics of the oscillator, or the physical parameters that describe

Figure 2.7

Frequency vs. temperature characteristics for a typical tempera-

ture-compensated crystal oscillator. (

After [2].)

© 1999 CRC Press LLC

the device; such as motional time constant and equivalent circuit parameters [1].
Some factors that influence aging include the following:

·

Surface deterioration

·

Surface contamination

·

Electrode composition

·

Environmental conditions

Deterioration on the surface contributes to strain in the crystal, and this type of strain

can impact the operating frequency of the oscillator. More importantly, the surface
strain is sensitive to humidity and temperature early in the life of the crystal. One
method of combating this surface degradation mechanism is to etch away the strained
surface layers. The initial slope of the frequency change versus etch time curve is expo-
nential. However, after the strained surface layers have been removed, the slope be-
comes linear.

Another surface issue is contamination from foreign materials. The presence of for-

eign substances will load the oscillator with an additional mass, and this can be re-
flected in the equivalent circuit parameters. The contaminants can be introduced during
fabrication processes, from out-gassing of other materials after the device has been
packaged, or from contaminants in the environment during storage. The contamination
process is often increased in higher temperature environments, and temperature cy-
cling will lead to stability problems for the device.

Aging problems associated with the device electrodes includes delamination, lead-

ing to increased resistance or corrosion effects. Typical electrode materials include
gold, silver, and aluminum. Of these, silver is the most popular material.

2.1.4 Operating Class

Amplifier stage operating efficiency is a key element in the design and application of
an electron tube device or system. As the power level of an RF generator increases,
the overall efficiency of the system becomes more important. Increased efficiency
translates into lower operating costs and, usually, improved reliability of the system.
Put another way, for a given device dissipation, greater operating efficiency translates
into higher power output. The operating mode of the final stage, or stages, is the pri-
mary determining element in the maximum possible efficiency of the system.

An electron amplifying device is classified by its individual class of operation. Four

primary class divisions apply to vacuum tube devices:

·

Class A--A mode wherein the power amplifying device is operated over its linear
transfer characteristic. This mode provides the lowest waveform distortion, but
also the lowest efficiency. The basic operating efficiency of a class A stage is 50
percent. Class A amplifiers exhibit low intermodulation distortion, making them
well suited to linear RF amplifier applications.

© 1999 CRC Press LLC

·

Class B--A mode wherein the power amplifying device is operated just outside
its linear transfer characteristic. This mode provides improved efficiency at the
expense of some waveform distortion. Class AB is a variation on class B opera-
tion. The transfer characteristic for an amplifying device operating in this mode
is, predictably, between class A and class B.

·

Class C--A mode wherein the power amplifying device is operated significantly
outside its linear transfer characteristic, resulting in a pulsed output waveform.
High efficiency (up to 90 percent) can be realized with class C operation, but sig-
nificant distortion of the waveform will occur. Class C is used extensively as an
efficient RF power generator.

·

Class D--A mode that essentially results in a switched device state. The power
amplifying device is either on or off. This is the most efficient mode of operation.
It is also the mode that produces the greatest waveform distortion.

The angle of current flow determines the class of operation for a power amplifying

device. Typically, the following generalizations regarding conduction angle apply:

·

Class A: 360°

·

Class AB: between 180 and 360°

·

Class B: 180°

·

Class C: less than 180°

Subscripts also may be used to denote grid current flow. The subscript "1" means that
no grid current flows in the stage; the subscript "2" denotes grid current flow.

Figure

2.8

charts operating efficiency as a function of conduction angle for an RF amplifier.

The class of operation is not directly related to the type of amplifying circuit. Vac-

uum tube stages may be grid- or cathode-driven without regard to the operating class.

Operating Efficiency

The design goal of all RF amplifiers is to convert input power into an RF signal at the
greatest possible efficiency. Direct current input power that is not converted into a
useful output signal is, for the most part, converted to heat. This heat is wasted energy,
which must be removed from the amplifying device. Removal of heat is a problem
common to all high-power RF amplifiers. Cooling methods include:

·

Natural convection

·

Radiation

·

Forced convection

·

Liquid

·

Conduction

·

Evaporation

© 1999 CRC Press LLC

The type of cooling method chosen is dictated in large part by the type of active de-

vice used and the power level involved. For example, liquid cooling is used almost ex-
clusively for high-power (100 kW) vacuum tubes; conduction is used most often for
low-power (10 kW) devices. (Power-device cooling techniques are discussed in detail
in Section 8.2.)

2.1.5 Broadband Amplifier Design

RF design engineers face a continuing challenge to provide adequate bandwidth for
the signals to be transmitted while preserving as much efficiency from the overall sys-
tem as possible. These two parameters, although not mutually exclusive, often involve
tradeoffs for both designers and operators.

An ideal RF amplifier will operate over a wide band of frequencies with minimum

variations in output power, phase, distortion, and efficiency. The bandwidth of the am-
plifier depends to a great extent on the type of active device used, the frequency range
required, and the operating power. As a general rule, a bandwidth of 20 percent or
greater at frequencies above 100 MHz can be considered broadband. Below 100 MHz,
broadband amplifiers typically have a bandwidth of one octave or more.

Stagger Tuning

Several stages with narrowband response (relative to the desired system bandwidth)
can be cascaded and, through the use of stagger tuning, made broadband. Although
there is an efficiency penalty for this approach, it has been used for years in all types
of equipment. The concept is simple: Offset the center operating frequencies (and,

Figure 2.8 Plate efficiency as a function of conduction angle for an amplifier with a
tuned load.

© 1999 CRC Press LLC

therefore, peak amplitude response) of the cascaded amplifiers so that the resulting
passband is flat and broad.

For example, the first stage in a three-stage amplifier is adjusted for peak response at

the center operating frequency of the system. The second stage is adjusted above the
center frequency, and the third stage is adjusted below center. The resulting composite
response curve yields a broadband trace. The efficiency penalty for this scheme varies
depending on the power level of each stage, the amount of stagger tuning required to
achieve the desired bandwidth, and the number of individual stages.

Matching Circuits

The individual stages of an RF generator must be coupled together. Rarely do the out-
put impedance and power level of one stage precisely match the input impedance and
signal-handling level of the next stage. There is a requirement, therefore, for broad-
band matching circuits. Matching at radio frequencies can be accomplished with sev-
eral different techniques, including:

·

A 1/4-wave transformer: A matching technique using simply a length of trans-
mission line 1/4-wave long, with a characteristic impedance of
Z

Z

Z

line

in

out

=

×

, where Z

in

and Z

out

= the terminating impedances. Multiple

1/4-wave transformers can be cascaded to achieve more favorable matching
characteristics. Cascaded transformers permit small matching ratios for each in-
dividual section.

·

A balun transformer: A transmission-line transformer in which the turns are ar-
ranged physically to include the interwinding capacitance as a component of the
characteristic impedance of the transmission line. This technique permits wide
bandwidths to be achieved in the device without unwanted resonances.

Balun transformers usually are made of twisted wire pairs or twisted coaxial lines.

Ferrite toroids may be used as the core material.

Power Combining

The two most common methods of extending the operating power of active devices
are direct paralleling of components and hybrid splitting/combining. Direct parallel-
ing has been used for tube designs, but application of this simple approach is limited
by variations in device operating parameters. Two identical devices in parallel do not
necessarily draw the same amount of current (supply the same amount of power to the
load). Paralleling at UHF and above can be difficult because of the restrictions of op-
erating wavelength.

The preferred approach involves the use of identical stages driven in parallel from a

hybrid coupler. The coupler provides a constant-source impedance and directs any re-
flected energy from the driven stages to a reject port for dissipation. A hybrid coupler
offers a standing-wave-canceling effect that improves system performance. Hybrids
also provide a high degree of isolation between active devices in a system.

© 1999 CRC Press LLC

2.1.6 Thermal and Circuit Noise

All electronic circuits are affected by any number of factors that cause their perfor-
mance to be degraded from the ideal assumed in simple component models, in ways
that can be controlled but not eliminated entirely [4]. One limitation is the failure of
the model to account properly for the real behavior of components, either the result of
an oversimplified model description or variations in manufacture. Usually, by careful
design, the engineer can work around the limitations of the model and produce a de-
vice or circuit whose operation is very close to predictions. One source of perfor-
mance degradation that cannot be easily overcome, however, is noise.

When vacuum tubes first came into use in the early part of the twentieth century,

they were extensively used in radios as signal amplifiers and for other related signal
conditioning functions. Thus, the measure of performance that was of greatest impor-
tance to circuit designers was the quality of the sound produced at the radio speaker. It
was immediately noticed that the sound coming from speakers not only consisted of the
transmitted signal but also of popping, crackling, and hissing sounds, which seemed to
have no pattern and were distinctly different from the sounds that resulted from other
interfering signal sources, such as other radio stations using neighboring frequencies.
This patternless or random signal was labeled "noise," and has become the standard
term for signals that are random and are combined with the circuit signal to affect the
overall performance of the system.

As the study of noise has progressed, engineers have come to realize that there are

many sources of noise in circuits. The following definitions are commonly used in dis-
cussions of circuit noise:

·

White noise: Noise whose energy is evenly distributed over the entire frequency
spectrum, within the frequency range of interest (typically below frequencies in
the infrared range). Because noise is totally random it may seem inappropriate to
refer to its frequency range, because it is not really periodic in the ordinary sense.
Nevertheless, by examining an oscilloscope trace of white noise, one can verify
that every trace is different--as the noise never repeats itself--and yet each trace
looks the same. Similarly, a TV set tuned to an unused frequency displays
never-ending "snow" that always looks the same, yet clearly is always changing.
There is a strong theoretical foundation to represent the frequency content of such
signals as covering the frequency spectrum evenly. In this way the impact on
other periodic signals can be analyzed. The term "white noise" arises from the
fact that--similar to white light, which has equal amounts of all light frequen-
cies--white noise has equal amounts of noise at all frequencies within circuit op-
erating range.

·

Interference: The name given to any predictable, periodic signal that occurs in an
electronic circuit in addition to the signal the circuit is designed to process. This
component is distinguished from a noise signal by the fact that it occupies a rela-
tively small frequency range, and because it is predictable, it can often be filtered
out. Interference usually results from another electronic system, such as an inter-
fering radio source.

© 1999 CRC Press LLC

·

Thermal noise: Any temperature-dependent noise generated within a circuit.
This signal usually is the result of the influence of temperature directly on the op-
erating characteristics of circuit elements, which--because of the random motion
of molecules as a result of temperature--in turn creates a random fluctuation of
the signal being processed.

·

Shot noise: A type of circuit noise that is not temperature-dependent, and is not
white noise in the sense that it tends to diminish at higher frequencies. (See

Figure

2.9

.) This noise usually occurs in components whose operation depends on a

mean particle residence time for the active electrons within the device. The cutoff
frequency above which noise disappears is closely related to the inverse of this
characteristic particle residence time. It is called "shot noise" because in a radio it
can make a sound similar to buckshot hitting a drum, as opposed to white noise,
which tends to sound more like hissing (because of the higher frequency content).

Thermal Noise

By definition, thermal noise is internally generated noise that is temperature-depend-
ent [4]. While first observed in vacuum tube devices (because their amplifying capa-
bilities tend to bring out thermal noise), it is also observed in semiconductor devices.
It is a phenomenon resulting from the ohmic resistance of devices that dissipate the
energy lost in them as heat. Heat consists of random motion of molecules, which are
more energetic as temperature increases. Because the motion is random, it is to be ex-
pected that as electrons pass through the ohmic device incurring resistance, there
should be some random deviations in the rate of energy loss. This fluctuation has the
effect of causing variation in the resulting current, which is noise. As the temperature
of the device increases, the random motion of the molecules increases, and so does
the corresponding noise level.

Figure 2.9 A plot of signal
power vs frequency for white
noise and frequency limited
noise. (

From [4]. Used with

permission.)

© 1999 CRC Press LLC

Noise in Systems of Cascaded Stages

The noise level produced by thermal noise sources is not necessarily large, however
because source signal power may also be low, it is usually necessary to amplify the
source signal. Because noise is combined with the source signal, and both are then
amplified, with more noise added at each successive stage of amplification, noise can
become a noticeable phenomenon [4].

2.2

Amplitude Modulation

In the simplest form of amplitude modulation, an analog carrier is controlled by an
analog modulating signal. The desired result is an RF waveform whose amplitude is
varied by the magnitude of the applied modulating signal and at a rate equal to the fre-
quency of the applied signal. The resulting waveform consists of a carrier wave plus
two additional signals:

·

An upper-sideband signal, which is equal in frequency to the carrier plus the fre-
quency of the modulating signal

·

A lower-sideband signal, which is equal in frequency to the carrier minus the fre-
quency of the modulating signal

This type of modulation system is referred to as double-sideband amplitude modula-
tion (DSAM).

The radio carrier wave signal onto which the analog amplitude variations are to be

impressed is expressed as:

( )

( )

e t

A E

t

c

c

=

cos

(2.4)

Where:
e(t) = instantaneous amplitude of carrier wave as a function of time (t)
A = a factor of amplitude modulation of the carrier wave

c

= angular frequency of carrier wave (radians per second)

E

c

= peak amplitude of carrier wave

If A is a constant, the peak amplitude of the carrier wave is constant, and no modula-

tion exists. Periodic modulation of the carrier wave results if the amplitude of A is
caused to vary with respect to time, as in the case of a sinusoidal wave:

( )

( )

A

t

E

E

m

m

c

= +

1

cos

(2.5)

Where:
E

m

/E

c

= the ratio of modulation amplitude to carrier amplitude

The foregoing relationship leads to:

( )

( )

( ) ( )

[

]

e t

E

t

t

c

E

E

m

c

m

c

=

+

1

cos

cos

(2.6)

© 1999 CRC Press LLC

This is the basic equation for periodic (sinusoidal) amplitude modulation. When all
multiplications and a simple trigonometric identity are performed, the result is:

( )

( ) ( ) (

) ( ) (

)

e t

E

t

t

t

t

t

c

c

M

c

m

M

c

m

=

+

+

+

-

cos

cos

cos

2

2

(2.7)

Where:
M = the amplitude modulation factor (E

m

/E

c

)

Amplitude modulation is, essentially, a multiplication process in which the time

functions that describe the modulating signal and the carrier are multiplied to produce a
modulated wave containing intelligence (information or data of some kind). The fre-
quency components of the modulating signal are translated in this process to occupy a
different position in the spectrum.

The bandwidth of an AM transmission is determined by the modulating frequency.

The bandwidth required for full-fidelity reproduction in a receiver is equal to twice the
applied modulating frequency.

The magnitude of the upper sideband and lower sideband will not normally exceed

50 percent of the carrier amplitude during modulation. This results in an upper-side-
band power of one-fourth the carrier power. The same power exists in the lower side-
band. As a result, up to one-half of the actual carrier power appears additionally in the
sum of the sidebands of the modulated signal. A representation of the AM carrier and
its sidebands is shown in

Figure 2.10

. The actual occupied bandwidth, assuming pure

sinusoidal modulating signals and no distortion during the modulation process, is equal
to twice the frequency of the modulating signal.

The extent of the amplitude variations in a modulated wave is expressed in terms of

the degree of modulation or percentage of modulation. For sinusoidal variation, the de-
gree of modulation m is determined from the following:

Figure 2.10 Frequency-domain representation of an amplitude-modulated signal at
100 percent modulation.

E

c

= carrier power,

F

c

= frequency of the carrier, and

F

m

= fre-

quency of the modulating signal.

© 1999 CRC Press LLC

m

E

E

E

avg

avg

=

-

min

(2.8)

Where:
E

avg

= average envelope amplitude

E

min

= minimum envelope amplitude

Full (100 percent) modulation occurs when the peak value of the modulated envelope
reaches twice the value of the unmodulated carrier, and the minimum value of the en-
velope is zero. The envelope of a modulated AM signal in the time domain is shown
in

Figure 2.11

.

When the envelope variation is not sinusoidal, it is necessary to define the degree of

modulation separately for the peaks and troughs of the envelope:

m

E

E

E

pp

avg

avg

=

-

×

max

100

(2.9)

m

E

E

E

np

avg

avg

=

-

×

min

100

(2.10)

Where:
m

pp

= positive peak modulation (percent)

E

max

= peak value of modulation envelope

m

np

= negative peak modulation (percent)

Figure 2.11 Time-domain representation of an amplitude-modulated signal. Modula-
tion at 100 percent is defined as the point at which the peak of the waveform reaches
twice the carrier level, and the minimum point of the waveform is zero.

© 1999 CRC Press LLC

E

avg

= average envelope amplitude

E

min

= minimum envelope amplitude

When modulation exceeds 100 percent on the negative swing of the carrier, spurious

signals are emitted. It is possible to modulate an AM carrier asymmetrically; that is, to
restrict modulation in the negative direction to 100 percent, but to allow modulation in
the positive direction to exceed 100 percent without a significant loss of fidelity. In
fact, many modulating signals normally exhibit asymmetry, most notably human
speech waveforms.

The carrier wave represents the average amplitude of the envelope and, because it is

the same regardless of the presence or absence of modulation, the carrier transmits no
information. The information is carried by the sideband frequencies. The amplitude of
the modulated envelope may be expressed as follows [5]:

(

)

(

)

E

E

E

f t

E

f t

=

+

+

+

+

0

1

1

1

2

2

2

2

2

sin

sin

(2.11)

Where:
E = envelope amplitude
E

0

= carrier wave crest value (volts)

E

1

= 2

× first sideband crest amplitude (volts)

f

1

= frequency difference between the carrier and the first upper/lower sidebands

E

2

= 2

× second sideband crest amplitude (volts)

f

2

= frequency difference between the carrier and the second upper/lower sidebands

1

= phase of the first sideband component

2

= phase of the second sideband component

2.2.1 High-Level AM Modulation

High-level anode modulation is the oldest and simplest method of generating a
high-power AM signal. In this system, the modulating signal is amplified and com-
bined with the dc supply source to the anode of the final RF amplifier stage. The RF
amplifier is normally operated class C. The final stage of the modulator usually con-
sists of a pair of tubes operating class B in a push-pull configuration. A basic
high-level modulator is shown in

Figure 2.12

.

The RF signal normally is generated in a low-level transistorized oscillator. It is then

amplified by one or more solid-state or vacuum tube stages to provide final RF drive at
the appropriate frequency to the grid of the final class C amplifier. The modulation in-
put is applied to an intermediate power amplifier (usually solid-state) and used to drive
two class B (or class AB) push-pull output devices. The final amplifiers provide the
necessary modulating power to drive the final RF stage. For 100 percent modulation,
this modulating power is equal to 50 percent of the actual carrier power.

The modulation transformer shown in

Figure 2.12

does not usually carry the dc sup-

ply current for the final RF amplifier. The modulation reactor and capacitor shown pro-
vide a means to combine the signal voltage from the modulator with the dc supply to the
final RF amplifier. This arrangement eliminates the necessity of having direct current

© 1999 CRC Press LLC

flow through the secondary of the modulation transformer, which would result in mag-
netic losses and saturation effects. In some transmitter designs, the modulation reactor
is eliminated from the system, thanks to improvements in transformer technology.

The RF amplifier normally operates class C with grid current drawn during positive

peaks of the cycle. Typical stage efficiency is 75 to 83 percent. An RF tank following
the amplifier resonates the output signal at the operating frequency and, with the assis-
tance of a low-pass filter, eliminates harmonics of the amplifier caused by class C oper-
ation.

This type of system was popular in AM applications for many years, primarily be-

cause of its simplicity. The primary drawback is low overall system efficiency. The
class B modulator tubes cannot operate with greater than 50 percent efficiency. Still,
with inexpensive electricity, this was not considered to be a significant problem. As en-
ergy costs increased, however, more efficient methods of generating high-power AM
signals were developed. Increased efficiency normally came at the expense of added
technical complexity.

2.2.2 Vestigial-Sideband Amplitude Modulation

Because the intelligence (modulating signal) of conventional AM transmission is
identical in the upper and lower sidebands, it is possible to eliminate one sideband and

Figure 2.12 Simplified diagram of a high-level amplitude-modulated amplifier.

© 1999 CRC Press LLC

still convey the required information. This scheme is implemented in vestigial-side-
band AM (VSBAM). Complete elimination of one sideband (for example, the lower
sideband) requires an ideal high-pass filter with infinitely sharp cutoff. Such a filter is
difficult to implement in any practical design. VSBAM is a compromise technique
wherein one sideband (typically the lower sideband) is attenuated significantly. The
result is a savings in occupied bandwidth and transmitter power.

VSBAM is used for television broadcast transmission and other applications. A typ-

ical bandwidth trace for a VSBAM TV transmitter is shown in

Figure 2.13

.

2.2.3 Single-Sideband Amplitude Modulation

The carrier in an AM signal does not convey any intelligence. All of the modulating
information is in the sidebands. It is possible, therefore, to suppress the carrier upon
transmission, radiating only one or both sidebands of the AM signal. The result is
much greater efficiency at the transmitter (that is, a reduction in the required transmit-
ter power). Suppression of the carrier may be accomplished with DSAM and SSBAM
signals. Single-sideband suppressed carrier AM (SSB-SC) is the most spectrum- and
energy-efficient mode of AM transmission.

Figure 2.14

shows representative wave-

forms for suppressed carrier transmissions.

A waveform with carrier suppression differs from a modulated wave containing a

carrier primarily in that the envelope varies at twice the modulating frequency. In addi-
tion, it will be noted that the SSB-SC wave has an apparent phase that reverses every
time the modulating signal passes through zero. The wave representing a single side-
band consists of a number of frequency components, one for each component in the

Figure 2.13 Idealized amplitude characteristics of the FCC standard waveform for
monochrome and color TV transmission. (

Adapted from: FCC Rules, Sec. 73.699.)

© 1999 CRC Press LLC

Figure 2.14 Types of suppressed carrier amplitude modulation: (

a) the modulating sig-

nal, (

b) double-sideband AM signal, (c) double-sideband suppressed carrier AM, (d)

single-sideband suppressed carrier AM.

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

original signal. Each of these components has an amplitude proportional to the ampli-
tude of the corresponding modulating component and a frequency differing from that
of the carrier by the modulating frequency. The result is that, in general, the envelope
amplitude of the single sideband signal increases with the degree of modulation, and
the envelope varies in amplitude in accordance with the difference frequencies formed
by the various frequency components of the single sideband interacting with each
other.

An SSB-SC system is capable of transmitting a given intelligence within a fre-

quency band only half as wide as that required by a DSAM waveform. Furthermore, the
SSB system saves more than two-thirds of the transmission power because of the elimi-
nation of one sideband and the carrier.

The drawback to suppressed carrier systems is the requirement for a more compli-

cated receiver. The carrier must be regenerated at the receiver to permit demodulation
of the signal. Also, in the case of SSBAM transmitters, it is usually necessary to gener-
ate the SSB signal in a low-power stage and then amplify the signal with a linear power
amplifier to drive the antenna. Linear amplifiers generally exhibit low efficiency.

2.2.4 Quadrature Amplitude Modulation (QAM)

Single sideband transmission makes very efficient use of the spectrum; for example,
two SSB signals can be transmitted within the bandwidth normally required for a sin-
gle DSB signal. However, DSB signals can achieve the same efficiency by means of
quadrature amplitude modulation (QAM), which permits two DSB signals to be
transmitted and received simultaneously using the same carrier frequency [6]. A basic
QAM DSB modulator is shown schematically in

Figure 2.15

.

Two DSB signals coexist separately within the same bandwidth by virtue of the 90°

phase shift between them. The signals are, thus, said to be in quadrature. Demodulation
uses two local oscillator signals that are also in quadrature, i.e., a sine and a cosine sig-
nal, as illustrated in

Figure 2.16

.

Figure 2.15

Simplified

QAM modulator. (

From

[6].

Used with permis-

sion.)

© 1999 CRC Press LLC

The chief disadvantage of QAM is the need for a coherent local oscillator at the re-

ceiver exactly in phase with the transmitter oscillator signal. Slight errors in phase or
frequency can cause both loss of signal and interference between the two signals
(cochannel interference or crosstalk).

The relative merits of the various AM systems are summarized in

Table 2.2

2.3

Frequency Modulation

Frequency modulation is a technique whereby the phase angle or phase shift of a car-
rier is varied by an applied modulating signal. The magnitude of frequency change of
the carrier is a direct function of the magnitude of the modulating signal. The rate at
which the frequency of the carrier is changed is a direct function of the frequency of
the modulating signal. In FM modulation, multiple pairs of sidebands are produced.
The actual number of sidebands that make up the modulated wave is determined by
the modulation index (MI) of the system.

2.3.1 Modulation Index

The modulation index is a function of the frequency deviation of the system and the
applied modulating signal:

MI

F

M

d

f

=

(2.12)

Where:
MI = the modulation index
F

d

= frequency deviation

M

f

= modulating frequency

Figure 2.16

Simplified

QAM demodulator. (

From

[6].

Used with permis-

sion.)

© 1999 CRC Press LLC

The higher the MI, the more sidebands produced. It follows that the higher the modu-
lating frequency for a given deviation, the fewer number of sidebands produced, but
the greater their spacing.

Modulation

Scheme

Advantages

Disadvantages

Comments

DSB SC

Good power effi-
ciency; good
low-frequency re-
sponse.

More difficult to generate
than DSB+C; detection
requires coherent local
oscillator, pilot, or
phase-locked loop; poor
spectrum efficiency.

DSB+C

Easier to generate
than DSB SC, es-
pecially at
high-power levels;
inexpensive receiv-
ers using envelope
detection.

Poor power efficiency;
poor spectrum efficiency;
poor low-frequency re-
sponse; exhibits thresh-
old effect in noise.

Used in commercial
AM broadcasting.

SSB SC

Excellent spectrum
efficiency.

Complex transmitter de-
sign; complex receiver
design (same as DSB
SC); poor low-frequency
response.

Used in military
communication sys-
tems, and to multi-
plex multiple phone
calls onto long-haul
microwave links.

SSB+C

Good spectrum effi-
ciency; low receiver
complexity.

Poor power efficiency;
complex transmitters;
poor low-frequency re-
sponse; poor noise per-
formance.

VSB SC

Good spectrum effi-
ciency; excellent
low-frequency re-
sponse; transmitter
easier to build than
for SSB.

Complex receivers (same
as DSB SC).

VSB+C

Good spectrum effi-
ciency; good
low-frequency re-
sponse; inexpen-
sive receivers using
envelope detection.

Poor power efficiency;
poor performance in
noise.

Used in commercial
TV broadcasting.

QAM

Good
low-frequency re-
sponse; good spec-
trum efficiency.

Complex receivers; sensi-
tive to frequency and
phase errors.

Two SSB signals
may be preferable.

Table 2.2 Comparison of Amplitude Modulation Techniques (

After [6].)

© 1999 CRC Press LLC

To determine the frequency spectrum of a transmitted FM waveform, it is necessary

to compute a Fourier series or Fourier expansion to show the actual signal components
involved. This work is difficult for a waveform of this type, because the integrals that
must be performed in the Fourier expansion or Fourier series are not easily solved. The
result, however, is that the integral produces a particular class of solution that is identi-
fied as the Bessel function, illustrated in

Figure 2.17

.

The carrier amplitude and phase, plus the sidebands, can be expressed mathemati-

cally by making the modulation index the argument of a simplified Bessel function.
The general expression is given from the following equations:

RF output voltage = E

E

S

S

S

S

S

S

S

S

c

u

l

u

l

u

l

nu

nl

1

1

1

2

2

3

3

=

+

-

+

-

+

-

+

-

Carrier amplitude =

( )

( )

[

]

E

A J

M

c t

c

=

0

sin

First-order upper sideband =

( ) (

)

S

J

M

c

m t

u

1

1

=

+

sin

First-order lower sideband =

( ) (

)

S

J

M

c

m t

l

1

1

=

-

sin

Second-order upper sideband =

( ) (

)

S

J

M

c

m t

u

2

2

2

=

+

sin

Figure 2.17 Plot of Bessel functions of the first kind as a function of modulation index.

© 1999 CRC Press LLC

Second-order lower sideband =

( ) (

)

S

J

M

c

m t

l

2

2

2

=

-

sin

Third-order upper sideband =

( ) (

)

S

J

M

c

m t

u

3

3

3

=

+

sin

Third-order lower sideband =

( ) (

)

S

J

M

c

m t

l

3

3

3

=

-

sin

Nth-order upper sideband =

( ) (

)

S

J

M

c

n m t

nu

n

=

+

sin

Nth-order lower sideband =

( ) (

)

S

J

M

c

n m t

nl

n

=

-

sin

Where:
A = the unmodulated carrier amplitude constant
J

0

= modulated carrier amplitude

J

1

, J

2

, J

3

...J

n

= amplitudes of the nth-order sidebands

M = modulation index

c

F

c

= 2

, the carrier frequency

m

F

m

= 2

, the modulating frequency

Further supporting mathematics will show that an FM signal using the modulation in-
dices that occur in a wideband system will have a multitude of sidebands. From the
purist point of view, all sidebands would have to be transmitted, received, and demod-
ulated to reconstruct the modulating signal with complete accuracy. In practice, how-
ever, the channel bandwidths permitted FM systems usually are sufficient to recon-
struct the modulating signal with little discernible loss in fidelity, or at least an ac-
ceptable loss in fidelity.

Figure 2.18

illustrates the frequency components present for a modulation index of

5.

Figure 2.19

shows the components for an index of 15. Note that the number of signif-

icant sideband components becomes quite large with a high MI. This simple represen-
tation of a single-tone frequency-modulated spectrum is useful for understanding the
general nature of FM, and for making tests and measurements. When typical modula-
tion signals are applied, however, many more sideband components are generated.
These components vary to the extent that sideband energy becomes distributed over the
entire occupied bandwidth, rather than appearing at discrete frequencies.

Although complex modulation of an FM carrier greatly increases the number of fre-

quency components present in the frequency-modulated wave, it does not, in general,
widen the frequency band occupied by the energy of the wave. To a first approximation,
this band is still roughly twice the sum of the maximum frequency deviation at the peak
of the modulation cycle plus the highest modulating frequency involved.

FM is not a simple frequency translation, as with AM, but involves the generation of

entirely new frequency components. In general, the new spectrum is much wider than
the original modulating signal. This greater bandwidth may be used to improve the sig-
nal-to-noise ratio (S/N) of the transmission system. FM thereby makes it possible to
exchange bandwidth for S/N enhancement.

The power in an FM system is constant throughout the modulation process. The out-

put power is increased in the amplitude modulation system by the modulation process,

© 1999 CRC Press LLC

but the FM system simply distributes the power throughout the various frequency com-
ponents that are produced by modulation. During modulation, a wideband FM system
does not have a high amount of energy present in the carrier. Most of the energy will be
found in the sum of the sidebands.

The constant-amplitude characteristic of FM greatly assists in capitalizing on the

low noise advantage of FM reception. Upon being received and amplified, the FM sig-

Figure 2.18 RF spectrum of a frequency-modulated signal with a modulation index of 5
and other operating parameters as shown.

Figure 2.19 RF spectrum of a frequency-modulated signal with a modulation index of
15 and operating parameters as shown.

© 1999 CRC Press LLC

nal normally is clipped to eliminate all amplitude variations above a certain threshold.
This removes noise picked up by the receiver as a result of man-made or atmospheric
signals. It is not possible (generally speaking) for these random noise sources to change
the frequency of the desired signal; they can affect only its amplitude. The use of hard
limiting in the receiver will strip off such interference.

2.3.2 Phase Modulation

In a phase modulation (PM) system, intelligence is conveyed by varying the phase of
the RF wave. Phase modulation is similar in many respects to frequency modulation,
except in the interpretation of the modulation index. In the case of PM, the modula-
tion index depends only on the amplitude of the modulation; MI is independent of the
frequency of the modulating signal. It is apparent, therefore, that the phase-modulated
wave contains the same sideband components as the FM wave and, if the modulation
indices in the two cases are the same, the relative amplitudes of these different com-
ponents also will be the same.

The modulation parameters of a PM system relate as follows:

f m

f

p

m

=

×

(2.13)

Where:
f = frequency deviation of the carrier
m

p

= phase shift of the carrier

f

m

= modulating frequency

In a phase-modulated wave, the phase shift m

p

is independent of the modulating fre-

quency; the frequency deviation

f is proportional to the modulating frequency. In con-

trast, with a frequency-modulated wave, the frequency deviation is independent of
modulating frequency. Therefore, a frequency-modulated wave can be obtained from a
phase modulator by making the modulating voltage applied to the phase modulator in-
versely proportional to frequency. This can be readily achieved in hardware.

2.3.3 Modifying FM Waves

When a frequency-modulated wave is passed through a harmonic generator, the effect
is to increase the modulation index by a factor equal to the frequency multiplication
involved. Similarly, if the frequency-modulated wave is passed through a frequency
divider, the effect is to reduce the modulation index by the factor of frequency divi-
sion. Thus, the frequency components contained in the wave--and, consequently, the
bandwidth of the wave--will be increased or decreased, respectively, by frequency
multiplication or division. No distortion in the nature of the modulation is introduced
by the frequency change.

When an FM wave is translated in the frequency spectrum by heterodyne action, the

modulation index--hence the relative positions of the sideband frequencies and the
bandwidths occupied by them--remains unchanged.

© 1999 CRC Press LLC

2.3.4 Preemphasis and Deemphasis

The FM transmission/reception system offers significantly better noise-rejection
characteristics than AM. However, FM noise rejection is more favorable at low modu-
lating frequencies than at high frequencies because of the reduction in the number of
sidebands at higher frequencies. To offset this problem, the input signal to the FM
transmitter may be preemphasized to increase the amplitude of higher-frequency sig-
nal components in normal program material. FM receivers utilize complementary
deemphasis to produce flat overall system frequency response.

FM broadcasting, for example, uses a 75

µs preemphasis curve, meaning that the

time constant of the resistance-inductance (RL) or resistance-capacitance (RC) circuit
used to provide the boost of high frequencies is 75

µs. Other values of preemphasis are

used in different types of FM communications systems.

Figure 2.20

shows three com-

mon preemphasis curves.

2.3.5 Modulation Circuits

Early FM transmitters used reactance modulators that operated at a low frequency.
The output of the modulator then was multiplied to reach the desired output fre-
quency. This approach was acceptable for some applications and unsuitable for oth-
ers. Modern FM systems utilize what is referred to as direct modulation; that is, the

Figure 2.20 Preemphasis curves for time constants of 50, 75, and 100

µs.

© 1999 CRC Press LLC

frequency modulation occurs in a modulated oscillator that operates on a center fre-
quency equal to the desired transmitter output frequency.

Various techniques have been developed to generate the direct-FM signal. One of

the most popular uses a variable-capacity diode as the reactive element in the oscillator.
The modulating signal is applied to the diode, which causes the capacitance of the de-
vice to vary as a function of the magnitude of the modulating signal. Variations in the
capacitance cause the frequency of the oscillator to change. The magnitude of the fre-
quency shift is proportional to the amplitude of the modulating signal, and the rate of
frequency shift is equal to the frequency of the modulating signal.

The direct-FM modulator is one element of an FM transmitter exciter, which gener-

ates the composite FM waveform. A block diagram of a complete FM exciter is shown
in

Figure 2.21

. Input signals are buffered, filtered, and preemphasized before being

summed to feed the modulated oscillator. Note that the oscillator is not normally cou-
pled directly to a crystal, but to a free-running oscillator adjusted as closely as possible
to the carrier frequency of the transmitter. The final operating frequency is maintained
carefully by an automatic frequency control system employing a phase locked loop
(PLL) tied to a reference crystal oscillator or frequency synthesizer.

A solid-state class C amplifier typically follows the modulated oscillator and raises

the operating power of the FM signal to 20 to 30 W. One or more subsequent amplifiers
in the transmitter raise the signal power to several hundred watts for application to the
final power amplifier stage. Nearly all high-power FM broadcast transmitters use
solid-state amplifiers up to the final RF stage, which is generally a vacuum tube for op-
erating powers of 15 kW and above. All stages operate in the class C mode. In contrast
to AM systems, each stage in an FM power amplifier can operate class C; no informa-
tion is lost from the frequency-modulated signal because of amplitude changes. As
mentioned previously, FM is a constant-power system.

Figure 2.21 Block diagram of an FM exciter.

© 1999 CRC Press LLC

Direct-FM Modulator

Many types of circuits have been used to produce a direct-FM signal [7]. In each case,
a reactance device is used to shunt capacitive or inductive reactance across an oscilla-
tor. The value of capacitive or inductive reactance is made to vary as the amplitude of
the modulating signal varies. Because the reactive load is placed across an oscillator's
tuned circuit, the frequency of the oscillator will therefore shift by a predetermined
amount, thereby creating an FM signal.

A typical example of a reactance modulator is illustrated in

Figure 2.22

. The circuit

uses a field-effect transistor (FET), where the modulating signal is applied to the modu-
lator through C

1

. The actual components that affect the overall reactance consist of R

1

and C

2

. Typically, the value of C

2

is small as this is the input capacitance to the FET,

which may only be a few picofarads. However, this capacitance will generally be much
larger by a significant amount as a result of the Miller effect. Capacitor C

3

has no signif-

icant effect on the reactance of the modulator; it is strictly a blocking capacitor that
keeps dc from changing the gate bias of the FET.

To further understand the performance of the reactance modulator, the equivalent

circuit of

Figure 2.22

is represented in

Figure 2.23

. The FET is represented as a current

source, gmV

g

, with the internal drain resistance, r

d

. The impedances Z

1

and Z

2

are a com-

bination of resistance and capacitive reactance, which are designed to provide a 90º
phase shift.

Using vector diagrams, we can analyze the phase relationship of the reactance mod-

ulator. Referring to

Figure 2.22

, the resistance of R

1

is typically high compared to the

capacitive reactance of C

2

. The R

1

C

2

circuit is then resistive. Because this circuit is re-

sistive, the current, I

AB

, that flows through it, is in phase with the voltage V

AB

. Voltage V

AB

is also across R

1

C

2

(or Z

1

Z

2

in

Figure 2.23

). This is true because current and voltage tend

to be in phase in a resistive network. However, voltage V

C2

, which is across C

2

, is out of

phase with I

AB

. This is because the voltage that is across a capacitor lags behind its cur-

rent by 90°. (See

Figure 2.24.

)

Figure 2.22 Simplified
reactance modulator.
(

From [7]. Used with

permission.)

© 1999 CRC Press LLC

VCO Direct-FM Modulator

One of the more common direct-FM modulation techniques uses an analog voltage
controlled oscillator (VCO) in a phase locked loop arrangement [7]. In this configu-
ration, shown in

Figure 2.25

, a VCO produces the desired carrier frequency that

is--in turn--modulated by applying the input signal to the VCO input via a variactor
diode. The variactor is used to vary the capacitance of an oscillator tank circuit.
Therefore, the variactor behaves as a variable capacitor whose capacitance changes as
the signal voltage across it changes. As the input capacitance of the VCO is varied by

Figure 2.23 Equiva-
lent circuit of the
reactance modulator.
(

From [7]. Used with

permission.)

Figure 2.24 Vector
d i a g r a m

o f

a

reactance modulator
producing FM. Note:
V

g

=

V

C2

. (

From [7].

Used with permis-
sion.)

Figure 2.25

Volt-

age-controlled di-
rect-FM modulator.
(

From [7]. Used with

permission.)

© 1999 CRC Press LLC

the variactor, the output frequency of the VCO is shifted, which produces a direct-FM
modulated signal.

FM Power Amplifiers

Nearly all high-power FM transmitters employ cavity designs. The 1/4-wavelength
cavity is the most common. The design is simple and straightforward. A number of
variations can be found in different transmitters, but the underlying theory of opera-
tion is the same. The goal of any cavity amplifier is to simulate a resonant tank circuit
at the operating frequency and provide a means to couple the energy in the cavity to
the transmission line.

A typical 1/4-wave cavity is shown in

Figure 2.26

. The plate of the tube connects di-

rectly to the inner section (tube) of the plate-blocking capacitor. The blocking capacitor
can be formed in one of several ways. In at least one design, it is made by wrapping the
outside surface of the inner tube conductor with multiple layers of insulating film. The
exhaust chimney/inner conductor forms the other element of the blocking capacitor.
The cavity walls form the outer conductor of the 1/4-wave transmission line circuit.
The dc plate voltage is applied to the PA tube by a cable routed inside the exhaust chim-

Figure 2.26 Mechanical layout of a common type of 1/4-wave PA cavity for FM service.

© 1999 CRC Press LLC

ney and inner tube conductor. In this design, the screen-contact fingerstock ring
mounts on a metal plate that is insulated from the grounded cavity deck by a blocking
capacitor. This hardware makes up the screen-blocker assembly. The dc screen voltage
feeds to the fingerstock ring from underneath the cavity deck using an insulated
feedthrough.

Some transmitters that employ the 1/4-wave cavity design use a grounded-screen

configuration in which the screen-contact fingerstock ring is connected directly to the
grounded cavity deck. The PA cathode then operates at below ground potential (in other
words, at a negative voltage), establishing the required screen voltage for the tube.

Coarse-tuning of the cavity is accomplished by adjusting the cavity length. The top

of the cavity (the cavity shorting deck) is fastened by screws or clamps and can be raised
or lowered to set the length of the assembly for the particular operating frequency.
Fine-tuning is accomplished by a variable-capacity plate-tuning control built into the
cavity. In the example shown in

Figure 2.26

, one plate of this capacitor--the stationary

plate--is fastened to the inner conductor just above the plate-blocking capacitor. The
movable tuning plate is fastened to the cavity box, the outer conductor, and is linked
mechanically to the front-panel tuning control. This capacity shunts the inner conduc-
tor to the outer conductor and varies the electrical length and resonant frequency of the
cavity.

The theory of operation of cavity amplifier systems is discussed in Section 4.3.

2.4

Pulse Modulation

The growth of digital processing and communications has led to the development of
modulation systems tailor-made for high-speed, spectrum-efficient transmission. In a
pulse modulation system, the unmodulated carrier usually consists of a series of re-
current pulses. Information is conveyed by modulating some parameter of the pulses,
such as amplitude, duration, time of occurrence, or shape. Pulse modulation is based
on the sampling principle, which states that a message waveform with a spectrum of
finite width can be recovered from a set of discrete samples if the sampling rate is
higher than twice the highest sampled frequency (the Nyquist criteria). The samples
of the input signal are used to modulate some characteristic of the carrier pulses.

2.4.1 Digital Modulation Systems

Because of the nature of digital signals (on or off), it follows that the amplitude of the
signal in a pulse modulation system should be one of two heights (present or ab-
sent/positive or negative) for maximum efficiency. Noise immunity is a significant
advantage of such a system. It is necessary for the receiving system to detect only the
presence or absence (or polarity) of each transmitted pulse to allow complete recon-
struction of the original intelligence. The pulse shape and noise level have minimal
effect (to a point). Furthermore, if the waveform is to be transmitted over long dis-
tances, it is possible to regenerate the original signal exactly for retransmission to the
next relay point. This feature is in striking contrast to analog modulation systems in
which each modulation step introduces some amount of noise and signal corruption.

© 1999 CRC Press LLC

In any practical digital data system, some corruption of the intelligence is likely to

occur over a sufficiently large span of time. Data encoding and manipulation schemes
have been developed to detect and correct or conceal such errors. The addition of er-
ror-correction features comes at the expense of increased system overhead and (usu-
ally) slightly lower intelligence throughput.

2.4.2 Pulse Amplitude Modulation

Pulse amplitude modulation (PAM) is one of the simplest forms of data modulation.
PAM departs from conventional modulation systems in that the carrier exists as a se-
ries of pulses, rather than as a continuous waveform. The amplitude of the pulse train
is modified in accordance with the applied modulating signal to convey intelligence,
as illustrated in

Figure 2.27

. There are two primary forms of PAM sampling:

·

Natural sampling (or top sampling), where the modulated pulses follow the am-
plitude variation of the sampled time function during the sampling interval.

·

Instantaneous sampling (or square-topped sampling), where the amplitude of the
pulses is determined by the instantaneous value of the sampled time function cor-
responding to a single instant of the sampling interval. This "single instant" may
be the center or edge of the sampling interval.

There are two common methods of generating a PAM signal:

·

Variation of the amplitude of a pulse sequence about a fixed nonzero value (or
pedestal). This approach constitutes double-sideband amplitude modulation.

·

Double-polarity modulated pulses with no pedestal. This approach constitutes
double-sideband suppressed carrier modulation.

2.4.3 Pulse Time Modulation (PTM)

A number of modulating schemes have been developed to take advantage of the noise
immunity afforded by a constant amplitude modulating system. Pulse time modula-
tion (PTM) is one of those systems. In a PTM system, instantaneous samples of the
intelligence are used to vary the time of occurrence of some parameter of the pulsed
carrier. Subsets of the PTM process include:

·

Pulse duration modulation (PDM), where the time of occurrence of either the
leading or trailing edge of each pulse (or both pulses) is varied from its
unmodulated position by samples of the input modulating waveform. PDM also
may be described as pulse length or pulse width modulation (PWM).

·

Pulse position modulation (PPM), where samples of the modulating input signal
are used to vary the position in time of pulses, relative to the unmodulated wave-
form. Several types of pulse time modulation waveforms are shown in

Figure

2.28

.

© 1999 CRC Press LLC

Figure 2.27

Pulse amplitude modulation waveforms: (

a) modulating signal; (b)

square-topped sampling, bipolar pulse train; (

c) topped sampling, bipolar pulse train; (d)

square-topped sampling, unipolar pulse train; (

e) top sampling, unipolar pulse train.

(

a)

(

b)

(

c)

(

d)

(

e)

© 1999 CRC Press LLC

·

Pulse frequency modulation (PFM), where samples of the input signal are used to
modulate the frequency of a series of carrier pulses. The PFM process is illus-
trated in

Figure 2.29

.

It should be emphasized that all of the pulse modulation systems discussed thus far

may be used with both analog and digital input signals. Conversion is required for either
signal into a form that can be accepted by the pulse modulator.

Figure 2.28

Pulse time modulation waveforms: (

a) modulating signal and sam-

ple-and-hold (S/H) waveforms, (

b) sawtooth waveform added to S/H, (c) leading-edge

PTM, (

d) trailing-edge PTM.

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

2.4.4 Pulse Code Modulation

The pulse modulation systems discussed previously are unencoded systems. Pulse
code modulation (PCM) is a scheme wherein the input signal is quantized into dis-
crete steps and then sampled at regular intervals (as in conventional pulse modula-
tion). In the quantization process, the input signal is sampled to produce a code repre-
senting the instantaneous value of the input within a predetermined range of values.

Figure 2.30

illustrates the concept. Only certain discrete levels are allowed in the

Figure 2.29 Pulse frequency modulation.

Figure 2.30 The quantization process.

© 1999 CRC Press LLC

quantization process. The code is then transmitted over the communications system
as a pattern of pulses.

Quantization inherently introduces an initial error in the amplitude of the samples

taken. This quantization error is reduced as the number of quantization steps is in-
creased. In system design, tradeoffs must be made regarding low quantization error,
hardware complexity, and occupied bandwidth. The greater the number of quantization
steps, the wider the bandwidth required to transmit the intelligence or, in the case of
some signal sources, the slower the intelligence must be transmitted.

In the classic design of a PCM encoder, the quantization steps are equal. The

quantization error (or quantization noise) usually can be reduced, however, through the
use of nonuniform spacing of levels. Smaller quantization steps are provided for
weaker signals, and larger steps are provided near the peak of large signals.
Quantization noise is reduced by providing an encoder that is matched to the level dis-
tribution (probability density) of the input signal.

Nonuniform quantization typically is realized in an encoder through processing of

the input (analog) signal to compress it to match the desired nonuniformity. After com-
pression, the signal is fed to a uniform quantization stage.

2.4.5 Delta Modulation

Delta modulation (DM) is a coding system that measures changes in the direction of
the input waveform, rather than the instantaneous value of the wave itself.

Figure 2.31

illustrates the concept. The clock rate is assumed to be constant. Transmitted pulses
from the pulse generator are positive if the signal is changing in a positive direction;
they are negative if the signal is changing in a negative direction.

As with the PCM encoding system, quantization noise is a parameter of concern for

DM. Quantization noise can be reduced by increasing the sampling frequency (the
pulse generator frequency). The DM system has no fixed maximum (or minimum) sig-
nal amplitude. The limiting factor is the slope of the sampled signal, which must not
change by more than one level or step during each pulse interval.

2.4.6 Digital Coding Systems

A number of methods exist to transmit digital signals over long distances in analog
transmission channels. Some of the more common systems include:

·

Binary on-off keying (BOOK), a method by which a high-frequency sinusoidal
signal is switched on and off corresponding to 1 and 0 (on and off) periods in the
input digital data stream. In practice, the transmitted sinusoidal waveform does
not start or stop abruptly, but follows a predefined ramp up or down.

·

Binary frequency-shift keying (BFSK), a modulation method in which a continu-
ous wave is transmitted that is shifted between two frequencies, representing 1s
and 0s in the input data stream. The BFSK signal may be generated by switching
between two oscillators (set to different operating frequencies) or by applying a
binary baseband signal to the input of a voltage-controlled oscillator (VCO). The

© 1999 CRC Press LLC

transmitted signals often are referred to as a mark (binary digit 1) or a space (bi-
nary digit 0).

Figure 2.32

illustrates the transmitted waveform of a BFSK system.

·

Binary phase-shift keying (BPSK), a modulating method in which the phase of
the transmitted wave is shifted 180° in synchronism with the input digital signal.
The phase of the RF carrier is shifted by

/2 radians or /2 radians, depending

upon whether the data bit is a 0 or a 1.

Figure 2.33

shows the BPSK transmitted

waveform.

Figure 2.31 Delta modulation waveforms: (

a) modulating signal, (b) quantized modu-

lating signal, (

c) pulse train, (d) resulting delta modulation waveform.

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

·

Quadriphase-shift keying (QPSK), a modulation scheme similar to BPSK except
that quaternary modulation is employed, rather than binary modulation. QPSK
requires half the bandwidth of BPSK for the same transmitted data rate.

2.4.7 Baseband Digital Pulse Modulation

After the input samples have been quantized, they are transmitted through a channel,
received, and converted back to their approximate original form [8].The format (mod-
ulation scheme) applied to the quantized samples is determined by a number of fac-
tors, not the least of which is the channel through which the signal passes. A number
of different formats are possible and practical.

Figure 2.32 Binary FSK waveform.

Figure 2.33 Binary PSK waveform.

© 1999 CRC Press LLC

Several common digital modulation formats are shown in

Figure 2.34

. The first (a)

is referred to as non-return-to-zero (NRZ) polar because the waveform does not return
to zero during each signaling interval, but switches from +V to V, or vice versa, at the
end of each signaling interval (NRZ unipolar uses the levels V and 0). On the other
hand, the unipolar return-to-zero (RZ) format, shown in (b) returns to zero in each sig-
naling interval. Because bandwidth is inversely proportional to pulse duration, it is ap-
parent that RZ requires twice the bandwidth that NRZ does. Also, RZ has a nonzero dc
component, whereas NRZ does not necessarily have a nonzero component (unless
there are more 1s than 0s or vice versa). An advantage of RZ over NRZ is that a pulse
transition is guaranteed in each signaling interval, whereas this is not the case for NRZ.
Thus, in cases where there are long strings of 1s or 0s, it may be difficult to synchronize
the receiver to the start and stop times of each pulse in NRZ-based systems. A very im-
portant modulation format from the standpoint of synchronization considerations is
NRZ-mark, also known as differential encoding, where an initial reference bit is chosen
and a subsequent 1 is encoded as a change from the reference and a 0 is encoded as no
change. After the initial reference bit, the current bit serves as a reference for the next
bit, and so on. An example of this modulation format is shown in (c).

Manchester is another baseband data modulation format that guarantees a transition

in each signaling interval and does not have a dc component. Also known as biphase or
split phase, this scheme is illustrated in (d). The format is produced by ORing the data
clock with an NRZ-formatted signal. The result is a + to transition for a logic 1, and a
to + zero crossing for a logic 0.

A number of other data formats have been proposed and employed in the past, but

further discussion is beyond the scope of this chapter.

2.4.8 Spread Spectrum Systems

As the name implies, a spread spectrum system requires a frequency range substan-
tially greater than the basic information-bearing signal. Spread spectrum systems
have some or all of the following properties:

·

Low interference to other communications systems

·

Ability to reject high levels of external interference

·

Immunity to jamming by hostile forces

·

Provision for secure communications paths

·

Operability over multiple RF paths

Spread spectrum systems operate with an entirely different set of requirements than

transmission systems discussed previously. Conventional modulation methods are de-
signed to provide for the easiest possible reception and demodulation of the transmitted
intelligence. The goals of spread spectrum systems, on the other hand, are secure and
reliable communications that cannot be intercepted by unauthorized persons. The most
common modulating and encoding techniques used in spread spectrum communica-
tions include:

© 1999 CRC Press LLC

Figure 2.34 Various baseband modulation formats: (

a) non-return-to zero, (b) unipolar

return-to-zero, (

c) differential encoded (NRZ-mark), (d) split phase. (From [8]. Used with

permission.)

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

·

Frequency hopping, where a random or pseudorandom number (PN) sequence is
used to change the carrier frequency of the transmitter. This approach has two ba-
sic variations: slow frequency hopping, where the hopping rate is smaller than the
data rate, and fast frequency hopping, where the hopping rate is larger than the
data rate. In a fast frequency-hopping system, the transmission of a single piece of
data occupies more than one frequency. Frequency-hopping systems permit mul-
tiple-access capability to a given band of frequencies because each transmitted
signal occupies only a fraction of the total transmitted bandwidth.

·

Time hopping, where a PN sequence is used to switch the position of a mes-
sage-carrying pulse within a series of frames.

·

Message corruption, where a PN sequence is added to the message before modu-
lation.

·

Chirp spread spectrum, where linear frequency modulation of the main carrier is
used to spread the transmitted spectrum. This technique is commonly used in ra-
dar and also has been applied to communications systems.

In a spread spectrum system, the signal power is divided over a large bandwidth. The

signal, therefore, has a small average power in any single narrowband slot. This means
that a spread spectrum system can share a given frequency band with one or more
narrowband systems. Furthermore, because of the low energy in any particular band,
detection or interception of the transmission is difficult.

2.5

References

1. Tate, Jeffrey P., and Patricia F. Mead, "Crystal Oscillators," in The Electronics

Handbook, Jerry C. Whitaker (ed.), CRC Press, Boca Raton, FL, pp. 185199,
1996.

2. Frerking, M. E., Crystal Oscillator Design and Temperature Compensation, Van

Nostrand Reinhold, New York, 1978.

3. Dye, D. W., Proc. Phys. Soc., Vol. 38, pp. 399457, 1926.
4. Douglass, Barry G., "Thermal Noise and Other Circuit Noise," in The Electronics

Handbook, Jerry C. Whitaker (ed.), pp. 3036, 1996.

5. Terman, F. E., Radio Engineering, 3rd ed., McGraw-Hill, New York, pg. 468, 1947.
6. Kubichek, Robert, "Amplitude Modulation," in The Electronics Handbook, Jerry C.

Whitaker (ed.), CRC Press, Boca Raton, FL, pp. 11751187, 1996.

7. Seymour, Ken, "Frequency Modulation," in The Electronics Handbook, Jerry C.

Whitaker (ed.), CRC Press, Boca Raton, FL, pp. 11881200, 1996.

8. Ziemer, Rodger E., "Pulse Modulation," in The Electronics Handbook, Jerry C.

Whitaker (ed.), CRC Press, Boca Raton, FL, pp. 12011212, 1996.

2.6

Bibliography

Benson, K. B., and Jerry. C. Whitaker (eds.), Television Engineering Handbook,

McGraw-Hill, New York, 1986.

© 1999 CRC Press LLC

Benson, K. B., and Jerry. C. Whitaker, Television and Audio Handbook for Techni-

cians and Engineers, McGraw-Hill, New York, 1989.

Crutchfield, E. B. (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, 1991.

Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 3rd Ed.,

McGraw-Hill, New York, 1989.

Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 2nd Ed.,

McGraw-Hill, New York, 1982.

Hulick, Timothy P., "Using Tetrodes for High Power UHF," Proceedings of the Soci-

ety of Broadcast Engineers, Vol. 4, SBE, Indianapolis, IN, pp. 52-57, 1989.

Jordan, Edward C., Reference Data for Engineers: Radio, Electronics, Computer and

Communications, 7th Ed., Howard W. Sams, Indianapolis, IN, 1985.

Laboratory Staff, The Care and Feeding of Power Grid Tubes, Varian Eimac, San

Carlos, CA, 1984.

Mendenhall, G. N., "Fine Tuning FM Final Stages," Broadcast Engineering, Intertec

Publishing, Overland Park, KS, May 1987.

Whitaker, Jerry. C., Maintaining Electronic Systems, CRC Press, Boca Raton, FL,

1992.

Whitaker, Jerry. C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1991.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

3 Vacuum Tube Principles"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

3

Vacuum Tube Principles

3.1

Introduction

A power grid tube is a device using the flow of free electrons in a vacuum to produce
useful work [1]. It has an emitting surface (the cathode), one or more grids that con-
trol the flow of electrons, and an element that collects the electrons (the anode). Power
tubes can be separated into groups according to the number of electrodes (grids) they
contain. The physical shape and location of the grids relative to the plate and cathode
are the main factors that determine the amplification factor (

µ) and other parameters

of the device (see Section 3.2.5). The physical size and types of material used to con-
struct the individual elements determine the power capability of the tube. A wide vari-
ety of tube designs are available to commercial and industrial users. By far the most
common are triodes and tetrodes.

3.2

Characteristics of Electrons

Electrons are minute, negatively charged particles that are constituents of all matter.
They have a mass of 9

× 10

28

g (1/1840 that of a hydrogen atom) and a charge of 1.59

× 10

19

coulomb. Electrons are always identical, irrespective of their source. Atoms

are composed of one or more such electrons associated with a much heavier nucleus,
which has a positive charge equal to the number of the negatively charged electrons
contained in the atom; an atom with a full quota of electrons is electrically neutral.
The differences in chemical elements arise from differences in the nucleus and in the
number of associated electrons.

Free electrons can be produced in a number of ways. Thermonic emission is the

method normally employed in vacuum tubes (discussed in Section 3.2.2). The principle
of thermonic emission states that if a solid body is heated sufficiently, some of the elec-
trons that it contains will escape from the surface into the surrounding space. Electrons
also are ejected from solid materials as a result of the impact of rapidly moving elec-
trons or ions. This phenomenon is referred to as secondary electron emission, because
it is necessary to have a primary source of electrons (or ions) before the secondary
emission can be obtained (Section 3.2.3). Finally, it is possible to pull electrons directly
out of solid substances by an intense electrostatic field at the surface of the material.

© 1999 CRC Press LLC

Positive ions represent atoms or molecules that have lost one or more electrons and

so have become charged bodies having the weight of the atom or molecule concerned,
and a positive charge equal to the negative charge of the lost electrons. Unlike electrons,
positive ions are not all alike and may differ in charge or weight, or both. They are much
heavier than electrons and resemble the molecule or atom from which they are derived.
Ions are designated according to their origin, such as mercury ions or hydrogen ions.

3.2.1 Electron Optics

Electrons and ions are charged particles and, as such, have forces exerted upon them
by an electrostatic field in the same way as other charged bodies. Electrons, being
negatively charged, tend to travel toward the positive or anode electrode, while the
positively charged ions travel in the opposite direction (toward the negative or cathode
electrode). The force F exerted upon a charged particle by an electrostatic field is pro-
portional to the product of the charge e of the particle and the voltage gradient G of
the electrostatic field [1]:

F

G

e

= × ×10

7

(3.1)

Where:
F = force in dynes
G = voltage gradient in volts per centimeter
e = charge in coulombs

This force upon the ion or electron is exerted in the direction of the electrostatic flux
lines at the point where the charge is located. The force acts toward or away from the
positive terminal, depending upon whether a negative or positive charge, respectively,
is involved.

The force that the field exerts on the charged particle causes an acceleration in the

direction of the field at a rate that can be calculated by the laws of mechanics where the
velocity does not approach that of light:

A

F

M

=

(3.2)

Where:
A = acceleration in centimeters per second per second
F = force in dynes
m = mass in grams

The velocity an electron or ion acquires in being acted upon by an electrostatic field

can be expressed in terms of the voltage through which the electron (or ion) has fallen in
acquiring the velocity. For velocities well below the speed of light, the relationship be-
tween velocity and the acceleration voltage is:

© 1999 CRC Press LLC

v

V

e

m

=

× × ×

2

10

7

(3.3)

Where:
v = velocity in centimeters per second corresponding to V
V = accelerating voltage
e = charge in coulombs
m = mass in grams

Electrons and ions move at great velocities in even moderate-strength fields. For ex-

ample, an electron dropping through a potential difference of 2500 V will achieve a ve-
locity of approximately one-tenth the speed of light.

Electron optics, as discussed in this section, relies on the principles of classical

physics. While modern tube design uses computer simulation almost exclusively, the
preceding information is still valid and provides a basis for the understanding of elec-
tron motion within a vacuum tube device.

Magnetic Field Effects

An electron in motion represents an electric current of magnitude ev, where e is the
magnitude of the charge on the electron and v is its velocity. A magnetic field accord-
ingly exerts a force on a moving electron exactly as it exerts a force on an electric cur-
rent in a wire. The magnitude of the force is proportional to the product of the equiva-
lent current ev represented by the moving electron and the strength of the component
of the magnetic field in a direction at right angles to the motion of the electron. The
resulting force is, then, in a direction at right angles both to the direction of motion of
the electron and to the component of the magnetic field that is producing the force. As
a result, an electron entering a magnetic field with a high velocity will follow a
curved path. Because the acceleration of the electron that the force of the magnetic
field produces is always at right angles to the direction in which the electron is travel-
ing, an electron moving in a uniform magnetic field will follow a circular path. The
radius of this circle is determined by the strength of the magnetic field and the speed
of the electron moving through the field.

When an electron is subjected to the simultaneous action of both electric and mag-

netic fields, the resulting force acting on the electron is the vector sum of the force re-
sulting from the electric field and the force resulting from the magnetic field, each con-
sidered separately.

Magnetic fields are not used for conventional power grid tubes. Microwave power

tubes, on the other hand, use magnetic fields to confine and focus the electron stream.

3.2.2 Thermal Emission From Metals

Thermonic emission is the phenomenon of an electric current leaving the surface of a
material as the result of thermal activation. Electrons with sufficient thermal energy
to overcome the surface-potential barrier escape from the surface of the material. This

© 1999 CRC Press LLC

thermally emitted electron current increases with temperature because more electrons
have sufficient energy to leave the material.

The number of electrons released per unit area of an emitting surface is related to the

absolute temperature of the emitting material and a quantity b that is a measure of the
work an electron must perform in escaping through the surface, according to the equa-
tion [1]:

I

A T

b

T

=

-

2

(3.4)

Where:
T = absolute temperature of the emitting material
b = the work an electron must perform in escaping the emitter surface
I = electron current in amperes per square centimeter
A = a constant (value varies with type of emitter)

The exponential term in the equation accounts for most of the variation in emission
with temperature. The temperature at which the electron current becomes appreciable
is accordingly determined almost solely by the quantity b.

Figure 3.1

plots the emis-

sion resulting from a cathode operated at various temperatures.

Thermal electron emission can be increased by applying an electric field to the cath-

ode. This field lowers the surface-potential barrier, enabling more electrons to escape.
This field-assisted emission is known as the Schottky effect.

Figure 3.2

illustrates common heater-cathode structures for power tubes.

3.2.3 Secondary Emission

Almost all metals and some insulators will emit low-energy electrons (secondary
electrons) when bombarded by other energetic electrons. The number of secondary
electrons emitted per primary electron is determined by the velocity of the primary
bombarding electrons and the nature and condition of the material composing the sur-
face being bombarded.

Figure 3.3

illustrates a typical relationship for two types of

surfaces. As shown in the figure, no secondary electrons are produced when the pri-
mary velocity is low. However, with increasing potential (and consequently higher ve-
locity), the ratio of secondary to primary electrons increases, reaching a maximum
and then decreasing. With pure metal surfaces, the maximum ratio of secondary to
primary electrons ranges from less than 1 to approximately 3. Some complex surfaces
based on alkali metal compounds yield ratios of secondary to primary electrons as
high as 5 to 10.

The majority of secondary electrons emitted from a conductive surface have rela-

tively low velocity. However, a few secondary electrons usually are emitted with a ve-
locity nearly equal to the velocity of the bombarding primary electrons.

For insulators, the ratio of secondary to primary electrons as a function of primary

electron potential follows along the same lines as for metals. The net potential of the in-
sulating surface being bombarded is affected by the bombardment. If the ratio of sec-
ondary to primary current is less than unity, the insulator acquires a net negative charge

© 1999 CRC Press LLC

because more electrons arrive than depart. This causes the insulator to become more
negative and, finally, to repel most of the primary electrons, resulting in a blocking ac-
tion. In the opposite extreme, when the ratio of secondary to primary electrons exceeds
unity, the insulating surface loses electrons through secondary emission faster than

Figure 3.1 Variation of electron emission as a function of absolute temperature for a
thoriated-tungsten emitter.

Figure 3.2 Common types of heater and cathode structures. (

From [2]. Used with per-

mission.)

© 1999 CRC Press LLC

they arrive; the surface becomes increasingly positive. This action continues until the
surface is sufficiently positive that the ratio of secondary to primary electrons de-
creases to unity as a result of the increase in the velocity of the bombarding electrons, or
until the surface is sufficiently positive that it attracts back into itself a significant num-
ber of secondary electrons. This process makes the number of electrons gained from all
sources equal to the number of secondary electrons emitted.

3.2.4 Diode

A diode is a two-electrode vacuum tube containing a cathode, which emits electrons
by thermonic emission, surrounded by an anode (or plate). (See

Figure 3.4

) Such a

tube is inherently a rectifier because when the anode is positive, it attracts electrons;
current, therefore, passes through the tube. When the anode is negative, it repels the
electrons and no current flows.

The typical relationship between anode voltage and current flowing to the positive

anode is shown in

Figure 3.5

. When the anode voltage is sufficiently high, electrons are

Figure 3.3 Ratio of secondary emission current to primary current as a function of pri-
mary electron velocity.

© 1999 CRC Press LLC

drawn from the cathode as rapidly as they are emitted. The anode current is then limited
by the electron emission of the cathode and, therefore, depends upon cathode tempera-
ture rather than anode voltage.

At low anode voltages, however, plate current is less than the emission of which the

cathode is capable. This occurs because the number of electrons in transit between the
cathode and plate at any instant cannot exceed the number that will produce a negative
space charge, which completely neutralizes the attraction of the positive plate upon the
electrons just leaving the cathode. All electrons in excess of the number necessary to

Figure 3.5 Anode current as a function of anode voltage in a two-electrode tube for
three cathode temperatures.

(

a)

(

b)

Figure 3.4 Vacuum diode: (

a) directly heated cathode, (b) indirectly heated cathode.

(

From [2]. Used with permission.)

© 1999 CRC Press LLC

neutralize the effects of the plate voltage are repelled into the cathode by the negative
space charge of the electrons in transit; this situation applies irrespective of how many
excess electrons the cathode emits. When the plate current is limited in this way by
space charge, plate current is determined by plate potential and is substantially inde-
pendent of the electron emission of the cathode.

Detailed examination of the space-charge situation will reveal that the negative

charge of the electrons in transit between the cathode and the plate is sufficient to give
the space in the immediate vicinity of the cathode a slight negative potential with re-
spect to the cathode. The electrons emitted from the cathode are projected out into this
field with varying emission velocities. The negative field next to the cathode causes the
emitted electrons to slow as they move away from the cathode, and those having a low
velocity of emission are driven back into the cathode. Only those electrons having the
highest velocities of emission will penetrate the negative field near the cathode and
reach the region where they are drawn toward the positive plate. The remainder (those
electrons having low emission velocities) will be brought to a stop by the negative field
adjacent to the cathode and will fall back into the cathode.

The energy that is delivered to the tube by the source of anode voltage is first ex-

pended in accelerating the electrons traveling from the cathode to the anode; it is con-
verted into kinetic energy. When these swiftly moving electrons strike the anode, this
kinetic energy is then transformed into heat as a result of the impact and appears at the
anode in the form of heat that must be radiated to the walls of the tube.

The basic function of a vacuum tube diode--to rectify an ac voltage--has been su-

perseded by solid-state devices. An understanding of how the diode operates, however,
is important in understanding the operation of triodes, tetrodes, and pentodes.

3.2.5 Triode

The power triode is a three-element device commonly used in a wide variety of RF
generators. Triodes have three internal elements: the cathode, control grid, and plate.
Most tubes are cylindrically symmetrical. The filament or cathode structure, the grid,
and the anode all are cylindrical in shape and are mounted with the axis of each cylin-
der along the center line of the tube, as illustrated in

Figure 3.6

.

The grid normally is operated at a negative potential with respect to the cathode, and

so attracts no electrons. However, the extent to which it is negative affects the electro-
static field in the vicinity of the cathode and, therefore, controls the number of electrons
that pass between the grid and the plate. The grid, in effect, functions as an imperfect
electrostatic shield. It allows some, but not all, of the electrostatic flux from the anode
to leak between its wires. The number of electrons that reach the anode in a triode tube
under space-charge-limited conditions is determined almost solely by the electrostatic
field near the cathode; the field in the rest of the interelectrode space has little effect.
This phenomenon results because the electrons near the cathode are moving slowly
compared with the electrons that have traveled some distance toward the plate. The re-
sult of this condition is that the volume density of electrons in proportion to the rate of
flow is large near the cathode and low in the remainder of the interelectrode space. The
total space charge of the electrons in transit toward the plate, therefore, consists almost

© 1999 CRC Press LLC

solely of the electrons in the immediate vicinity of the cathode. After an electron has
traveled beyond this region, it reaches the plate so quickly as to contribute to the space
charge for only a brief additional time interval. The result is that the space current in a
three-electrode vacuum tube is, for all practical purposes, determined by the electro-
static field that the combined action of the grid and plate potentials produces near the
cathode.

When the grid structure is symmetrical, the field E at the surface of the cathode is

proportional to the quantity:

E

E

c

b

+

µ

(3.5)

Where:
E

c

= control grid voltage (with respect to cathode)

E

b

= anode voltage (with respect to cathode)

µ = a constant determined by the geometry of the tube

The constant

µ, the amplification factor, is independent of the grid and plate voltages.

It is a measure of the relative effectiveness of grid and plate voltages in producing
electrostatic fields at the surfaces of the cathode. Placement of the control grid rela-
tive to the cathode and plate determines the amplification factor. The

µ values of tri-

Figure 3.6 Mechanical configura-
tion of a power triode.

© 1999 CRC Press LLC

odes generally range from 5 to 200. Key mathematical relationships include the fol-
lowing:

µ =

E

E

b

c1

(3.6)

R

E

I

p

b

b

=

(3.7)

S

I

E

m

b

c

=

1

(3.8)

Where:
µ = amplification factor (with plate current held constant)
R

p

= dynamic plate resistance

S

m

= transconductance (also may be denoted G

m

)

E

b

= total instantaneous plate voltage

E

c1

= total instantaneous control grid voltage

I

b

= total instantaneous plate current

The total cathode current of an ideal triode can be determined from the equation:

I

E

E

k

c

b

=

+

µ

3

2

(3.9)

Where:
I

k

= cathode current

K = a constant determined by tube dimensions
E

c

= grid voltage

E

b

= plate voltage

µ = amplification factor

Figure 3.7

plots plate and grid current as a function of plate voltage at various grid

voltages for a triode with a

µof 12. The tube, a 304TL, is a classic design and, while not

used in new equipment, provides a common example of the relationship between the
parameters plotted.

Figure 3.8

plots the same parameters for a tube with a

µ of 20. Ob-

serve how much more plate current at a given plate voltage can be obtained from the
304TL (

µ = 12) without driving the grid into the positive grid region. Note also how

much more bias voltage is required for the 304TL to cut the plate current off at some
given plate voltage. With this increased bias, there is a corresponding increase in grid
voltage swing to drive up the zero grid voltage point on the curve. Low-

µ tubes have

lower voltage gain by definition. This fact can be seen by comparing

Figures 3.7

and

3.8

.

© 1999 CRC Press LLC

Triodes with a

µof 20 to 50 generally are used in conventional RF amplifiers and os-

cillators. High-

µ triodes (200 or more) may be designed so that the operating bias is

zero, as depicted in

Figure 3.9

. These zero-bias triodes are available with plate dissipa-

tion ratings of 400 W to 10 kW or more. The zero-bias triode commonly is used in
grounded-grid amplification. The tube offers good power gain and circuit simplicity.
No bias power source is required. Furthermore, no protection circuits for loss of bias or
drive are needed. Despite these attributes, present-day use of the zero-bias triode is lim-
ited.

Low- and medium-

µ devices usually are preferred for induction heating applica-

tions because of the wide variations in load that an induction or dielectric heating oscil-
lator normally works into. Such tubes exhibit lower grid-current variation with a chang-
ing load. The grid current of a triode with a

µ of 20 will rise substantially less under a

light- or no-load condition than a triode with a

µ of 40. High-µ triode oscillators can be

designed for heating applications, but extra considerations must be given to current rise
under no-load conditions.

Vacuum tubes specifically designed for induction heating are available, intended for

operation under adverse conditions. The grid structure is ruggedized with ample dissi-

Figure 3.7 Constant-current characteristics for a triode with a

µ of 12.

© 1999 CRC Press LLC

Figure 3.8 Constant-current characteristics for a triode with a

µ of 20.

Figure 3.9 Grounded-grid constant-current characteristics for a zero-bias triode with a

µ of 200.

© 1999 CRC Press LLC

pation capability to deal with wide variations in load. As the load decreases, grid dissi-
pation increases.

Triodes also are manufactured with the cathode, grid, and anode in the shape of a flat

surface, as shown in

Figure 3.10

. Tubes so constructed are called planar triodes. This

construction technique permits operation at high frequencies. The close spacing re-
duces electron transit time, allowing the tube to be used at high frequencies (up to 3
GHz or so). The physical construction of planar triodes results in short lead lengths,
which reduces lead inductance. Planar triodes are used in both continuous wave (CW)
and pulsed modes. The contacting surfaces of the planar triode are arranged for easy in-
tegration into coaxial and waveguide resonators.

3.2.6 Tetrode

The tetrode is a four-element tube with two grids. The control grid serves the same
purpose as the grid in a triode, while a second (screen) grid with the same number of
vertical elements (bars) as the control grid is mounted between the control grid and
the anode. The grid bars of the screen grid are mounted directly behind the con-
trol-grid bars, as observed from the cathode surface, and serve as a shield or screen
between the input circuit and the output circuit of the tetrode. The principal advan-
tages of a tetrode over a triode include:

·

Lower internal plate-to-grid feedback.

Figure 3.10 Internal configuration of a planar triode.

© 1999 CRC Press LLC

·

Lower drive power requirements. In most cases, the driving circuit need supply
only 1 percent of the output power.

·

More efficient operation. Tetrodes allow the design of compact, simple, flexible
equipment with little spurious radiation and low intermodulation distortion.

Plate current is almost independent of plate voltage in a tetrode.

Figure 3.11

plots

plate current as a function of plate voltage at a fixed screen voltage and various grid
voltages. In an ideal tetrode, a change in plate current does not cause a change in plate
voltage. The tetrode, therefore, can be considered a constant-current device. The volt-
ages on the screen and control grids determine the amount of plate current.

The total cathode current of an ideal tetrode is determined by the equation:

I

K

E

E

E

k

c

c

s

b

p

=

+

+

1

2

3

2

µ

µ

(3.10)

Where:
I

k

= cathode current

K = a constant determined by tube dimensions
E

c1

= control grid voltage

E

c2

= screen grid voltage

µ

s

= screen amplification factor

µ

p

= plate amplification factor

E

b

= plate voltage

Figure 3.11 Tetrode plate current characteristics. Plate current is plotted as a function
of plate voltage, with grid voltages as shown.

© 1999 CRC Press LLC

The arithmetic value of the screen

µ generally is not used in the design of an RF am-

plifier. In most tetrode applications, the screen amplification factor is useful to cate-
gorize roughly the performance to be expected.

Application Example

Figure 3.12

shows a radial beam power tetrode (4CX15000A) designed for class AB

1

or class C power amplification. The device is particularly well suited for RF linear
power amplifier service. The tube has a directly heated thoriated-tungsten mesh fila-
ment for mechanical ruggedness and high efficiency. The maximum rated plate dissi-
pation of the tube is 15 kW using air cooling.

The tube must be protected from damage that may result from an internal arc occur-

ring at high plate voltage. A protective resistance typically is inserted in series with the
tube anode to help absorb stored power supply energy in case an internal arc occurs.

The maximum control grid dissipation is 200 W, determined (approximately) by the

product of the dc grid current and the peak positive grid voltage.

Screen grid maximum dissipation is 450 W. With no ac applied to the screen grid,

dissipation is the product of dc screen voltage and dc screen current. Plate voltage, plate
loading, and/or bias voltage must never be removed while filament and screen voltages
are present.

The 4CX15000A must be mounted vertically, base up or down. The tube requires

forced-air cooling in all applications. The tube socket is mounted in a pressurized com-
partment where cooling air passes through the socket and is guided to the anode cooling
fins by an air chimney. Adequate movement of cooling air around the base of the tube

Figure 3.12

Radial beam power

tetrode (4CX15000A).

© 1999 CRC Press LLC

keeps the base and socket contact fingers at a safe operating temperature. Although the
maximum temperature rating for seals and the anode is 250°C, good engineering prac-
tice dictates that a safety factor be provided.

Table 3.1

lists cooling parameters for the

tube with the cooling air at 50°C and a maximum anode temperature of 225°C. The fig-
ures given in the table apply to designs in which air passes in the base-to-anode direc-
tion. Pressure drop values shown are approximate and apply to the tube/socket/chim-
ney combination.

At altitudes significantly above sea level, the flow rate must be increased for equiva-

lent cooling. At 5000 ft above sea level, both the flow rate and the pressure drop is in-
creased by a factor of 1.20; at 10,000 ft, both flow rate and pressure drop are increased
by 1.46.

Anode and base cooling is applied before or simultaneously with filament voltage

turn-on, and normally should continue for a brief period after shutdown to allow the
tube to cool properly.

An outline of the principal tube dimensions is given in

Figure 3.13

. General specifi-

cations are listed in

Table 3.2

.

3.2.7 Pentode

The pentode is a five-electrode tube incorporating three grids. The control and screen
grids perform the same function as in a tetrode. The third grid, the suppressor grid, is
mounted in the region between the screen grid and the anode. The suppressor grid
produces a potential minimum, which prevents secondary electrons from being inter-
changed between the screen and plate. The pentode's main advantages over the
tetrode include:

·

Reduced secondary emission effects.

·

Good linearity.

·

Ability to let plate voltage swing below the screen voltage without excessive
screen dissipation. This allows slightly higher power output for a given operating
plate voltage.

Table 3.1 Minimum Cooling Airflow Requirements for the 4CX15000A Power Tetrode
at Sea Level

Plate Dissipation (W)

Airflow (CFM)

Pressure Drop (inches of

water)

7500

230

0.7

12,500

490

2.7

15,000

645

4.6

© 1999 CRC Press LLC

Because of the design of the pentode, plate voltage has even less effect on plate cur-
rent than in the tetrode. The same total space-current equation applies to the pentode
as with the tetrode:

I

K

E

E

E

k

c

c

s

b

p

=

+

+

1

2

3

2

µ

µ

(3.11)

Where:
I

k

= cathode current

K = a constant determined by tube dimensions
E

c1

= control grid voltage

E

c2

= screen grid voltage

µ

s

= screen amplification factor

µ

p

= plate amplification factor

E

b

= plate voltage

The suppressor grid may be operated negative or positive with respect to the cath-

ode. It also may be operated at cathode potential. It is possible to control plate current

Figure 3.13 Principal dimensions of 4CX15000A tetrode.

© 1999 CRC Press LLC

Table 3.2 General Characteristics of the 4CX15000A Power Tetrode (

Courtesy of

Svetlana Electron Devices.)

Electrical Characteristics

Filament type

Thoriated-tungsten mesh

Filament voltage

6.3 ± 0.3 V

Filament current

164 A (at 6.3 V)

Amplification factor (average), grid to screen

4.5

Direct interelectrode capacitance
(grounded cathode)

C

in

158 pF

C

out

25.8 pF

C

pk

1.3 pF

Direct interelectrode capacitance
(grounded grid)

C

in

67 pF

C

out

25.6 pF

C

gk

0.21 pF

Maximum frequency for full ratings (CW)

110 MHz

Mechanical Characteristics:

Length

238 mm (9.38 in)

Diameter

193 mm (7.58 in)

Net weight

5.8 kg (12.8 lb)

Operating position

Axis vertical, base up or
down

Maximum operating temperature (seals/envelope)

250°C

Cooling method

Forced air

Base type

Coaxial

Radio Frequency Power Amplifier (class C FM) (absolute maximum ratings)

DC plate voltage

10,000 V

DC screen voltage

2000 V

DC grid voltage

750 V

DC plate current

5.0 A

Plate dissipation

15 kW

Screen dissipation

450 W

Grid dissipation

200 W

Typical Operation (frequencies up to 110 MHz)

DC plate voltage

7.5 kV dc

10.0 kV dc

DC screen voltage

750 V dc

750 V dc

DC grid voltage

510 V dc

550 V dc

DC plate current

4.65 A dc

4.55 A dc

DC screen current

0.59 A dc

0.54 A dc

DC grid current

0.30 A dc

0.27 A dc

Peak RF grid voltage

730 V

790 V

Calculated driving power

220 W

220 W

Plate dissipation

8.1

9.0 kW

Plate output power

26.7

36.5 kW

© 1999 CRC Press LLC

by varying the potential on the suppressor grid. Because of this ability, a modulating
voltage can be applied to the suppressor to achieve amplitude modulation. The required
modulating power is low because of the low electron interception of the suppressor.

Application Example

The 5CX1500B is a ceramic/metal power pentode designed for use as a class AB

1

lin-

ear amplifier or FM VHF amplifying device. The exterior of the device is shown in

Figure 3.14

. Basic specifications for the tube are given in

Table 3.3

.

The filament of the 5CX1500B is rated for 5.0 V. Filament voltage, measured at the

socket, is maintained to ±5 percent of the rated value for maximum tube life.

The rated dissipation of the control grid is 25 W. This value is approximately the

product of dc grid current and peak positive grid voltage. Operation at bias and drive
levels near those listed in

Table 3.3

will ensure safe operation.

The power dissipated by the screen grid must not exceed 75 W. Screen dissipation, in

cases where there is no ac applied to the screen, is the simple product of screen voltage
and screen current. If the screen voltage is modulated, screen dissipation will depend on
the rms screen current and voltage. Screen dissipation is likely to rise to excessive lev-
els if the plate voltage, bias voltage, or plate load are removed with filament and screen
voltages applied. Suitable protective means must be provided to limit screen dissipa-
tion to 75 W in the event of a circuit failure.

The rated dissipation of the suppressor grid is 25 W. Suppressor current is zero or

nearly zero for all typical operating conditions specified in

Table 3.3

. The 5CX1500B

Figure 3.14

5CX1500B power

pentode device.

© 1999 CRC Press LLC

Electrical Characteristics

Filament type

Thoriated-tungsten mesh

Filament voltage

5.0 ± 0.25 V

Filament current

38.5 A (at 5.0 V)

Transconductance (avg.),

I

b

= 1.0 A dc,

E

c2

= 500

V dc

24,000

µmhos

Amplification factor (average), grid to screen

5.5

Direct interelectrode ca-
pacitance (grounded cath-
ode)

Input

75 pF

Output

17.8 pF

Feedback

0.20 pF

Frequency (maximum) CW

110 MHz

Mechanical Characteristics

Cooling method

Forced air

Base

Ring and breechblock

Recommended air system socket

SK-840 series

Recommended (air) chimney

SK-806

Operating position

Axis vertical, base down or up

Maximum operating temperature

250°C

Maximum dimensions

Length

130 mm (5.2 in)

Diameter

85.6 mm (3.37 in)

Net weight

850 gm (30 oz)

Radio Frequency Linear Amplifier (class C, CW conditions) Maximum Ratings

Plate voltage

5000 V

Screen voltage

750 V

Plate dissipation

1500 W

Suppressor dissipation

25 W

Screen dissipation

75 W

Grid dissipation

25 W

Typical Operation (frequencies to 30 MHz)

Plate voltage

3000 V dc

4500 V dc

Suppressor voltage

0

0

Screen voltage

500 V dc

500 V dc

Grid voltage

200 V dc

200 V dc

Plate current

900 mA dc

900 mA dc

Screen current

94 mA dc

88 mA dc

Grid current

35 mA dc

34 mA dc

Peak RF grid voltage

255 V

255 V

Calculated driving power

9.0 W

9.0 W

Plate input power

2700 W

4050 W

Plate dissipation

720 W

870 W

Plate output power

1980 W

3180 W

Table 3.3 General Characteristics of the 5CX1500B Power Pentode (

Courtesy of

Svetlana Electron Devices.)

© 1999 CRC Press LLC

has been designed for zero voltage operation of the suppressor grid for most applica-
tions.

The plate dissipation rating of the 5CX1500B is 1500 W. The tube and associated

circuitry should be protected against surge current in the event of an arc inside the tube.
A current-limiting resistance of 10 to 25

between the power supply and plate is suffi-

cient. The resistor should be capable of withstanding the surge currents that may de-
velop; it should not be used as a fuse.

Cooling requirements for the 5CX1500B follow along the same lines as for the

4CX15000A discussed in the previous section.

Table 3.4

lists the airflow requirements

for the pentode. Note that the power dissipated by the device is equal to the power dissi-
pated by the plate, heater, and grids (control, screen, and suppressor) combined. In the
case of the 5CX1500B, the heater and three grids account for approximately 350 W dis-
sipation.

Mechanical dimensions for the 5CX1500B are given in

Figure 3.15

.

3.2.8 High-Frequency Operating Limits

As with most active devices, performance of a given vacuum tube deteriorates as the
operating frequency is increased beyond its designed limit. Electron transit time is a
significant factor in the upper-frequency limitation of electron tubes. A finite time is
taken by electrons to traverse the space from the cathode, through the grid, and travel
on to the plate. As the operating frequency increases, a point is reached at which the
electron transit-time effects become significant. This point is a function of the accel-
erating voltages at the grid and anode and their respective spacings. Tubes with re-
duced spacing in the grid-to-cathode region exhibit reduced transit-time effects.

A power limitation also is interrelated with the high-frequency limit of a device. As

the operating frequency is increased, closer spacing and smaller-sized electrodes must
be used. This reduces the power-handling capability of the tube.

Figure 3.16

illustrates

the relationship.

Gridded tubes at all power levels for frequencies up to about 1 GHz are invariably

cylindrical in form. At higher frequencies, planar construction is almost universal. As
the operating frequency is increased beyond design limits, output power and efficiency
both decrease.

Figure 3.17

illustrates the relationship.

Table 3.4 Minimum Cooling Airflow Requirements for the 5CX1500B Power Tetrode

Plate Dissipation (W)

Airflow (CFM)

Pressure Drop (inches of water)

At sea level

1000

27

0.33

1550

47

0.76

At 6000 ft

1000

33

0.40

1550

58

0.95

© 1999 CRC Press LLC

Transit time typically is not a problem for power grid tubes operating below 30

MHz. Depending on the application, power grid tubes can be used at 100 MHz and
above without serious consideration of transit time effects. Klystrons and other micro-
wave tubes actually take advantage of transit time, as discussed in Chapter 6.

Transit-Time Effects

When class C, class B, or similar amplifier operations are carried out at frequencies
sufficiently high that the transit time of the electrons is not a negligible fraction of the
waveform cycle, the following complications are observed in grid-based vacuum
tubes:

·

Back-heating of the cathode

·

Loading of the control grid circuit as a result of energy transferred to electrons
that do not necessarily reach the grid to produce a dc grid current

·

Debunching of plate current pulses

Figure 3.15 Principal dimensions of 5CX1500B pentode.

© 1999 CRC Press LLC

Figure 3.16 Continuous wave output power capability of a gridded vacuum tube.

Figure 3.17 Performance of a class C amplifier as the operating frequency is increased
beyond the design limits of the vacuum tube.

© 1999 CRC Press LLC

·

Phase differences between the plate current and the exciting voltage applied to
the control grid

Back-heating of the cathode occurs when the transit time in the grid-cathode space

is sufficiently great to cause an appreciable number of electrons to be in transit at the in-
stant the plate current pulse would be cut off in the case of low-frequency operation. A
considerable fraction of the electrons thus trapped in the interelectrode space are re-
turned to the cathode by the negative field existing in the grid-cathode space during the
cutoff portion of the cycle. These returning electrons act to heat the cathode. At very
high frequencies, this back-heating is sufficient to supply a considerable fraction of the
total cathode heating required for normal operation. Back-heating may reduce the life
of the cathode as a result of electron bombardment of the emitting surface. It also
causes the required filament current to depend upon the conditions of operation within
the tube.

Energy absorbed by the control grid as a result of input loading is transferred di-

rectly to the electron stream in the tube. Part of this stream acts to produce back-heating
of the cathode. The remainder affects the velocity of the electrons as they arrive at the
anode of the tube. This portion of the energy is not necessarily all wasted. In fact, a con-
siderable percentage of it may, under favorable conditions, appear as useful output in
the tube. To the extent that this is the case, the energy supplied by the exciting voltage to
the electron stream is simply transferred directly from the control grid to the output cir-
cuits of the tube without amplification.

An examination of the total time required by electrons to travel from the cathode to

the anode in a triode, tetrode, or pentode operated as a class C amplifier reveals that the
resulting transit times for electrons at the beginning, middle, and end of the current
pulse will differ as the operating frequency is increased. In general, electrons traversing
the distance during the first segment of the pulse will have the shortest transit time,
while those near the middle and end of the pulse will have the longest transit times, as il-
lustrated in

Figure 3.18

. The first electrons in the pulse have a short transit time because

they approach the plate before the plate potential is at its minimum value. Electrons
near the middle of the pulse approach the plate with the instantaneous plate potential at
or near minimum and, consequently, travel less rapidly in the grid-plate space. Finally,
those electrons that leave the cathode late in the current pulse (those just able to escape
being trapped in the control grid-cathode space and returned toward the cathode) will
be slowed as they approach the grid, and so have a large transit time. The net effect is to
cause the pulse of plate current to be longer than it would be in operation at a low fre-
quency. This causes the efficiency of the amplifier to drop at high frequencies, because
a longer plate current pulse increases plate losses.

3.3

Vacuum Tube Design

Any particular power vacuum tube may be designed to meet a number of operating
parameters, the most important usually being high operating efficiency and high
gain/bandwidth properties. Above all, the tube must be reliable and provide long op-
erating life. The design of a new power tube is a lengthy process that involves com-

© 1999 CRC Press LLC

Figure 3.18 Transit-time effects in a class C amplifier: (

a) control grid voltage, (b) elec-

tron position as a function of time (triode case), (

c) electron position as a function of time

(tetrode case), (

d) plate current (triode case).

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

puter-aided calculations and modeling. The design engineers must examine a laundry
list of items, including:

·

Cooling: how the tube will dissipate heat generated during normal operation. A
high-performance tube is of little value if it will not provide long life in typical ap-
plications. Design questions include whether the tube will be air-cooled or wa-
ter-cooled, the number of fins the device will have, and the thickness and spacing
of the fins.

·

Electro-optics: how the internal elements line up to achieve the desired perfor

-

mance. A careful analysis must be made of what happens to the electrons in their
paths from the cathode to the anode, including the expected power gain of the
tube.

·

Operational parameters: what the typical interelectrode capacitances will be,
and the manufacturing tolerances that can be expected. This analysis includes:
spacing variations among elements within the tube, the types of materials used in
construction, the long-term stability of the internal elements, and the effects of
thermal cycling.

3.3.1 Device Cooling

The main factor that separates tube types is the method of cooling used: air, water, or
vapor. Air-cooled tubes are common at power levels below 50 kW. A water cooling
system, although more complicated, is more effective than air cooling--by a factor of
5 to 10 or more--in transferring heat from the device. Air cooling at the 100 kW level
is virtually impossible because it is difficult to physically move enough air through
the device (if the tube is to be of reasonable size) to keep the anode sufficiently cool.
Vapor cooling provides an even more efficient method of cooling a power amplifier
(PA) tube than water cooling, for a given water flow and a given power dissipation.
Naturally, the complexity of the external blowers, fans, ducts, plumbing, heat
exchangers, and other hardware must be taken into consideration in the selection of a
cooling method.

Figure 3.19

shows how the choice of cooling method is related to an-

ode dissipation.

Air Cooling

A typical air cooling system for a transmitter is shown in

Figure 3.20

. Cooling system

performance for an air-cooled device is not necessarily related to airflow volume. The
cooling capability of air is a function of its mass, not its volume. An appropriate air-
flow rate within the equipment is established by the manufacturer, resulting in a given
resistance to air movement.

The altitude of operation is also a consideration in cooling system design. As alti-

tude increases, the density (and cooling capability) of air decreases. To maintain the
same cooling effectiveness, increased airflow must be provided. (Air cooling systems
are discussed in detail in Section 8.2.1.)

© 1999 CRC Press LLC

Figure 3.20 A typical transmitter PA stage cooling system.

Figure 3.19 The relationship between anode dissipation and cooling method.

© 1999 CRC Press LLC

Water Cooling

Water cooling usually is preferred over air cooling for power outputs above about 50
kW. Multiple grooves on the outside of the anode, in conjunction with a cylindrical
jacket, force the cooling water to flow over the surface of the anode, as illustrated in

Figure 3.21

.

Because the water is in contact with the outer surface of the anode, a high degree of

purity must be maintained. A resistivity of 1 m

-cm (at 25°C) typically is specified by

tube manufacturers. Circulating water can remove about 1 kW/cm

2

of effective internal

anode area. In practice, the temperature of water leaving the tube must be limited to
70°C to prevent the possibility of spot boiling.

After leaving the anode, the heated water is passed through a heat exchanger where it

is cooled to 30 to 40°C before being pumped back to the tube.

In typical modern water-cooled tubes, the dissipation ranges from 500 W to 1 kW

per square centimeter. Water-cooled anodes have deen designed that can operate at 10
kW per square centimeter and above. However, with the exception of "big science" ap-
plications, dissipations this high are seldom required. (Water cooling systems are dis-
cussed in detail in Section 8.2.2.)

Vapor-Phase Cooling

Vapor cooling allows the permissible output temperature of the water to rise to the
boiling point, giving higher cooling efficiency compared with water cooling. The
benefits of vapor-phase cooling are the result of the physics of boiling water. In-
creasing the temperature of one gram of water from 40 to 70°C requires 30 calories of
energy. However, transforming 1 g of water at 100°C into steam vapor requires 540
calories. Thus, a vapor-phase cooling system permits essentially the same cooling ca-
pacity as water cooling, but with greatly reduced water flow. Viewed from another

Figure 3.21 Water-cooled anode with grooves
for controlled water flow.

© 1999 CRC Press LLC

perspective, for the same water flow, the dissipation of the tube may be increased sig-
nificantly (all other considerations being the same).

A typical vapor-phase cooling system is shown in

Figure 3.22

. A tube incorporating

a specially designed anode is immersed in a boiler filled with distilled water. When
power is applied to the tube, anode dissipation heats the water to the boiling point, con-
verting the water to steam vapor. The vapor passes to a condenser, where it gives up its
energy and reverts to a liquid state. The condensate then is returned to the boiler, com-
pleting the cycle. Electric valves and interlocks are included in the system to provide for
operating safety and maintenance. A vapor-phase cooling system for a transmitter with
multiple PA tubes is shown in

Figure 3.23

. (Vapor-phase cooling systems are discussed

in detail in Section 8.2.3.)

Special Applications

Power devices used for research applications must be designed for transient overload-
ing, requiring special considerations with regard to cooling. Oil, heat pipes, refriger-
ants (such as Freon), and, where high-voltage hold-off is a problem, gases (such as
sulfahexafluoride) are sometimes used to cool the anode of a power tube.

3.3.2 Cathode Assembly

The ultimate performance of any vacuum tube is determined by the accuracy of de-
sign and construction of the internal elements. The requirements for a successful tube

Figure 3.22 Typical vapor-phase cooling system.

© 1999 CRC Press LLC

Figure 3.23 Vapor-phase cooling system for a 4-tube transmitter using a common water supply.

©

1
999
CRC Press LLC

include the ability to operate at high temperatures and withstand physical shock. Each
element is critical to this objective.

The cathode used in a power tube obtains the energy required for electron emission

from heat. The cathode may be directly heated (filament type) or indirectly heated. The
three types of emitting surfaces most commonly used are:

·

Thoriated-tungsten

·

Alkaline-earth oxides

·

Tungsten-barium-aluminate-impregnated emitters

The thoriated-tungsten and tungsten-impregnated cathodes are preferred in power
tube applications because they are more tolerant to ion bombardment. The character-
istics of the three emitting surfaces are summarized in

Table 3.5

.

A variety of materials may be used as a source of electrons in a vacuum tube. Certain

combinations of materials are preferred, however, for reasons of performance and eco-
nomics.

Oxide Cathode

The conventional production-type oxide cathode consists of a coating of barium and
strontium oxides on a base metal such as nickel. Nickel alloys, in general, are stron-
ger, tougher, and harder than most nonferrous alloys and many steels. The most im-
portant property of nickel alloys is their ability to retain strength and toughness at ele-
vated temperatures. The oxide layer is formed by first coating a nickel structure (a can
or disk) with a mixture of barium and strontium carbonates, suspended in a binder
material. The mixture is approximately 60 percent barium carbonate and 40 percent
strontium carbonate.

During vacuum processing of the tube, these elements are baked at high tempera-

tures. As the binder is burned away, the carbonates subsequently are reduced to oxides.
The cathode is then said to be activated and will emit electrons.

An oxide cathode operates CW at 700 to 820°C and is capable of an average emis-

sion density of 100 to 500 mA/cm

2

. High emission current capability is one of the main

Table 3.5 Characteristics of Common Thermonic Emitters

Emitter

Heating Method

1

Operating

Temp. (°C)

Emission Density (A/cm

2

)

Average

Peak

Oxide

Direct and indirect 700 to 820

0.10 to 0.50

0.10 to 20

Thoriated
tungsten

Direct

1600 to 1800

0.04 to 0.43

0.04 to 10

Impregnated
tungsten

Direct and indirect 900 to 1175

0.5 to 8.0

0.5 to 12

1

Directly heated refers to a filament-type cathode.

© 1999 CRC Press LLC

advantages of the oxide cathode. Other advantages include high peak emission for
short pulses and low operating temperature. As shown in

Table 3.5

, peak emission of up

to 20 A/cm

2

is possible from an oxide cathode. A typical device is shown in

Figure 3.24

.

Although oxide-coated emitters provide more peak emission per watt of heating

power than any other type, they are not without their drawbacks. Oxide emitters are
more easily damaged or "poisoned" than other emitters and also deteriorate more rap-
idly when subjected to bombardment from high-energy particles.

The oxide cathode material will evaporate during the life of the tube, causing free

barium to migrate to other areas within the device. This evaporation can be minimized
in the design stage by means of a high-efficiency cathode that runs as cool as possible
but still is not emission-limited at the desired heater voltage. In the field, the heater volt-
age must not exceed the design value. An oxide cathode that is overheated produces lit-
tle, if any, additional emission. The life of the tube under such operation, however, is
shortened significantly.

Thoriated-Tungsten Cathode

The thoriated-tungsten filament is another form of atomic-film emitter commonly
used in power grid tubes. Tungsten is stronger than any other common metal at tem-
peratures of more than 3500°F. The melting point of tungsten is 6170°F, higher than
that of any other metal. The electrical conductivity of tungsten is approximately
one-third that of copper, but much better than the conductivity of nickel, platinum, or
iron-based alloys. The resistivity of tungsten in wire form is exploited in filament ap-
plications. The thoriated-tungsten filament (or cathode) is created in a high-tempera-
ture gaseous atmosphere to produce a layer of ditungsten carbide on the surface of the

Figure 3.24 Photo of a typical oxide cath-
ode. (

Courtesy of Varian.)

© 1999 CRC Press LLC

cathode element(s). Thorium is added to tungsten in the process of making tungsten
wire. The thorium concentration is typically about 1.5 percent, in the form of thoria.
By proper processing during vacuum pumping of the tube envelope, the metallic tho-
rium is brought to the surface of the filament wire. The result is an increase in emis-
sion of approximately 1000 times over a conventional cathode.

At a typical operating temperature of 1600 to 1800°C, a thoriated-tungsten filament

will produce an average emission of 40 to 430 mA/cm

2

. Peak current ranges up to 10

A/cm

2

or more.

One of the advantages of a thoriated-tungsten cathode over an oxide cathode is the

ability to operate the plate at higher voltages. Because oxide cathodes are susceptible to
deterioration caused by ion bombardment, plate voltage must be limited. A
thoriated-tungsten cathode is more tolerant of ion bombardment, so higher plate volt-
ages can be safely applied.

The end of useful life for a thoriated-tungsten tube occurs when most of the carbon

has evaporated or has combined with residual gas, depleting the carbide surface layer.
Theoretically, a 3 percent increase in filament voltage will result in a 20 degree Kelvin
(K) increase in cathode temperature, a 20 percent increase in peak emission, and a 50
percent decrease in tube life because of carbon loss. This cycle works in reverse, too.
For a small decrease in temperature and peak emission, the life of the carbide
layer--hence, the tube--may be increased.

Tungsten-Impregnated Cathode

The tungsten-impregnated cathode typically operates at 900 to 1175°C and provides
the highest average emission density of the three types of cathodes discussed in this
section (500 mA/cm

2

to 8 A/cm

2

). Peak power performance ranges up to 12 A/cm

2

.

Tungsten, as an element, is better able than other emitters to withstand bombard-

ment by high-energy positive ions without suffering emission impairment. These posi-
tive ions are always present in small numbers in vacuum tubes as a result of ionization
by collision with the residual gas.

Cathode Construction

Power tube filaments can be assembled in several different configurations.

Figure

3.25

shows a spiral-type filament, and

Figure 3.26

shows a bar-type design. The spiral

filament is used extensively in low-power tubes. As the size of the tube increases, me-
chanical considerations dictate a bar-type filament with spring loading to compensate
for thermal expansion. A mesh filament can be used for both small and large tubes. It
is more rugged than other designs and less subject to damage from shock and vibra-
tion. The rigidity of a cylindrical mesh cathode is determined by the following param-
eters:

·

The diameter of the cathode

·

The number, thickness, and length of the wires forming the cathode

·

The ratio of welded to total wire crossings

© 1999 CRC Press LLC

A mesh cathode is shown in

Figure 3.27

.

Most power grid tubes are designed as a series of electron gun structures arranged in

a cylinder around a centerline. This construction allows large amounts of plate current
to flow and to be controlled with a minimum of grid interception. With reduced grid in-

Figure 3.25 Spiral-type tungsten filament.
(

Courtesy of Varian.)

Figure 3.26

Bar-type tungsten filament.

(

Courtesy of Varian.)

© 1999 CRC Press LLC

terception, less power is dissipated in the grid structures. In the case of the control grid,
less driving power is required for the tube.

In certain applications, the construction of the filament assembly may affect the per-

formance of the tube, and that of the RF system as a whole. For example, filaments built
in a basket-weave mesh arrangement usually offer lower distortion in critical high-level
AM modulation circuits.

Velocity of Emission

The electrons emitted from a hot cathode depart with a velocity that represents the
difference between the kinetic energy possessed by the electron just before emission
and the energy that must be given up to escape. Because the energy of different elec-
trons within the emitter is not the same, the velocity of emission will vary as well,
ranging from zero to a maximum value determined by the type and style of emitter.

3.3.3 Grid Structures

The type of grid used for a power tube is determined principally by the power level
and operating frequency required. For most medium-power tubes (5 to 25 kW dissipa-
tion) welded-wire construction is common. At higher power levels, laser-cut pyrolytic
graphite grids may be found. The grid structures of a power tube must maintain their
shape and spacing at elevated temperatures. They also must withstand shock and vi-
bration.

Figure 3.27

Mesh tungsten filament.

(

Courtesy of Varian.)

© 1999 CRC Press LLC

Wire Grids

Conventional wire grids are prepared by operators that wind the assemblies using spe-
cial mandrels (forms) that include the required outline of the finished grid. The opera-
tors spot weld the wires at intersecting points, as shown in

Figure 3.28

. Most grids of

this type are made with tungsten or molybdenum, which exhibit stable physical prop-
erties at elevated temperatures. On a strength basis, pure molybdenum generally is
considered the most suitable of all refractory metals at temperatures of 1600 to
3000°F. The thermal conductivity of molybdenum is more than three times that of
iron and almost half that of copper.

Grids for higher-power tubes typically are built using a bar-cage type of construc-

tion. A number of vertical supports are fastened to a metal ring at the top and to a base
cone at the bottom. The lower end of the assembly is bonded to a contact ring. The con-
struction of the ring, metal base cone, and cylindrical metal base give the assembly low
lead inductance and low RF resistance.

The external loading of a grid during operation and the proximity of the grid to the

hot cathode impose severe demands on both the mechanical stability of the structure
and the physical characteristics of its surface. The grid absorbs a high proportion of the
heat radiated by the cathode. It also intercepts the electron beam, converting part of its
kinetic energy into heat. Furthermore, high-frequency capacitive currents flowing in
the grid create additional heat.

The result is that grids are forced to work at temperatures as high as 1500°C. Their

primary and secondary emissions, however, must be low. To prevent grid emission,
high electron affinity must be ensured throughout the life of the tube, even though it is

Figure 3.28 Mesh grid structure. (

Courtesy

of Varian.)

© 1999 CRC Press LLC

impossible to prevent material that evaporates from the cathode from contaminating
the grid surface.

In tubes with oxide cathodes, grids made of tungsten or molybdenum wire are

coated with gold to reduce primary emission caused by deposition. The maximum safe
operating temperature for gold plating, however, is limited (about 550°C). Special
coatings developed for high-temperature applications are effective in reducing grid
emission. In tubes with thoriated-tungsten cathodes, grids made of tungsten or molyb-
denum are coated with proprietary compounds to reduce primary emission.

Primary grid emission is usually low in a thoriated tungsten cathode device. In the

case of an oxide cathode, however, free barium may evaporate from the cathode coating
material and find its way to the control and screen grids. The rate of evaporation is a
function of cathode temperature and time. A grid contaminated with barium will be-
come another emitting surface. The hotter the grid, the greater the emissions.

K-Grid

To permit operation at higher powers (and, therefore, higher temperatures) the K-grid
has been developed (Philips). The K-grid is a spot-welded structure of molybdenum
wire doped to prevent brittleness and recrystallization. To eliminate mechanical
stresses, the grid is annealed at 1800 K. It is then baked under vacuum at 2300 K to
degas the molybdenum wire. The grid structure next is coated with zirconium carbide
(sintered at 2300 K) and, finally, with a thin layer of platinum.

Despite high grid temperatures, the platinum coating keeps primary emissions to a

minimum. The surface structure maintains low secondary emissions.

Pyrolytic Grid

Pyrolytic grids are a high-performance alternative to wire or bar grid assemblies.
Used primarily for high-power devices, pyrolytic grids are formed by laser-cutting a
graphite cup of the proper dimensions. The computer-controlled laser cuts numerous
holes in the cup to simulate a conventional-style grid.

Figure 3.29

shows a typical

pyrolytic-type grid before and after laser processing.

Pyrolytic (or oriented) graphite is a form of crystallized carbon produced by the de-

composition of a hydrocarbon gas at high temperatures in a controlled environment. A
layer of pyrolytic graphite is deposited on a grid form. The thickness of the layer is pro-
portional to the amount of time deposition is allowed to continue. The structural and
mechanical properties of the deposited graphite depend upon the imposed conditions.

Pyrolytic grids are ideal vacuum tube elements because they do not expand like

metal. Their small coefficient of expansion prevents movement of the grids inside the
tube at elevated temperatures. This preserves the desired electrical characteristics of
the device. Because tighter tolerances can be maintained, pyrolytic grids can be spaced
more closely than conventional wire grids. Additional benefits are that the grid:

·

Is a single structure having no weld points

·

Has a thermal conductivity in two of the three planes nearly that of copper

© 1999 CRC Press LLC

·

Can operate at high temperatures with low vapor pressure

Grid Physical Structure

The control, screen, and suppressor grids are cylindrical and concentric. Each is
slightly larger than the previous grid, as viewed from the cathode. Each is fastened to
a metal base cone, the lower end of which is bonded to a contact ring.

Figure 3.30

shows the construction of a typical screen grid assembly.

Figure 3.31

provides a

cutaway view of a tetrode power tube.

The shape of the control grid and its spacing from the cathode define, in large part,

the operating characteristics of the tube. For best performance, the grid must be essen-
tially transparent to the electron path from the cathode to the plate. In a tetrode, the con-
trol and screen grids must be precisely aligned to minimize grid current. For pentode
tubes, these two conventions apply, in addition to the requirement for precise alignment
and minimum beam interception for the suppressor grid.

Secondary Emission Considerations

The relationship of the properties of secondary electrons to the grid structures and
other elements of a vacuum tube must be considered carefully in any design. As the

Figure 3.29 Pyrolytic graphite grid: (

a) before laser processing, (b) completed assem-

bly. (

Courtesy of Varian.)

(

a) (b)

© 1999 CRC Press LLC

power capability of a tube increases, the physical size of the elements also increases.
This raises the potential for secondary emission from the control, screen, and sup-
pressor grids. Secondary emission may occur regardless of the type of cathode used.

Figure 3.30 The screen grid assembly of a typical tetrode PA tube.

Figure 3.31 The internal arrangement of the anode, screen, control grid, and cathode
assemblies of a tetrode power tube.

© 1999 CRC Press LLC

The yield of secondary electrons may be reduced through the application of surface
treatments.

In a tetrode, the screen is operated at a relatively low potential, necessary to acceler-

ate the electrons emitted from the cathode. Not all electrons pass through the screen on
their way to the plate. Some are intercepted by the screen grid. As the electrons strike
the screen, other low-energy electrons are emitted. If these electrons have a stronger at-
traction to the screen, they will fall back to that element. If, however, they pass into the
region between the screen and the plate, the much higher anode potential will attract
them. The result is electron flow from screen to plate.

Because of the physical construction of a tetrode, the control grid will have virtually

no control over screen-to-plate current flow as a result of secondary electrons. During a
portion of the operating cycle of the device, it is possible that more electrons will leave
the screen grid than will arrive. The result will be a reverse electron flow on the screen
element, a condition common to high-power tetrodes. A low-impedance path for re-
verse electron flow must be provided.

Tube manufacturers typically specify the recommended value of bleeding current

from the screen power supply to counteract the emission current. Two common ap-
proaches are illustrated in

Figures 3.32

and

3.33

. If the screen power supply impedance

is excessively high in the direction of reverse electron flow, the screen voltage will at-
tempt to rise to the plate voltage. Note the emphasis on low impedance in the reverse
electron flow direction. Most regulated power supplies are low impedance in the for-
ward electron flow direction only. If the supply is not well bled, the reverse electrons
will try to flow from anode to cathode in the regulator series pass element. As the screen
voltage rises, the secondary and plate currents will increase, and the tube will enter a
runaway condition.

Figure 3.32 Circuit for providing a low-impedance screen supply for a tetrode.

© 1999 CRC Press LLC

As shown in

Figure 3.33

, the addition of a 12.5 k

resistor from screen to ground

provides a path for screen grid emission (20 mA for the circuit shown). In the circuit of

Figure 3.33

, plate current flows through the screen power supply, swamping the screen

supply. The screen power supply must, obviously, carry the normal screen and plate
currents. This scheme is used extensively in circuits where the screen is operated at dc
ground potential. The plate-to-cathode voltage is then the sum of the E

b

and E

c2

power

supplies.

The suppressor grid of a pentode lessens the effects of secondary emission in the de-

vice, reducing the requirement to provide a reverse electron flow path for the screen
grid power supply. The screen current requirement for a pentode, however, may be
somewhat higher than for a tetrode of the same general characteristics.

The designer also must consider the impedance of the control grid circuit. Primary

grid emission can result in operational problems if the grid circuit impedance is exces-
sively high. Primary grid emission, in the case of an oxide cathode tube, will increase
with tube life.

The size and power of gridded tubes dictate certain characteristics of electrical po-

tential. As this geometry increases in electrical terms, significant secondary emission
from the control grid can occur. Control grid secondary emission can exist whether the
cathode is a thoriated-tungsten or oxide emitter, and it can occur in a triode, tetrode, or
pentode. A typical curve of grid current as a function of grid voltage for a high-power
thoriated-tungsten filament tetrode is shown in

Figure 3.34

. As shown in the figure,

grid current decreases and eventually takes a reverse direction as the grid voltage in-
creases. This reduction and reversal of grid current can be explained by the normal sec-
ondary emission characteristics of the metals used in the grid structure.

The secondary emission characteristics of common metals are presented in curve

form in

Figure 3.35

. The ratio of secondary-to-primary electron current is given as a

function of primary electron potential. An examination of the chart will show the re-
gion between 200 and 600 V to be rather critical as far as secondary emission is con-

Figure 3.33 Series power supply configuration for swamping the screen circuit of a
tetrode.

© 1999 CRC Press LLC

cerned. Any power grid tube that normally operates with 200 to 600 V on the grid can
exhibit the negative resistance characteristic of decreasing grid current with increasing
grid voltage when another electrode--such as the anode in a triode or the screen grid in
a tetrode--is at a sufficiently high potential to attract the emitted electrons from the
control grid. A driver stage that works into such a nonlinear load normally must be de-

Figure 3.34 Typical curve of grid current as a function of control grid voltage for a
high-power thoriated-tungsten filament tetrode.

Figure 3.35 Secondary emission characteristics of various metals under ordinary con-
ditions.

© 1999 CRC Press LLC

signed in such a manner as to tolerate this condition. One technique involves swamping
the driver so that changes in the load resulting from secondary grid emission represent
only a small percentage of the total load the driver works into.

3.3.4 Plate Assembly

The plate assembly of a power tube is typically a collection of many smaller parts that
are machined and assembled to tight specifications. Copper generally is used to con-
struct the anode. It is an excellent material for this purpose because it has the highest
electrical conductivity of any metal except pure silver. Furthermore, copper is easily
fabricated and ideally suited to cold-forming operations such as deep drawing, bend-
ing, and stamping.

The anode and cooling fins (in the case of an air-cooled device) begin as flat sheets

of copper. They are stamped by the tube manufacturer into the necessary sizes and
shapes. After all of the parts have been machined, the anode and cooling fins are
stacked in their proper positions, clamped, and brazed into one piece in a brazing fur-
nace.

The plate of a power tube resembles a copper cup with the upper half of a plate con-

tact ring welded to the mouth and cooling fins silver-soldered or welded to the outside
of the assembly. The lower half of the anode contact ring is bonded to a base ceramic
spacer. At the time of assembly, the two halves of the ring are welded together to form a
complete unit, as shown in

Figure 3.36

.

In most power tubes, the anode is a part of the envelope and, because the outer sur-

face is external to the vacuum, it can be cooled directly. Glass envelopes were used in
older power tubes. Most have been replaced, however, with devices that use ceramic as
the envelope material.

3.3.5 Ceramic Elements

Ceramics are an integral part of modern power vacuum tubes. Three types of ceramics
are in common usage in the production of vacuum devices:

·

Aluminum oxide-based ceramics

·

Beryllium oxide (BeO)-based ceramics

·

Aluminum nitride (AlNi)-based ceramics

Aluminum Oxide Ceramics

Aluminum oxide-based ceramic insulators are a common construction material for a
wide variety of electric components, including vacuum tubes. Aluminum oxide is 20
times higher in thermal conductivity than most oxides. The flexure strength of com-
mercial high-alumina ceramics is two to four times greater than that of most oxide ce-
ramics. There are drawbacks, however, to the use of alumina ceramics, including:

© 1999 CRC Press LLC

·

Relatively high thermal expansion (approximately 7 ppm/°C), compared to other
ceramic materials, such as BeO

·

Moderately high dielectric constant (approximately 10)

Aluminas are fabricated from aluminum oxide powders with various percentages of

sintering promoters, which produce the glassy phase. The later additives reduce the
densification temperatures to between 1500 and 1600°C. Based on the final applica-
tion, the powders may be pressed, extruded, or prepared in slurries for slip casting or
tape casting. The surface finish of aluminas is typically 3 to 25

µm/in as a result of nor-

mal processing. For very smooth finishes (2

µm/in), the surfaces may be lapped or pol-

ished.

Alumina ceramics for vacuum tube applications rarely are used apart from being

bonded to metals. The means by which this is accomplished frequently dictate the pro-
cessing technique. Metallization of aluminas usually is accomplished by either
high-temperature firing or low-temperature thick-film processing. Fired, shaped
aluminas usually are postmetallized by refiring the formed article after it has been
coated with a slurry of molybdenum and manganese powder or tungsten metal powder.
Based on the purity of the alumina, glass powder may be added to the metal powder.
The mechanism of metallization requires an internal glassy phase or the added glassy
phase for proper bonding. This is accomplished by firing in slightly reducing and moist
atmospheres at temperatures above 1500°C. The resulting metallization usually is
plated electrochemically with nickel, copper, or both.

Figure 3.36 A cutaway view of the anode structure of an RF power amplifier tube.

© 1999 CRC Press LLC

Thick-film processing of alumina ceramics traditionally has been performed in oxi-

dizing atmospheres at moderate temperatures (800 to 1000°C). Precious metals are
used, such as gold, silver, platinum, palladium, and their combinations. A glassy phase
usually is incorporated in the thick-film paste.

Each approach to metallization has advantages and disadvantages. The advantage of

the high-temperature metallization schemes with molybdenum or tungsten is their
moderate cost. The disadvantage is the high resistivity of the resulting films. These at-
tributes are reversed for thick-film materials. Another advantage of thick-film
metallization is that the process often is applied by the device fabricator, not the ce-
ramic vendor. This can allow for greater efficiency in design modification.

In tubes designed for operation at 100 MHz and below, alumina typically is the ce-

ramic of choice.

Beryllium Oxide Ceramics

Beryllium oxide-based ceramics are in many ways superior to alumina-based ceram-
ics. The major drawback is the toxicity of BeO. Beryllium and its compounds are a
group of materials that are potentially hazardous and must be handled properly. With
the necessary safeguards, BeO has been used successfully in many tube designs.

Beryllium oxide materials are particularly attractive for use in power vacuum tubes

because of their electrical, physical, and chemical properties. The thermal conductivity
of BeO is approximately 10 times higher than that of alumina-based materials.

Figure

3.37

compares the thermal conductivity of BeO with that of alumina and some alterna-

tive materials. As the chart illustrates, BeO has a lower dielectric constant and coeffi-
cient of thermal expansion than alumina. It is, however, also slightly lower in strength.

Figure 3.37 Thermal conductivity of common ceramics.

© 1999 CRC Press LLC

Beryllia materials are fabricated in much the same way as alumina compounds, al-

though the toxic properties of the powders mandate that they be processed in laborato-
ries equipped to handle them safely. Simple cutting, drilling, and postmetallization also
are handled by vendors with special equipment. Reliable thick-film systems may be ap-
plied to BeO substrates. Such coatings usually necessitate less elaborate safety precau-
tions and are often applied by the device fabricator. General safety precautions relating
to BeO ceramics are given in Section 11.3.2.

Other Ceramics

Aluminum nitride (AlNi)-based ceramics have been developed as an alternative to the
toxicity concerns of BeO-based materials. As shown in

Figure 3.37

, the thermal con-

ductivity of AlNi is comparable to that of BeO but deteriorates less with temperature.
The dielectric constant of AlNi is comparable to that of alumina (a drawback), but its
thermal expansion is low (4 ppm/°C).

Other ceramic materials that find some use in vacuum devices include silicon car-

bide (SiC)-based substances and boron nitride ceramics.

3.3.6 Tube Construction

Each type of power grid tube is unique insofar as its operating characteristics are con-
cerned. The basic physical construction, however, is common to most devices. A vac-
uum tube is built in essentially two parts:

·

The base, which includes the filament and supporting stem, control grid, screen
grid, and suppressor grid (if used)

·

The anode, which includes the heat-dissipating fins made in various machining
steps

The base subassembly is welded using a tungsten-inert gas (TIG) process in an oxy-
gen-free atmosphere (a process sometimes referred to as Heliarc welding) to produce
a finished base unit.

The ceramic elements used in a vacuum tube are critical parts of the device. Assem-

bled in sections, each element builds upon the previous one to form the base of the tube.
The ceramic-to-metal seals are created using a material that is painted onto the ceramic
and then heated in a brazing oven. After preparation in a high-temperature oven, the
painted area provides a metallic structure that is molecularly bonded to the ceramic and
provides a surface suitable for brazing.

This process requires temperature sequences that dictate completion of the high-

est-temperature stages first. As the assembly takes form, lower oven temperatures are
used so that completed bonds will not be compromised.

Despite all the advantages that make ceramics one of the best insulators for a tube

envelope, their brittleness is a potential cause of failure. The smallest cracks in the ce-
ramic, not in themselves damaging, can cause the ceramic to break when mechanically
stressed by temperature changes.

© 1999 CRC Press LLC

After the base assembly has been matched with the anode, the completed tube is

brazed into a single unit. The device then goes through a bake-out procedure. Baking
stations are used to evacuate the tube and bake out any oxygen or other gases from the
copper parts of the assembly. Although oxygen-free copper is used in tube construc-
tion, some residual oxygen exists in the metal and must be driven out for long compo-
nent life. A typical vacuum reading of 10

-8

Torr is specified for most power tubes (for-

merly expressed as 10

-8

mm of mercury). For comparison, this is the degree of vacuum

in outer space about 200 miles above the earth.

A vacuum offers excellent electrical insulation characteristics. This property is es-

sential for reliable operation of a vacuum tube, the elements of which typically operate
at high potentials with respect to each other and to the surrounding environment. An
electrode containing absorbed gases, however, will exhibit reduced breakdown voltage
because the gas will form on the electrode surface, increasing the surface gas pressure
and lowering the breakdown voltage in the vicinity of the gas pocket.

To maintain a high vacuum during the life of the component, power tubes contain a

getter device. The name comes from the function of the element: to "get" or trap and
hold gases that may exist inside the tube. Materials used for getters include zirconium,
cerium, barium, and titanium.

The operation of a vacuum tube is an evolving chemical process. End of life in a vac-

uum tube generally is caused by loss of emission.

3.3.7 Connection Points

The high power levels and frequencies at which vacuum tubes operate place stringent
demands on the connectors used to tie the outside world to the inside elements. Tubes
are designed to be mounted vertically on their electrical connectors. The connectors
provide a broad contact surface and mechanical support for the device.

The cathode and grids typically are mounted on ring-shaped Kovar bases, which

also serve as contact rings for the external connections. Kovar is an iron-nickel-cobalt
alloy whose coefficient of thermal expansion is comparable to that of aluminum oxide
ceramic. The different diameters of the various contact rings allow them to be grouped
coaxially. The concentric tube/connector design provides for operation at high frequen-
cies. Conductivity is improved by silver plating.

3.3.8 Tube Sockets

Any one tube design may have several possible socket configurations, depending
upon the frequency of operation. If the tube terminals are large cylindrical surfaces,
the contacting portions of the socket consist of either spring collets or an assembly of
spring fingerstock. Usually, these multiple-contacting surfaces are made of beryllium
copper to preserve spring tension at the high temperatures present at the tube termi-
nals. The fingers are silver-plated to reduce RF resistance.

Figure 3.38

shows a

cutaway view of the base of a tetrode socket.

If the connecting fingers of a power tube socket fail to provide adequate contact with

the tube element rings, a concentration of RF currents will result. Depending on the ex-

© 1999 CRC Press LLC

tent of this concentration, damage may result to the socket. After a connecting finger
loses its spring action, the heating effect is aggravated, and tube damage is possible.

A series of specialized power tubes is available with no sockets at all. Intended for

cathode-driven service, the grid assembly is formed into a flange that is bolted to the
chassis. The filament leads are connected via studs on the bottom of the tube. Such a
configuration eliminates the requirement for a socket. This type of device is useful for
low-frequency applications, such as induction heating.

3.4

Neutralization

An RF power amplifier must be properly neutralized to provide acceptable perfor-
mance in most applications. The means to accomplish this end vary considerably
from one design to another. An RF amplifier is neutralized when two operating condi-
tions are met:

·

The interelectrode capacitance between the input and output circuits is canceled.

·

The inductance of the screen grid and cathode assemblies (in a tetrode) is can-
celed.

Cancellation of these common forms of coupling between the input and output cir-
cuits of vacuum tube amplifiers prevents self-oscillation and the generation of spuri-
ous products.

Figure 3.38 Cross section of the base of a tetrode socket showing the connection
points.

© 1999 CRC Press LLC

3.4.1 Circuit Analysis

Figure 3.39

illustrates the primary elements that affect neutralization of a vacuum

tube RF amplifier operating in the VHF band. (Many of the following principles also
apply to lower frequencies.) The feedback elements include the residual grid-to-plate
capacitance (C

gp

), plate-to-screen capacitance (C

ps

), and screen grid lead inductance

(L

s

). The RF energy developed in the plate circuit (E

p

) causes a current (I) to flow

through the plate-to-screen capacitance and the screen lead inductance. The current
through the screen inductance develops a voltage (E) with a polarity opposite that of
the plate voltage (E

p

). The E potential often is used as a method of neutralizing

tetrode and pentode tubes operating in the VHF band.

Figure 3.40

graphically illustrates the electrical properties at work. The circuit ele-

ments of the previous figure have been arranged so that the height above or below the
zero potential line represents the magnitude and polarity of the RF voltage for each part
of the circuit with respect to ground (zero). For the purposes of this illustration, assume
that all of the circuit elements involved are pure reactances. The voltages represented
by each, therefore, are either in phase or out of phase and can be treated as positive or
negative with respect to each other.

The voltages plotted in the figure represent those generated as a result of the RF out-

put circuit voltage (E

p

). No attempt is made to illustrate the typical driving current on

the grid of the tube. The plate (P) has a high positive potential above the zero line, estab-
lished at the ground point. Keep in mind that the distance above the baseline represents
increasing positive potential. The effect of the out-of-phase screen potential developed
as a result of inductance L

s

is shown, resulting in the generation of E.

Figure 3.39 The elements involved in the neutralization of a tetrode PA stage.

© 1999 CRC Press LLC

As depicted, the figure constitutes a perfectly neutralized circuit. The grid potential

rests at the zero baseline. The grid operates at filament potential insofar as any action of
the output circuit on the input circuit is concerned.

The total RF voltage between plate and screen is made up of the plate potential and

screen lead inductance voltage, E. This total voltage is applied across a divider circuit
that consists of the grid-to-plate capacitance and grid-to-screen capacitance (C

gp

and

C

gs

). When this potential divider is properly matched for the values of plate RF voltage

(E

p

) and screen lead inductance voltage (E), the control grid will exhibit zero voltage

difference with respect to the filament as a result of E

p

.

3.4.2 Circuit Design

A variety of methods may be used to neutralize a vacuum tube amplifier. Generally
speaking, a grounded-grid, cathode-driven triode can be operated into the VHF band
without external neutralization components. The grounded-grid element is sufficient
to prevent spurious oscillations. Tetrode amplifiers generally will operate through the
MF band without neutralization. However, as the gain of the stage increases, the need
to cancel feedback voltages caused by tube interelectrode capacitances and external
connection inductances becomes more important. At VHF and above, it is generally
necessary to provide some form of stage neutralization.

Below VHF

For operation at frequencies below the VHF band, neutralization for a tetrode typi-
cally employs a capacitance bridge circuit to balance out the RF feedback caused by

Figure 3.40

A graphical representation of the elements involved in the

self-neutralization of a tetrode RF stage.

© 1999 CRC Press LLC

residual plate-to-grid capacitance. This method assumes that the screen is well by-
passed to ground, providing the expected screening action inside the tube.

Neutralization of low-power push-pull tetrode or pentode tubes can be accom-

plished with cross-neutralization of the devices, as shown in

Figure 3.41

. Small-value

neutralization capacitors are used. In some cases, neutralization can be accomplished
with a simple wire connected to each side of the grid circuit and brought through the
chassis deck. Each wire is positioned to "look at" the plate of the tube on the opposite
half of the circuit. Typically, the wire (or a short rod) is spaced a short distance from the
plate of each tube. Fine adjustment is accomplished by moving the conductor in or out
from its respective tube.

A similar method of neutralization can be used for a cathode-driven symmetrical

stage, as shown in

Figure 3.42

. Note that the neutralization capacitors (C

n

) are con-

nected from the cathode of one tube to the plate of the opposite tube. The neutralizing
capacitors have a value equal to the internal cathode-to-plate capacitance of the PA
tubes.

In the case of a single-ended amplifier, neutralization can be accomplished using ei-

ther a push-pull output or push-pull input circuit.

Figure 3.43

shows a basic push-pull

grid neutralization scheme that provides the out-of-phase voltage necessary for proper
neutralization. It is usually simpler to create a push-pull network in the grid circuit than
in the plate because of the lower voltages present. The neutralizing capacitor, C

n

, is

small and may consist of a simple feedthrough wire (described previously). A padding
capacitor in parallel with C

1

often is added to maintain the balance of the input circuit

while tuning. The padding capacitor generally is equal in size to the input capacitance
of the tube.

Single-ended tetrode and pentode stages also can be neutralized using the method

shown in

Figure 3.44

. The input resonant circuit is placed above ground by a small

amount because of the addition of capacitor C. The voltage to ground that develops

Figure 3.41 Push-pull grid neutralization.

© 1999 CRC Press LLC

across C upon the application of RF drive is out of phase with the grid voltage, and is fed
back to the plate through C

n

to provide neutralization. In such a design, C

n

is consider-

ably larger in value than the grid-to-plate interelectrode capacitance.

The single-ended grid neutralization circuit is redrawn in

Figure 3.45

to show the ca-

pacitance bridge that makes the design work. Balance is obtained when the following
condition is met:

C

C

C

C

n

gp

gf

=

(3.12)

Figure 3.43 Push-pull grid neutralization in a single-ended tetrode stage.

Figure 3.42 Symmetrical stage neutralization for a grounded-grid circuit.

© 1999 CRC Press LLC

Where:
C

n

= neutralization capacitance

C = input circuit bypass capacitor
C

gp

= grid-to-plate interelectrode capacitance

C

gf

= total input capacitance, including tube and stray capacitance

A single-ended amplifier also can be neutralized by taking the plate circuit slightly

above ground and using the tube capacitances as part of the neutralizing bridge. This
circuit differs from the usual RF amplifier design in that the plate bypass capacitor is re-
turned to the screen side of the screen bypass capacitor, as shown in

Figure 3.46

. The

size of screen bypass capacitor C

s

and the amount of stray capacitances in C

p

are chosen

Figure 3.44 Single-ended grid neutralization for a tetrode.

Figure 3.45 The previous figure redrawn to show the elements involved in neutraliza-
tion.

© 1999 CRC Press LLC

to balance the voltages induced in the input by the internal tube capacitances,
grid-to-plate (C

gp

) and screen-to-grid (C

sg

). This circuit is redrawn in

Figure 3.47

in the

usual bridge form. Balance is obtained when the following condition is met:

C

C

C

C

p

s

gp

sg

=

(3.13)

In usual tetrode and pentode structures, the screen-to-grid capacitance is approxi-

mately half the published tube input capacitance. The tube input capacitance is primar-
ily the sum of the grid-to-screen capacitance and the grid-to-cathode capacitance.

Note that in the examples given, it is assumed that the frequency of operation is low

enough that inductances in the socket and connecting leads can be ignored. This is basi-

Figure 3.46 Single-ended plate neutralization.

Figure 3.47 Single-ended plate neutralization showing the capacitance bridge present.

© 1999 CRC Press LLC

cally true in MF applications and below. At higher bands, however, the effects of stray
inductances must be considered, especially in single-ended tetrode and pentode stages.

VHF and Above

Neutralization of power grid tubes operating at very high frequencies provides special
challenges and opportunities to the design engineer. At VHF and above, significant
RF voltages may develop in the residual inductance of the screen, grid, and cathode
elements. If managed properly, these inductances can be used to accomplish neutral-
ization in a simple, straightforward manner.

At VHF and above, neutralization is required to make the tube input and output cir-

cuits independent of each other with respect to reactive currents. Isolation is necessary
to ensure independent tuning of the input and output. If variations in the output voltage
of the stage produce variations of phase angle of the input impedance, phase modula-
tion will result.

As noted previously, a circuit exhibiting independence between the input and output

circuits is only half of the equation required for proper operation at radio frequencies.
The effects of incidental inductance of the control grid also must be canceled for com-
plete stability. This condition is required because the suppression of coupling by capac-
itive currents between the input and output circuits is not, by itself, sufficient to negate
the effects of the output signal on the cathode-to-grid circuit. Both conditions--input
and output circuit independence and compensation for control grid lead induc-
tance--must be met for complete stage stability at VHF and above.

Figure 3.48

shows a PA stage employing stray inductance of the screen grid to

achieve neutralization. In this grounded-screen application, the screen is bonded to the
cavity deck using six short connecting straps. Two additional adjustable ground straps
are set to achieve neutralization.

3.4.3 Grounded-Grid Amplifier Neutralization

Grounded-grid amplifiers offer an attractive alternative to the more common
grid-driven circuit. The control grid is operated at RF ground and serves as a shield to
capacitive currents from the output to the input circuit. Generally, neutralization is not
required until the control grid lead inductive reactance becomes significant. The feed-
back from the output to the input circuit is no longer the result of plate-to-filament ca-
pacitance. The physical size of the tube and the operating frequency determine when
neutralization is required.

Two methods of neutralization commonly are used with grounded-grid amplifiers.

In the first technique, the grids of a push-pull amplifier are connected to a point having
zero impedance to ground, and a bridge of neutralizing capacitances is used that is
equal to the plate-filament capacitances of the tubes.

The second method of neutralization requires an inductance between the grid and

ground, or between the grids of a push-pull amplifier, of a value that will compensate
for the coupling between input and output circuits resulting from the internal capaci-
tances of the tubes.

© 1999 CRC Press LLC

Behavior of these two circuits is quite different. They may be considered as special

forms of the more general case in which the neutralizing capacitors have values differ-
ing from the internal capacitances of the tubes, and in which an appropriate reactance is
connected between the grids. Under these conditions, the value of neutralizing capaci-

Figure 3.48 A grounded-screen PA stage neutralized through the use of stray induc-
tance between the screen connector and the cavity deck: (

a) schematic diagram, (b)

mechanical layout of cavity.

(

a)

(

b)

© 1999 CRC Press LLC

tance permits continuous variation of power amplification, stability, and negative feed-
back.

Grid Impedance

In the special case of a grounded-grid amplifier having a grid impedance and the re-
active currents neutralized, the following equations apply (see

Figure 3.49

):

C

C

C

n

fp

fg

=

-

µ

(3.14)

(

)

Z

j C

C

g

fg

gp

= -

+

+

1

1

µ

(3.15)

If in solving the equation for C

n

the sign is negative, this indicates that in-phase neu-

tralization is required. Conversely, if the sign of C

n

is positive, then out-of-phase neu-

tralization is needed. A negative value of Z

g

indicates capacitive reactance required,

and a positive value indicates that inductive reactance is to be used.

Application Example

If the grids of a push-pull cathode-driven amplifier are not at ground potential be-
cause the inductance of the leads is not negligible, coupling may exist between the in-
put and output circuits through the plate-grid capacitances, cathode-grid capaci-
tances, and grid-to-grid inductance. One method of reducing this coupling is to insert
between the grids a series-tuned circuit that has zero reactance at the operating fre-
quency. This technique is illustrated in

Figure 3.50

. This neutralization scheme is use-

Figure 3.49 Circuit of grounded-grid amplifier having grid impedance and neutralized
for reactive currents.

© 1999 CRC Press LLC

ful only for the case in which no grid current flows. If grid current flows, a grid resis-
tance will appear in parallel with the grid-to-filament capacitance. If the resistance is
small in comparison with the reactance of this grid-to-filament capacitance, phase
modulation will result.

Another important property of the preceding neutralization scheme is that power

amplification is a function of the neutralizing capacitance, while the independence of
cathode and plate circuits from the viewpoint of reactive currents may be obtained with
any value of neutralizing capacitance. If the neutralizing capacitance is less than the
plate-to-filament capacitance of the tube, the stage will operate with low excitation
power and high power amplification.

If the neutralizing capacitance is greater than the plate-to-filament capacitance, the

power amplification will be quite low, but the total output power possible will be in-
creased.

3.4.4 Self-Neutralizing Frequency

The voltage-dividing action between the plate-to-grid capacitance (C

pg

) and the

grid-to-screen capacitance (C

gs

) will not change with variations in operating fre-

quency. The voltage division between the plate and screen, and screen and ground
caused by the charging current (I) will, however, vary significantly with frequency.
There will be a particular frequency, therefore, at which this potential-dividing circuit

Figure 3.50 Neutralization by cross-connected capacitors of a symmetrical cath-
ode-excited amplifier wth compensation of lead inductance.

© 1999 CRC Press LLC

will effectively place the grid at filament potential insofar as the plate is concerned.
This point is known as the self-neutralizing frequency, illustrated in

Figure 3.51

.

At the self-neutralizing frequency, the tetrode or pentode is inherently neutralized

by the circuit elements within the device itself, and external screen inductance to
ground. When a device is operated below its self-neutralizing frequency, the normal
cross-neutralization circuits apply. When the operating frequency is above the self-neu-
tralizing frequency, the voltage developed in the screen lead inductance is too large to
give the proper voltage division between the internal capacitances of the device. One
approach to neutralization in this case involves adjusting the inductive reactance of the
screen lead to ground to lower the total reactance. In the example shown in

Figure 3.52

,

this is accomplished with a series variable capacitor.

Another approach is shown in

Figure 3.53

, in which the potential divider network

made up of the tube capacitance is changed. In the example, additional plate-to-grid ca-
pacitance is added external to the tube. The external capacitance (C

ext

) can take the form

of a small wire or rod positioned adjacent to the plate of the tube. This approach is simi-
lar to the one described in Section 3.4.2, except that in this case the neutralizing probe is
connected to the grid of the tube, rather than to an opposite polarity in the circuit.

If the RF power amplifier is operating above the self-neutralizing frequency of the

tube and must be tuned over a range of frequencies, it is probably easier to use the
screen series-tuning capacitor method and make this control available to the operator. If
operation is desired over a range of frequencies including the self-neutralizing fre-
quency of the tube, this circuit is also desirable because the incidental lead inductance
in the variable capacitor lowers the self-neutralizing frequency of the circuit so that the
neutralizing series capacitor can be made to operate over the total desired frequency
range. If this range is too great, switching of neutralizing circuits will be required. A
small 50 to 100 pF variable capacitor in the screen lead often has been found to be satis-
factory.

Figure 3.51

Graphical representation of the elements of a tetrode when

self-neutralized.

© 1999 CRC Press LLC

Another method of changing the self-neutralizing frequency of a tetrode or pentode

can be fashioned from the general bypassing arrangement of the screen and filament
shown in

Figure 3.54

. The screen lead is bypassed with minimum inductance to the fil-

ament terminal of the tube. Some inductance is introduced in the common filament and
screen ground leads. The grid is shown below the zero voltage or chassis potential, indi-
cating that the voltage developed in the screen lead inductance to chassis is excessive. If
the filament is tapped on this inductance, a point can be found where the voltage differ-

Figure 3.52 Components of the output voltage of a tetrode when neutralized by adding
series screen-lead capacitance.

Figure 3.53 Components of the output voltage of a tetrode when neutralized by adding
external grid-to-plate capacitance.

© 1999 CRC Press LLC

ence between the grid and filament is zero, as far as the components of plate voltage are
concerned. This arrangement will be found to self-neutralize at a higher frequency than
if the filament and screen were bypassed separately to the chassis. Thus, increasing the
self-neutralizing frequency of the tube and screen bypass arrangement reduces the ten-
dency of the VHF parasitic to occur.

If the frequency of the VHF parasitic is reduced by increasing the inductance of the

plate lead (presuming this is the principal frequency-defining circuit), the circuit can be
made to approach the self-neutralizing frequency of the tube and, therefore, suppress
the parasitic.

3.4.5 Neutralization Adjustment

Most neutralization circuits must be adjusted for operation at a given frequency. The
exact procedure followed to make these adjustments varies from one circuit to the
next. The following generalizations, however, apply to most systems.

The first step in the process of neutralization is to break the dc connections to the

plate voltage and screen voltage supplies, leaving the RF circuits intact. If the direct
current path is not broken, some current can flow in either of these circuits even though
the voltages are zero. The presence of this current causes the amplifier to work in the
normal manner, generating RF power in the plate circuit. It would then be incorrect to
adjust for zero power in the plate circuit. Sufficient RF grid drive must be applied to
provide some grid current or to cause a sensitive RF meter coupled to the plate to give
an indication of feedthrough power. When the plate circuit is tuned through resonance,
the grid current will dip when the circuit is out of neutralization, or the RF meter will
peak. The neutralization adjustments are made until the indication is minimum.

Another powerful tool for roughly neutralizing an RF amplifier is to feed the power

output from a signal generator into the grid circuit. A sensitive RF detector is inserted

Figure 3.54 Components of the output voltage of a tetrode neutralized by adding induc-
tance common to the screen and cathode return.

© 1999 CRC Press LLC

between the output connector and the load. Neutralization can then be adjusted for min-
imum feedthrough. This technique is useful in working with prototype equipment. Ac-
tual qualitative measurements can be made. If the insertion loss of the amplifier is less
than the expected gain, oscillation will occur. Circuit modifications can be made until
the isolation is sufficient to warrant a test with high voltages applied. The advantages of
this "cold" system test are that:

·

No components are subjected to unusual stress if the amplifier is unstable.

·

Circuit adjustments can be made safely because no high voltages are present.

For final trimming of the neutralization adjustment, the stage should be returned to

operating condition at reduced power (similar to that used when testing for parasitic os-
cillations), or under the final loaded operating conditions. At higher frequencies, par-
ticularly in the VHF region, it will be found that a small additional trimming adjustment
of the neutralization circuit is usually required. When the plate circuit is tuned through
resonance, minimum plate current and maximum control grid current should occur si-
multaneously. In the case of the tetrode and pentode, the dc screen current should be
maximum at the same time.

These neutralizing procedures apply not only to the HF radio frequencies, but also to

the VHF and UHF regions.

3.5

References

1. Terman, F. E., Radio Engineering, 3rd Ed., McGraw-Hill, New York, 1947.
2 . Ferris, Clifford D., "Electron Tube Fundamentals," in The Electronics Handbook,

Jerry C. Whitaker (ed.), CRC Press, Boca Raton, FL, pp. 295305, 1996.

3.6

Bibliography

Birdsall, C. K., Plasma Physics via Computer Simulation, Adam Hilger, 1991.
Block, R., "CPS Microengineers New Breed of Materials," Ceramic Ind., pp. 5153,

April 1988.

Buchanan, R. C., Ceramic Materials for Electronics, Marcel Dekker, New York, 1986.
"Ceramic Products," Brush Wellman, Cleveland, OH.
Chaffee, E. L., Theory of Thermonic Vacuum Tubes, McGraw-Hill, New York, 1939.
"Combat Boron Nitride, Solids, Powders, Coatings," Carborundum Product Litera-

ture, form A-14, 011, September 1984.

"Coors Ceramics--Materials for Tough Jobs," Coors Data Sheet K.P.G.-2500-2/87

6429.

Cote, R. E., and R. J. Bouchard, "Thick Film Technology," in Electronic Ceramics, L.

M. Levinson (ed.), Marcel Dekker, New York, pp. 307370, 1988.

Dettmer, E. S., and H. K. Charles, Jr., "AlNi and SiC Substrate Properties and Pro-

cessing Characteristics," Advances in Ceramics, Vol. 31, American Ceramic Soci-
ety, Columbus, OH, 1989.

© 1999 CRC Press LLC

Dettmer, E. S., H. K. Charles, Jr., S. J. Mobley, and B. M. Romenesko, "Hybrid Design

and Processing Using Aluminum Nitride Substrates," ISHM 88 Proc., pp. 545553,
1988.

Eastman, Austin V., Fundamentals of Vacuum Tubes, McGraw-Hill, 1941.
Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 3rd Ed.,

McGraw-Hill, New York, 1989.

Floyd, J. R., "How to Tailor High-Alumina Ceramics for Electrical Applications," Ce-

ramic Ind., pp. 4447, February 1969; pp. 4649, March 1969.

Gray, Truman S., Applied Electronics, John Wiley & Sons, New York, 1954.
Harper, C. A., Electronic Packaging and Interconnection Handbook, McGraw-Hill,

New York, 1991.

"High Power Transmitting Tubes for Broadcasting and Research," Philips Technical

Publication, Eindhoven, the Netherlands, 1988.

Iwase, N., and K. Shinozaki, "Aluminum Nitride Substrates Having High Thermal

Conductivity," Solid State Technol., pp. 135137, October 1986.

Jordan, Edward C., (ed.), Reference Data for Engineers: Radio, Electronics, Computer

and Communications, 7th Ed., Howard W. Sams, Indianapolis, IN, 1985.

Kingery, W. D., H. K. Bowen, and D. R. Uhlmann, Introduction to Ceramics, John

Wiley & Sons, New York, pp. 637, 1976.

Kohl, Walter, Materials Technology for Electron Tubes, Reinhold, New York.
Laboratory Staff, The Care and Feeding of Power Grid Tubes, Varian Eimac, San

Carlos, CA, 1984.

Lafferty, J. M., Vacuum Arcs, John Wiley & Sons, New York, 1980.
Mattox, D. M., and H. D. Smith, "The Role of Manganese in the Metallization of High

Alumina Ceramics," J. Am. Ceram. Soc., Vol. 64, pp. 13631369, 1985.

Mistler, R. E., D. J. Shanefield, and R. B. Runk, "Tape Casting of Ceramics," in G. Y.

Onoda, Jr., and L. L. Hench (eds.), Ceramic Processing Before Firing, John Wiley
& Sons, New York, pp. 411448, 1978.

Muller, J. J., "Cathode Excited Linear Amplifiers," Electrical Communications, Vol.

23, 1946.

Powers, M. B., "Potential Beryllium Exposure While Processing Beryllia Ceramics for

Electronics Applications," Brush Wellman, Cleveland, OH.

Reich, Herbert J., Theory and Application of Electronic Tubes, McGraw-Hill, New

York, 1939.

Roth, A., Vacuum Technology, 3rd Ed., Elsevier Science Publishers B. V., 1990.
Sawhill, H. T., A. L. Eustice, S. J. Horowitz, J. Gar-El, and A. R. Travis, "Low Temper-

ature Co-Fireable Ceramics with Co-Fired Resistors," in Proc. Int. Symp. on Micro-
electronics, pp. 173180, 1986.

Schwartz, B., "Ceramic Packaging of Integrated Circuits," in Electronic Ceramics, L.

M. Levinson (ed.), Marcel Dekker, New York, pp. 144, 1988.

Strong, C. E., "The Inverted Amplifier," Electrical Communications, Vol. 19, no. 3,

1941.

Whitaker, J. C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1991.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter 4 Designing Vacuum Tube Circuits"
Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

4

Designing Vacuum Tube Circuits

4.1

Introduction

Any number of configurations may be used to generate RF signals using vacuum
tubes. Circuit design is dictated primarily by the operating frequency, output power,
type of modulation, duty cycle, and available power supply. Tube circuits can be di-
vided generally by their operating class and type of modulation employed. As dis-
cussed in Section 2.1.4, the angle of plate current flow determines the class of opera-
tion:

·

Class A = 360° conduction angle

·

Class B = 180° conduction angle

·

Class C = conduction angle less than 180°

·

Class AB = conduction angle between 180 and 360°

The class of operation has nothing to do with whether the amplifier is grid-driven or
cathode-driven. A cathode-driven amplifier, for example, can be operated in any de-
sired class. The class of operation is only a function of the plate current conduction
angle. The efficiency of an amplifier is also a function of the plate current conduction
angle.

The efficiency of conversion of dc to RF power is one of the more important charac-

teristics of a vacuum tube amplifier circuit. The dc power that is not converted into use-
ful output energy is, for the most part, converted to heat. This heat represents wasted
power; the result of low efficiency is increased operating cost for energy. Low effi-
ciency also compounds itself. This wasted power must be dissipated, requiring in-
creased cooling capacity. The efficiency of the amplifier must, therefore, be carefully
considered, consistent with the other requirements of the system.

Figure 4.1

shows the

theoretical efficiency attainable with a tuned or resistive load assuming that the peak ac
plate voltage is equal to the plate supply voltage.

© 1999 CRC Press LLC

4.1.1 Class A Amplifier

A class A amplifier is used in applications requiring low harmonic distortion in the
output signal. A class A amplifier can be operated with low intermodulation distor-
tion in linear RF amplifier service. Typical plate efficiency for a class A amplifier is
about 30 percent. Power gain is high because of the low drive power required. Gains
as high as 30 dB are typical.

4.1.2 Class B and AB Amplifiers

A class AB power amplifier is capable of generating more power--using the same
tube--than the class A amplifier, but more intermodulation distortion also will be
generated. A class B RF linear amplifier will generate still more intermodulation dis-
tortion, but is acceptable in certain applications. The plate efficiency is typically 66
percent, and stage gain is about 20 to 25 dB.

4.1.3 Class C Amplifier

A class C power amplifier is used where large amounts of RF energy need to be gen-
erated with high efficiency. Class C RF amplifiers must be used in conjunction with
tuned circuits or cavities, which restore the amplified waveform through the flywheel
effect.

The grounded-cathode class C amplifier is the building block of RF technology. It is

the simplest method of amplifying CW, pulsed, and FM signals. The basic configura-
tion is shown in

Figure 4.2

. Tuned input and output circuits are used for impedance

Figure 4.1 Plate efficiency as a function of conduction angle for an amplifier with a
tuned load.

© 1999 CRC Press LLC

matching and to resonate the stage at the desired operating frequency. The cathode is
bypassed to ground using low-value capacitors. Bias is applied to the grid as shown.
The bias power supply may be eliminated if a self-bias configuration is used. The typi-
cal operating efficiency of a class C stage ranges from 65 to 85 percent.

Figure 4.3

illustrates the application of a zero-bias triode in a grounded-grid ar-

rangement. Because the grid operates at RF ground potential, this circuit offers stable
performance without the need for neutralization (at MF and below). The input signal is
coupled to the cathode through a matching network. The output of the triode feeds a pi
network through a blocking capacitor.

4.2

Principles of RF Power Amplification

In an RF power amplifier, a varying voltage is applied to the control grid (or cathode,
in the case of a grounded-grid circuit) from a driver stage whose output is usually one
of the following:

·

Carrier-frequency signal only

·

Modulation (intelligence) signal only

·

Modulated carrier signal

Simultaneous with the varying control grid signal, the plate voltage will vary in a sim-
ilar manner, resulting from the action of the amplified current flowing in the plate cir-
cuit. In RF applications with resonant circuits, these voltage changes are smooth sine

Figure 4.2 Basic grounded-cathode RF amplifier circuit.

© 1999 CRC Press LLC

wave variations, 180° out of phase with the input. The relationship is illustrated in

Figure 4.4

. Note how these variations center about the dc plate voltage and the dc con-

trol grid bias. In

Figure 4.5

, the variations have been indicated next to the plate volt-

age and grid voltage scales of a typical constant-current curve. At some instant in
time, shown as t on the time scales, the grid voltage has a value denoted e

g

on the grid

voltage sine wave.

Any point on the operating line (when drawn on constant-current curves as illus-

trated in

Figure 4.5

) tells the instantaneous values of plate current, screen current, and

grid current that must flow when these particular values of grid and plate voltage are ap-
plied to the tube. Thus, by plotting the values of plate and grid current as a function of
time t, it is possible to produce a curve of instantaneous values of plate and grid current.
Such plots are shown in

Figure 4.6

.

By analyzing the plate and grid current values, it is possible to predict with accuracy

the effect on the plate circuit of a change at the grid. It follows that if a properly loaded
resonant circuit is connected to the plate, a certain amount of RF power will be deliv-
ered to that circuit. If the resonant circuit is tuned to the fundamental frequency (the
same frequency as the RF grid voltage), the power delivered will be that of the funda-
mental or principal RF component of plate current. If the circuit is tuned to a harmonic

Figure 4.3 Typical amplifier circuit using a zero-bias triode. Grid current is measured in
the return lead from ground to the filament.

© 1999 CRC Press LLC

Figure 4.4 Variation of plate voltage as a function of grid voltage.

Figure 4.5 Relationship between grid voltage and plate voltage plotted on a con-
stant-current curve.

© 1999 CRC Press LLC

of the grid voltage frequency, the power delivered will be the result of a harmonic com-
ponent of the plate current.

4.2.1 Drive Power Requirements

The technical data sheet for a given tube type lists the approximate drive power re-
quired for various operating modes. As the frequency of operation increases and the
physical size of the tube structure becomes large with respect to this frequency, the
drive power requirements also will increase.

The drive power requirements of a typical grounded-cathode amplifier consist of six

major elements:

·

The power consumed by the bias source, given by:

P

I

E

c

c

1

1

1

=

×

(4.1)

·

The power dissipated in the grid as a result of rectified grid current:

P

I

e

c

cmp

2

1

=

×

(4.2)

·

The power consumed in the tuned grid circuit:

P

I

R

c rms

rf

3

2

=

×

,

(4.3)

·

The power lost as a result of transit-time effects:

P

e

R

c rms

t

4

2

=

,

(4.4)

Figure 4.6

Instantaneous

values of plate and grid cur-
rent as a function of time.

© 1999 CRC Press LLC

R

t

is that part of the resistive component of the tube input impedance resulting

from transit-time effects, and is given by:

R

Kg

f

T

t

m

=

1

2

2

(4.5)

·

The power consumed in that part of the resistive component of the input imped-
ance resulting from cathode lead inductance:

P

e

R

g

s

5

2

=

(4.6)

Input resistance resulting from the inductance of the cathode leads is found
from the following:

R

g L C

s

m

k

gk

=

1

2

(4.7)

·

The power dissipated in the tube envelope because of dielectric loss:

P

f E

6

1

2

1 41

= .

(4.8)

Where:
I

c1

= dc grid current

E

c1

= dc grid voltage

E

cmp

= maximum positive grid voltage

I

c,rms

= rms value of RF grid current

R

rf

= RF resistance of grid circuit

e

c,rms

= rms value of RF grid voltage

R

t

= resistance resulting from transit-time loading

K = a constant (function of the tube geometry)
g

m

= transconductance

f = frequency (Hz)
T = transit-time, cathode to grid
R

s

= cathode lead inductance input resistance loading

= 2 f

L

k

= cathode lead inductance (Henrys)

C

gk

= grid-to-cathode capacitance (Farads)

E

1

= voltage gradient (kilovolt per inch, rms)

= loss factor of dielectric materials

The total drive power P

t

is, then, equal to:

P

P

P

P

P

P

P

t

=

+

+

+

+

+

1

2

3

4

5

6

(4.9)

© 1999 CRC Press LLC

Particular attention must be given to grid dissipation when a tube is operated in the

VHF and UHF regions. The total driving power required for a given output may be
greater than the grid dissipation capability of the device.

Operational Considerations for VHF and UHF

For operation of a tube in the VHF and UHF regions, several techniques may be ap-
plied to minimize the driving power without appreciably affecting plate conversion ef-
ficiency. The most common techniques are:

·

Use the minimum dc control bias. Frequently, it is advisable to bring the bias
down to approximately cutoff.

·

Maintain a high value of dc screen voltage, even though it appears to increase the
fraction of the cycle during which plate current flows.

·

Use the minimum RF excitation voltage necessary to obtain the desired plate cir-
cuit performance, even though the dc grid current is considerably lower than
would be expected at lower frequencies.

·

Keep the cathode lead inductance to the output and input circuits as low as possi-
ble. This can be accomplished by 1) using short and wide straps, 2) using two sep-
arate return paths for the input and output circuits, or 3) properly choosing a cath-
ode bypass capacitor(s).

These techniques do not necessarily decrease the plate efficiency significantly

when the circuit is operated at VHF and UHF. The steps should be tried experimentally
to determine whether the plate circuit efficiency is appreciably affected. It is usually ac-
ceptable, and even preferable, to sacrifice some plate efficiency for improved tube life
when operating at VHF and UHF.

Optimum power output at these frequencies is obtained when the loading is greater

than would be used at lower frequencies. Fortunately, the same condition reduces driv-
ing power and screen current (for the tetrode and pentode), and improves tube life ex-
pectancy in the process.

4.2.2 Mechanical and Electrical Considerations

To maintain proper isolation of the output and input circuits, careful consideration
must be given to the location of the component parts of the amplifier. All elements of
the grid or input circuit and any earlier stages must be kept out of the plate circuit
compartment. Similarly, plate circuit elements must be kept out of the input compart-
ment. Note, however, that in the case of the tetrode and pentode, the screen lead of the
tube and connections via the socket are common to both the output and input resonant
circuits. Because of the plate-to-screen capacitance of a tetrode or pentode, the RF
plate voltage (developed in the output circuit) causes an RF current to flow out of the
screen lead to the chassis. In the case of a push-pull stage, this current may flow from
the screen terminal of one tube to the screen terminal of the other tube. Similarly, be-
cause of the grid-to-screen capacitance of the tube, the RF voltage in the input circuit

© 1999 CRC Press LLC

will cause an RF current to flow in this same screen lead to the chassis, or to the oppo-
site tube of the push-pull circuit.

The inductance of the lead common to both the output and input circuits has the de-

sirable feature of providing voltage of opposite polarity to neutralize the feedback volt-
age of the residual plate to control grid capacitance of the tube. (The properties of neu-
tralization are discussed in Section 3.4.) Note, however, that the mutual coupling from
the screen lead to the input resonant circuit may be a possible source of trouble, depend-
ing on the design.

Lead Lengths

The interconnecting lead wires close to the tube should be designed with low induc-
tance to minimize the generation of VHF parasitic oscillations. If two or more tubes
are used in a given circuit, they should be placed reasonably close together to facili-
tate short interconnecting leads. The lead lengths of HF circuits usually can be much
longer; the length depends, to a large extent, upon the frequency of the fundamental.
All of the dc, keying, modulation, and control circuit wires can be relatively long if
properly filtered and arranged physically apart from the active RF circuits. The fol-
lowing interconnecting lead wires in a tetrode or pentode power amplifier should have
low inductance:

·

Filament and screen bypass leads

·

Suppressor bypass leads

·

Leads from the grid and the plate to the tuning capacitor of the RF circuit (and re-
turn)

·

Interconnections from tube to tube in push-pull or parallel arrangements (except
for parasitic suppressors in the plate)

For a lead to have low inductance, it must have a large surface and be short in length,
such as a strap or a ribbon. This consideration also applies to that portion of a lead in-
side a bypass capacitor.

Power Supply Considerations

The power supply requirements for a triode are straightforward. The degree of regula-
tion and ripple depends upon the requirements of the system. In the case of a linear
RF amplifier, it is important to have good plate power supply regulation. Without
tight regulation, the plate voltage will drop during the time the plate is conducting
current heavily. This drop will cause flat topping and will appear as distortion in the
output. In push-pull applications where grid current flows, it is important to keep the
grid circuit resistance to a minimum. If this is not done, positive peak clipping will
occur.

In the case of the tetrode and pentode, the need for screen voltage introduces some

new considerations and provides some new operating possibilities. Voltage for the
screen grid of a low-power tetrode or pentode can readily be taken from the power sup-

© 1999 CRC Press LLC

ply used for the plate of the tube. In this case, a series resistor or potential-dividing re-
sistor is chosen so that, with the intended screen current flowing, the voltage drop
through the resistor is adequate to give the desired screen voltage. The potential-divid-
ing resistor is the preferred technique for those tubes with significant secondary screen
emission.

For high-power tubes, screen voltage commonly is taken from a separate power sup-

ply. A combination scheme also may be employed, where a dropping resistor is used in
conjunction with a low-voltage or intermediate-voltage supply. Often, a combination
of series resistor and voltage source can be chosen so that the rated screen dissipation
will not be exceeded, regardless of variations in screen current. With a fixed screen sup-
ply, there are advantages in using an appreciable amount of fixed grid bias so as to pro-
vide protection against the loss of excitation, or for cases where the driver stage is being
keyed.

If the screen voltage is taken through a dropping resistor from the plate supply, there

is usually little point in using a fixed grid bias because an unreasonable amount of bias
would be required to protect the tube if the excitation failed. Under operating condi-
tions with normal screen voltage, the cutoff bias is low (screen voltage divided by the
screen

µ). When a stage loses excitation and runs statically, the screen current falls to

nearly zero. (See the static curves of the tube in question.) If the screen voltage is ob-
tained through a simple dropping resistor from the plate supply, the screen voltage will
then rise close to full plate voltage. Because the cutoff bias required is proportional to
screen voltage, the grid bias required will be much greater than the amount of bias de-
sired under normal operating conditions. When a screen dropping resistor is used, most
of the bias normally is supplied through a grid resistor, and other means are used for
tube protection.

The power output from a tetrode or pentode is sensitive to screen voltage. For this

reason, any application requiring a high degree of linearity through the amplifier re-
quires a well-regulated screen power supply. A screen dropping resistor from the plate
supply is not recommended in such cases.

The suppressor grid power supply requirements are similar to those of the control

grid power supply. The suppressor grid intercepts little current, so a low-power supply
may be used. Any variation in suppressor voltage as a result of ripple or lack of regula-
tion will appear at the output of the amplifier because of suppressor grid modulation of
the plate current.

4.2.3 Bypassing Tube Elements

Operation at high frequencies requires attention to bypassing of the tube elements.
Areas of concern include:

·

Filament circuit

·

Screen grid circuit

·

Suppressor grid circuit

© 1999 CRC Press LLC

Filament Bypassing

Low-inductance bypass capacitors should be used at the filament in an RF amplifier.
Good engineering practice calls for placement of the capacitor directly between the
filament socket terminals. If the circuit design allows it, strap one filament directly to
the chassis; if not, use a second bypass capacitor from one terminal to the chassis.

If two or more tubes are used in a push-pull or parallel circuit, a short strap intercon-

necting one of the filament terminals of each socket can be used. The midpoint of the
interconnecting strap is then bypassed or grounded directly.

For tubes equipped with an isolating screen cone terminal, the general circuit ar-

rangement is usually different. The filament or cathode should go directly, or through
bypass capacitors, to the cavity wall or chassis to which the screen terminal is bypassed.

Screen and Suppressor Grid Bypassing

Low-inductance leads are generally advisable for screen and suppressor grid terminal
connections. For all frequencies, good engineering practice calls for routing of the
screen bypass capacitors directly from the screen to one filament terminal. The sup-
pressor grid is bypassed in the same manner when the suppressor is operated with a
potential other than cathode potential. With the suppressor operating at cathode po-
tential, the suppressor should be grounded to the chassis directly in a circuit where the
cathode is at chassis potential. This applies to tubes in push-pull as well as tubes in
single-ended stages.

In the VHF region, the connection to the screen terminals--for those tubes with two

screen pins--should be made to the midpoint of a strap placed between the two screen
terminals of the socket. This arrangement provides for equal division of the RF currents
in the screen leads and minimizes heating effects.

At operating frequencies above the self-neutralizing frequency of the tetrode or

pentode being used, the screen bypass capacitors are sometimes variable. By proper ad-
justment of this variable capacitor, the amount and phase of the screen RF voltage can
be made to cancel the effects of the feedback capacitance within the tube. Thus, neutral-
ization is accomplished. The screen lead inductance and the variable capacitor are not
series resonant. The variable capacitor is adjusted so that a net inductive reactance re-
mains to provide the proper voltage and phase for neutralization.

The preceding guidelines apply directly to tubes having the screen and suppressor

grids mounted on internal supporting lead rods. Tube types having isolating screen
cone terminals tend to work best when the screen or suppressor bypass capacitor is a flat
sandwich type built directly onto the peripheral screen-contacting collet of the socket.
The size of the bypass capacitor is a function of the operating frequency. The dielectric
material may be of Teflon, mica, isomica, or a similar material. (Teflon is a trademark of
DuPont.)

Application Examples

The use of bypassing capacitors for a grounded-cathode RF amplifier is illustrated in

Figure 4.7

. Bypassing for a grounded-grid amplifier is shown in

Figure 4.8

. The con-

© 1999 CRC Press LLC

trol grid may be operated at dc ground potential, shown in

Figure 4.8a

, or bypassed

directly at the socket, shown in

Figure 4.8b

. For the sake of simplicity, operating the

grid at dc ground potential is preferred. Functionally, these two circuits are identical.
They vary only in the method used to measure grid current. In

Figure 4.8a

, grid cur-

rent is measured in the return lead from ground to filament. In

Figure 4.8b

, the grid is

raised 1

above dc ground to allow grid current to be measured.

4.2.4 Parasitic Oscillations

Most self-oscillations in RF power amplifiers using gridded tubes have been found to
fall within the following three classes:

·

Oscillation at VHF from about 40 to 200 MHz, regardless of the operating fre-
quency of the amplifier

·

Self-oscillation on the fundamental frequency of the amplifier

·

Oscillation at a low radio frequency below the normal frequency of the amplifier

Low-frequency oscillation in an amplifier usually involves the RF chokes, especially
when chokes are used in both the output and input circuits. Oscillation near the funda-
mental frequency involves the normal resonant circuits and brings up the question of
neutralization. (Neutralization is discussed in Section 3.4.) When a parasitic self-os-

Figure 4.7 Schematic of a tetrode RF power amplifier showing tube element bypass ca-
pacitors.

© 1999 CRC Press LLC

Figure 4.8 Schematic of a triode RF power amplifier showing tube element bypass ca-
pacitors: (

a) control grid at dc ground potential, (b) control grid bypassed to ground.

(

a)

(

b)

© 1999 CRC Press LLC

cillation is found on a very high frequency, the interconnecting leads of the tube, tun-
ing capacitor, and bypass capacitors typically are involved.

VHF oscillation occurs commonly in amplifiers where the RF circuits consist of

coils and capacitors, as opposed to cavity designs. As illustrated in

Figure 4.9

, the tube

capacitances effectively form a tuned-plate, tuned-grid oscillator.

The frequency of a VHF parasitic typically is well above the self-neutralizing fre-

quency of the tube. However, if the self-neutralizing frequency of the device can be in-
creased and the frequency of the parasitic lowered, complete suppression of the para-
sitic may result, or its suppression by resistor-inductor parasitic suppressors may be
made easier.

It is possible to predict, with the use of a grid dip wavemeter, the parasitic frequency

to be expected in a given circuit. The circuit should be complete and have no voltages
applied to the tube. Couple the meter to the plate or screen lead, and determine the reso-
nant frequency.

Elimination of the VHF parasitic oscillation may be accomplished using one of the

following techniques:

·

Place a small coil and resistor combination in the plate lead between the plate of
the tube and the tank circuit, as shown in

Figure 4.10a

. The resistor-coil combina-

tion usually is made up of a noninductive resistor of about 25 to 100

, shunted by

three or four turns of approximately 1/2 in diameter and frequently wound around
the resistor. In some cases, it may be necessary to use such a suppressor in both
the plate and grid leads. The resistor-coil combination operates on the principle
that the resistor loads the VHF circuit but is shunted by the coil for the lower fun-
damental frequency. In the process of adjusting the resistor-coil combination, it is

Figure 4.9

Interelectrode capacitances supporting VHF parasitic oscillation in a

high-frequency RF amplfier.

© 1999 CRC Press LLC

often found that the resistor runs hot. This heat usually is caused by the dissipa-
tion of fundamental power in the resistor, and it is an indication of too many turns
in the suppressor coil. Use only enough turns to suppress the parasitic, and no
more. Once the parasitic has been suppressed, there will be no parasitic voltage or
current present. Therefore, there is no parasitic power to be dissipated.

Figure 4.10 Placement of parasitic suppressors to eliminate VHF parasitic oscillations
in a high-frequency RF amplifier: (

a) resistor-coil combination, (b) parasitic choke.

(

a)

(

b)

© 1999 CRC Press LLC

·

Use small parasitic chokes in the plate lead, as shown in

Figure 4.10b

. The size of

the coil will vary considerably depending upon the tube and circuit layout. A coil
of four to ten turns with a diameter of approximately 1/2 in is typical. The pres-
ence of the choke in the frequency-determining part of the circuit lowers the fre-
quency of a possible VHF parasitic so that it falls near the self-neutralizing fre-
quency of the tube and bypass leads. In addition to variation in the size of the sup-
pressor choke, the amount of inductance common to the screen and filament in
the filament grounding strap may be a factor. This parameter can be varied simul-
taneously with the suppressor choke.

Of the two methods outlined for suppressing VHF parasitic oscillations, the first is
probably the simpler and is widely employed.

Dynatron Oscillation

Another form of commonly encountered self-oscillation is known as dynatron oscil-
lation, caused when an electrode in a vacuum tube has negative resistance. At times,
there may be more electrons leaving the screen grid than arriving. If the screen volt-
age is allowed to increase under these conditions, even more electrons will leave the
grid. This phenomenon implies a negative resistance characteristic. If there is high al-
ternating current impedance in the circuit from the screen grid through the screen grid
power supply, and from the plate power supply to the plate, dynatron oscillation may
be sustained.

Dynatron oscillation typically occurs in the region of 1 to 20 Hz. This low-frequency

oscillation usually is accompanied by another oscillation in the 1 to 2 kHz region. Sup-
pression of these oscillations can be accomplished by placing a large bypass capacitor
(1000

µF) across the output of the screen grid power supply. The circuit supporting the

oscillation also can be detuned by a large inductor. Increasing the circuit losses at the
frequency of oscillation is also effective.

Harmonic Energy

It is generally not appreciated that the pulse of grid current contains energy on har-
monic frequencies and that control of these harmonic energies may be important. The
ability of the tetrode and pentode to isolate the output circuit from the input over a
wide range of frequencies is significant in avoiding feedthrough of harmonic voltages
from the grid circuit. Properly designed tetrode and pentode amplifiers provide for
complete shielding in the amplifier layout so that coupling external to the tube is pre-
vented.

In RF amplifiers operating either on the fundamental or on a desired harmonic fre-

quency, the control of unwanted harmonics is important. The following steps permit re-
duction of the unwanted harmonic energies present in the output circuit:

·

Keep the circuit impedance between the plate and cathode low for the high har-
monic frequencies. This requirement may be achieved by having some or all of
the tuning capacitance of the resonant circuit close to the tube.

© 1999 CRC Press LLC

·

Completely shield the input and output compartments.

·

Use inductive output coupling from the resonant plate circuit and possibly a ca-
pacitive or Faraday shield between the coupling coil and the tank coil, or a
high-frequency attenuating circuit such as a pi or pi-L network.

·

Use low-pass filters on all supply leads and wires coming into the output and in-
put compartments.

·

Use resonant traps for particular frequencies.

·

Use a low-pass filter in series with the output transmission line.

4.2.5 Shielding

In an RF amplifier, shielding between the input and output circuits must be consid-
ered. Triode amplifiers are more tolerant of poor shielding because power gain is rela-
tively low. If the circuit layout is reasonable and no inductive coupling is allowed to
exist, a triode amplifier usually can be built without extensive shielding. Even if
shielding is not necessary to prevent fundamental-frequency oscillation, it will aid in
eliminating any tendency toward parasitic oscillation. The higher the gain of an am-
plifier, the more important the shielding.

Pierced Shields

Tetrode and pentode power amplifiers require comprehensive shielding to prevent in-
put-to-output circuit coupling. It is advisable to use nonmagnetic materials such as
copper, aluminum, or brass in the RF fields to provide the shielding. Quite often, a
shield must have holes through it to allow the passage of cooling air. In the LF and
part of the HF range, the presence of small holes will not impair shield effectiveness.
As the frequency is increased, however, the RF currents flowing around the hole in
one compartment cause fields to pass through the hole into another compartment.
Currents, therefore, are induced on the shield in the second compartment. This prob-
lem can be eliminated by using holes that have significant length. A piece of pipe with
a favorable length-to-diameter ratio as compared to the frequency of operation will
act as a waveguide-beyond-cutoff attenuator.

If more than one hole is required to pass air, a material resembling a honeycomb may

be used. This material is available commercially and provides excellent isolation with a
minimum of air pressure drop. A section of honeycomb shielding is shown in

Figure

4.11

. Some tube sockets incorporate a waveguide-beyond-cutoff air path. These sock-

ets allow the tube in the amplifier to operate at high gain and up through VHF.

Metal Base Shells and Submounted Sockets

Some tetrodes and pentodes are supplied with metal base shells. The shell is typically
grounded by the clips provided with the socket. This completes the shielding between

© 1999 CRC Press LLC

the output and input circuits because the base shell of the tube comes up opposite the
screen shield within the tube itself.

Some pentodes use this metal base shell as the terminal for the suppressor grid. If the

suppressor is to be at some potential other than ground, then the base shell must not be
dc grounded. The base shell is bypassed to ground for RF and insulated from ground for
dc.

There is a family of tetrodes and pentodes without the metal base shell. For this type

of tube structure, it is good practice to submount the socket so that the internal screen
shield is at the same level as the chassis deck. This technique will improve the in-
put-to-output circuit shielding. In submounting a tube, it is important that adequate
clearance be provided around the base of the device for the passage of cooling air.

Compartments

By placing the tube and related circuits in completely enclosed compartments and
properly filtering incoming supply wires, it is possible to prevent coupling of RF en-
ergy out of the circuit by means other than the desired output coupling. Such filtering
prevents the coupling of energy that may be radiated or fed back to the input section
or to earlier stages in the amplifier chain. Energy fed back to the input circuit causes
undesirable interaction in tuning and/or self-oscillation. If energy is fed back to ear-

Figure 4.11 A section of honeycomb shielding material used in an RF amplifier.

© 1999 CRC Press LLC

lier stages, significant operational problems may result because of the large power
gain over several stages.

In the design of an RF amplifier, doors or removable panels typically must be used.

The requirement for making a good, low-resistance joint at the discontinuity must be
met. There are several materials available commercially for this purpose. Fingerstock,
shown in

Figure 4.12

, has been used for many years. Teknit

1

is also a practical solution.

Sometimes, after the wiring has been completed, further shielding of a particular con-
ductor is required. Various types of shielding tapes

2

can be wound on a conductor as a

temporary or permanent solution.

4.2.6 Protection Measures

Power grid tubes are designed to withstand considerable abuse. The maximum ratings
for most devices are conservative. For example, the excess anode dissipation resulting
from detuning the plate circuit will have no ill effects on most tubes if it is not applied
long enough to overheat the envelope and the seal structure.

Similarly, the control, screen, and suppressor grids will stand some excess dissipa-

tion. Typically, the maximum dissipation for each grid indicated on the data sheet
should not be exceeded except for time intervals of less than 1 s. The maximum dissipa-
tion rating for each grid structure is usually considerably above typical values used for
maximum output so that ample operating reserve is provided. The time of duration of
overload on a grid structure is necessarily short because of the small heat-storage ca-
pacity of the grid wires. Furthermore, grid temperatures cannot be measured or seen, so
no warning of accidental overload is apparent.

Figure 4.12 A selection of common fingerstock.

1

Technical Wire Products, Inc., Cranford, NJ

2

Perfection Mica Co., Magnetic Shield Division, Chicago, IL

© 1999 CRC Press LLC

The type and degree of protection required in an RF amplifier against circuit failure

varies with the type of screen and grid voltage supply.

Table 4.1

lists protection criteria

for tetrode and pentode devices. The table provides guidelines on the location of a suit-
able relay that should act to remove the principal supply voltage from the stage or trans-
mitter to prevent damage to the tube.

For designs where screen voltage is taken through a dropping resistor from the plate

supply, a plate relay provides almost universal protection. For a fixed screen supply, a
relay provides protection in most cases. For protection against excessive antenna load-
ing and subsequent high plate dissipation, a screen undercurrent relay also may be used
in some services.

The plate, screen, and bias voltages may be applied simultaneously to a tetrode. The

same holds true for a pentode, plus the application of the suppressor voltage. In a
grid-driven amplifier, grid bias and excitation usually can be applied alone to the tube,
especially if a grid leak resistor is used. Plate voltage can be applied to the tetrode and
pentode before the screen voltage with or without excitation to the control grid. Never
apply screen voltage before the plate voltage. The only exceptions are when the tube is
cut off so that no space current (screen or plate current) will flow, or when the excitation
and screen voltage are low. If screen voltage is applied before the plate voltage and
screen current can flow, the maximum allowable screen dissipation will almost always
be exceeded, and tube damage will result.

Table 4.2

lists protection guidelines for a triode. The table covers the grid-driven tri-

ode amplifier and the high-

µ(zero-bias) cathode-driven triode amplifier. Drive voltage

must never be applied to a zero-bias triode amplifier without plate voltage. The table in-
dicates the recommended location of a suitable relay that should act to remove the prin-

Circuit Failure

Type

Fixed Screen Supply

Screen Voltage Through

Dropping Resistor

Fixed Grid

Bias

Resistor Grid

Bias

Fixed Grid

Bias

Resistor Grid

Bias

Loss of excita-
tion

No protection
required

Plate current
relay

Plate current
relay

Plate current
relay or screen
control circuit

Loss of an-
tenna loading

Screen current
relay

Screen current
relay

Grid current re-
lay

No protection
required

Excess an-
tenna loading

Screen under-
current relay

Screen under-
current relay

Plate current
relay

Plate current
relay

Failure of plate
supply

Screen current
relay

Screen current
relay

Grid current re-
lay

No protection
required

Failure of
screen supply

Grid current re-
lay

No protection
required

Does not apply Does not apply

Failure of grid
bias supply

Plate current
relay or screen
current relay

Does not apply Plate current

relay or grid
current relay

Does not apply

Table 4.1 Protection Guidelines for Tetrode and Pentode Devices

© 1999 CRC Press LLC

cipal supply voltage from the stage or transmitter to prevent damage to the tube or trans-
mitter.

4.3

Cavity Amplifier Systems

Power grid tubes are ideally suited for use as the power generating element in a cavity
amplifier. Because of the physical dimensions involved, cavity designs typically are
limited to VHF and above. Lower-frequency RF generators utilize discrete L and C
devices to create the needed tank circuit. Two types of cavity amplifiers commonly
are used: 1/4-wavelength and 1/2-wavelength systems.

In a cavity amplifier, the tube becomes part of a resonant transmission line. The

stray interelectrode and distributed capacity and inductance of the tube are used to ad-
vantage as part of the resonant line. This resonant line is physically larger than the
equivalent lumped constant LRC resonant circuit operating at the same frequency, and
this larger physical size aids in solving the challenges of high-power operation, skin ef-
fect losses, and high-voltage-standoff concerns.

A shorted 1/4-wavelength transmission line has a high, purely resistive input imped-

ance. Electrically, it appears as a parallel resonant circuit, as shown in

Figure 4.13

.

When the physical length of the line is less than wavelength, the impedance will be
lower and the line will appear inductive, as illustrated in

Figure 4.14

. This inductance is

used to resonate with the capacitive reactance in the tube and the surrounding circuit.

4.3.1 Bandwidth and Efficiency

Power amplifier bandwidth has a significant effect on modulation performance.
Available bandwidth determines the amplitude response, phase response, and
group-delay response. Performance tradeoffs often must be considered in the design
of a cavity amplifier, including bandwidth, gain, and efficiency.

Circuit Failure

Type

Triode

Zero-Bias Triode

Fixed Grid Bias

Resistor Grid Bias

Loss of excitation

No protection re-
quired

Plate overcurrent
relay

No protection re-
quired

Loss of antenna
loading

RF output detector
and relay

RF output detector
and relay

Grid overcurrent re-
lay

Excess antenna
loading

RF output detector
and relay

RF output detector
and relay

RF output detector
and relay

Failure of plate sup-
ply

No protection re-
quired

No protection re-
quired

Grid overcurrent re-
lay

Failure of grid bias
supply

Plate overcurrent
relay

Does not apply

Does not apply

Table 4.2 Protection Guidelines for a Triode

© 1999 CRC Press LLC

Power amplifier bandwidth is restricted by the equivalent load resistance across the

parallel tuned circuits in the stage. Tuned circuits are necessary to cancel low reactive
impedance presented by the relatively high input and output capacitances of the ampli-
fying device. The bandwidth for a single tuned circuit is proportional to the ratio of ca-
pacitive reactance X

c

to load resistance R

l

appearing across the tuned circuit:

( )

BW

K

f C R

K X

R

c

l

c

l

2

(4.10)

Where:
BW = bandwidth between half-power (- 3 dB) points
K = proportionality constant
R

l

= load resistance appearing across tuned circuit

C = total capacitance of tuned circuit (includes stray capacitances and output or in-

put capacitances of the tube)

X

c

= capacitive reactance of C

f

c

= carrier frequency

Figure 4.13 Shorted 1/4-wavelength line: (

a) physical circuit, (b) electrical equivalent.

Figure 4.14 Shorted line less than 1/4-wavelength: (

a) physical circuit, (b) electrical

equivalent.

(

a)

(

b)

(

a)

(

b)

© 1999 CRC Press LLC

The RF voltage swing across the tuned circuit also depends on the load resistance.

For the same power and efficiency, the bandwidth can be increased if the capacitance is
reduced.

The efficiency of a cavity PA depends primarily on the RF plate voltage swing, the

plate current conduction angle, and the cavity efficiency. Cavity efficiency is related to
the ratio of loaded to unloaded Q as follows:

N

Q

Q

l

u

= -

×

1

100

(4.11)

Where:
N = cavity efficiency in percent
Q

l

= cavity loaded Q

Q

u

= cavity unloaded Q

The loaded Q is determined by the plate load impedance and output circuit capaci-

tance. Unloaded Q is determined by the cavity volume and the RF resistivity of the con-
ductors resulting from the skin effect. (Skin effect is discussed in Chapter 7.) For best
cavity efficiency, the following conditions are desirable:

·

High unloaded Q

·

Low loaded Q

As the loaded Q increases, the bandwidth decreases. For a given tube output capaci-
tance and power level, loaded Q decreases with decreasing plate voltage or with in-
creasing plate current. The increase in bandwidth at reduced plate voltage occurs be-
cause the load resistance is directly related to the RF voltage swing on the tube anode.
For the same power and efficiency, the bandwidth also can be increased if the output
capacitance is reduced. Power tube selection and minimization of stray capacitance
are important considerations when designing for maximum bandwidth.

4.3.2 Current Paths

The operation of a cavity amplifier is an extension of the current paths inside the tube.
Two elements must be examined in this discussion:

·

The input circulating currents

·

The output circulating currents

Input Circuit

The grid/cathode assembly resembles a transmission line whose termination is the RF
resistance of the electron stream within the tube.

Figure 4.15

shows the current path

of an RF generator (the RF driver stage output) feeding a signal into the grid/cathode
circuit.

© 1999 CRC Press LLC

The outer contact ring of the cathode heater assembly makes up the inner conductor

of a transmission line formed by the cathode and control grid assemblies. The filament
wires are returned down the center of the cathode. For the input circuit to work cor-
rectly, the cathode must have a low RF impedance to ground. This cathode bypassing
may be accomplished in one of several ways.

Below 30 MHz, the cathode can be grounded to RF voltages by simply bypassing the

filament connections with capacitors, as shown in

Figure 4.16a

. Above 30 MHz, this

technique does not work well because of the stray inductance of the filament leads. No-
tice that in (b), the filament leads appear as RF chokes, preventing the cathode from be-
ing placed at RF ground potential. This causes negative feedback and reduces the effi-
ciency of the input and output circuits.

In

Figure 4.16c

, the cathode circuit is configured to simulate a 1/2-wave transmis-

sion line. The line is bypassed to ground with large-value capacitors 1/2 wavelength
from the center of the filament (at the filament voltage feed point). This transmission
line RF short circuit is repeated 1/2 wavelength away at the cathode (heater assembly),
effectively placing it at ground potential.

Because 1/2-wavelength bypassing is usually bulky at VHF (and may be expensive),

RF generators often are designed using certain values of inductance and capacitance in
the filament/cathode circuit to create an artificial transmission line that will simulate a
1/2-wavelength shorted transmission line. As illustrated in the figure, the inductance
and capacitance of the filament circuit can resemble an artificial transmission line of
1/2 wavelength, if the values of L and C are properly selected.

Figure 4.15 A simplified representation of the grid input circuit of a PA tube.

© 1999 CRC Press LLC

Output Circuit

The plate-to-screen circulating current of the tetrode is shown in

Figure 4.17

. For the

purposes of example, consider that the output RF current is generated by an imagi-
nary current generator located between the plate and screen grid. The RF current trav-
els along the inside surface of the plate structure (because of the skin effect), through
the ceramic at the lower half of the anode contact ring, across the bottom of the fins,
and to the band around the outside of the fins. The RF current then flows through the
plate bypass capacitor to the RF tuned circuit and load, and returns to the screen grid.

The return current travels through the screen bypass capacitor (if used) and screen

contact ring, up the screen base cone to the screen grid, and back to the imaginary gen-
erator.

The screen grid has RF current returning to it, but because of the assembly's low im-

pedance, the screen grid is effectively at RF ground potential. The RF current genera-
tor, therefore, appears to be feeding an open-ended transmission line consisting of the
anode (plate) assembly and the screen assembly. The RF voltage developed by the an-
ode is determined by the plate impedance (Z

p

) presented to the anode by the resonant

circuit and its load.

Figure 4.16 Three common methods for RF bypassing of the cathode of a tetrode PA
tube: (

a) grounding the cathode below 30 MHz, (b) grounding the cathode above 30

MHz, (

c) grounding the cathode via a 1/2-wave transmission line.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

When current flows on one conductor of a transmission line cavity circuit, an

equal-magnitude current flows in the opposite direction on the other conductor. This
means that a large value of RF circulating current is flowing in the cavity amplifier
outer conductor (the cavity box). All of the outer conductor circulating currents start at
and return to the screen grid (in a tetrode-based system). The front or back access panel
(door) of the cavity is part of the outer conductor, and large values of circulating current
flow into it, through it, and out of it. A mesh contact strap generally is used to electri-
cally connect the access panel to the rest of the cavity.

4.3.3 The 1/4-Wavelength Cavity

The 1/4-wavelength PA cavity is common in transmitting equipment. The design is
simple and straightforward. A number of variations can be found in different RF gen-
erators, but the underlying theory of operation is the same.

A typical 1/4-wave cavity is shown in

Figure 4.18

. The plate of the tube connects di-

rectly to the inner section (tube) of the plate-blocking capacitor. The exhaust chim-
ney/inner conductor forms the other plate of the blocking capacitor. The cavity walls
form the outer conductor of the 1/4-wave transmission line circuit. The dc plate voltage
is applied to the PA tube by a cable routed inside the exhaust chimney and inner tube
conductor. In the design shown in the figure, the screen-contact fingerstock ring
mounts on a metal plate that is insulated from the grounded cavity deck by an insulating
material. This hardware makes up the screen-blocking capacitor assembly. The dc

Figure 4.17 RF circulating current path between the plate and screen in a tetrode PA
tube.

© 1999 CRC Press LLC

screen voltage feeds to the fingerstock ring from underneath the cavity deck through an
insulated feedthrough assembly.

A grounded screen configuration also may be used in this design in which the

screen-contact fingerstock ring is connected directly to the grounded cavity deck. The
PA cathode then operates at below ground potential (in other words, at a negative volt-
age), establishing the required screen voltage to the tube.

The cavity design shown in the figure is set up to be slightly shorter than a full 1/4

wavelength at the operating frequency. This makes the load inductive and resonates the
tube's output capacity. Thus, the physically foreshortened shorted transmission line is
resonated and electrically lengthened to 1/4 wavelength.

Figure 4.19

illustrates the paths taken by the RF circulating currents in the circuit.

RF energy flows from the plate, through the plate-blocking capacitor, along the inside
surface of the chimney/inner conductor (because of the skin effect), across the top of
the cavity, down the inside surface of the cavity box, across the cavity deck, through the
screen-blocking capacitor, over the screen-contact fingerstock, and into the screen
grid.

Figure 4.20

shows a graph of RF current, voltage, and impedance for a 1/4-wave-

length coaxial transmission line. The plot shows that infinite impedance, zero RF cur-
rent, and maximum RF voltage occur at the feed point. This would not be suitable for a

Figure 4.18 The physical layout of a common type of 1/4-wavelength PA cavity de-
signed for VHF operation.

© 1999 CRC Press LLC

Figure 4.19 The RF circulating current paths for the 1/4-wavelength cavity shown in

Figure 4.18

.

Figure 4.20 Graph of the RF current, RF voltage, and RF impedance for a 1/4-wave-
length shorted transmission line. At the feed point, RF current is zero, RF voltage is
maximum, and RF impedance is infinite.

© 1999 CRC Press LLC

practical PA circuit because arcing would result from the high RF voltage, and poor ef-
ficiency would be caused by the mismatch between the tube and the load.

Notice, however, the point on the graph marked at slightly less than 1/4 wavelength.

This length yields an impedance of 600 to 800

and is ideal for the PA plate circuit. It is

necessary, therefore, to physically foreshorten the shorted coaxial transmission line
cavity to provide the correct plate impedance. Shortening the line also is a requirement
for resonating the tube's output capacity, because the capacity shunts the transmission
line and electrically lengthens it.

Figure 4.21

shows a graph of the RF current, voltage, and impedance presented to

the plate of the tube as a result of the physically foreshortened line. This plate imped-
ance is now closer to the ideal 600 to 800

value required by the tube anode.

Tuning the Cavity

Coarse-tuning of the cavity is accomplished by adjusting the cavity length. The top of
the cavity (the cavity shorting deck) is fastened by screws or clamps and can be raised
or lowered to set the length of the cavity for the particular operating frequency.
Fine-tuning is accomplished by a variable-capacity plate-tuning control built into the
cavity. In a typical design, one plate of the tuning capacitor--the stationary plate--is
fastened to the inner conductor just above the plate-blocking capacitor. The movable

Figure 4.21 Graph of the RF current, RF voltage, and RF impedance produced by the
physically foreshortened coaxial transmission line cavity.

© 1999 CRC Press LLC

tuning plate is fastened to the cavity box--the outer conductor--and mechanically
linked to the front-panel tuning control. This capacity shunts the inner conductor to
the outer conductor and is used to vary the electrical length and resonant frequency of
the cavity.

The 1/4-wavelength cavity is inductively coupled to the output port. This coupling is

usually on the side opposite the cavity access door. The inductive pickup loop can take
one of several forms. In one design it consists of a half-loop of flat copper bar stock that
terminates in the loading capacitor at one end and feeds the output transmission line in-
ner conductor at the other end. This configuration is shown in

Figure 4.22

. The induc-

tive pickup ideally would be placed at the maximum current point in the 1/4-wave-
length cavity. However, this point is located at the cavity shorting deck, and, when the
deck is moved for coarse-tuning, the magnetic coupling will be changed. A compro-
mise in positioning, therefore, must be made. The use of a broad, flat copper bar for the
coupling loop adds some capacitive coupling to augment the reduced magnetic cou-
pling.

Adjustment of the loading capacitor couples the 50

transmission line impedance

to the impedance of the cavity. Heavy loading lowers the plate impedance presented to
the tube by the cavity. Light loading reflects a higher load impedance to the amplifier
plate.

Methods commonly used to increase the operating bandwidth of the cavity include

minimizing added tuning capacitance. The ideal case would be to resonate the plate ca-

Figure 4.22 Basic loading mechanism for a VHF 1/4-wave cavity.

© 1999 CRC Press LLC

pacitance alone with a "perfect" inductor. Practical cavities, however, require either the
addition of a variable capacitor or a variable inductor using sliding contacts for tuning.
Other inherent mechanical and electrical compromises include:

·

The requirement for a plate dc-blocking capacitor.

·

The presence of maximum RF current at the grounded end of the simulated trans-
mission line where the conductor may be nonhomogeneous. This can result in ac-
companying losses in the contact resistance.

4.3.4 The 1/2-Wavelength Cavity

The operation of a 1/2-wavelength cavity follows the same basic idea as the 1/4-wave-
length design outlined previously. The actual construction of the system depends
upon the power level used and the required bandwidth. An example will help to illus-
trate how a 1/2-wavelength cavity operates.

The design of a basic 1/2-wavelength PA cavity for operation in the region of 100

MHz is shown in

Figure 4.23

. The tube anode and a silver-plated brass pipe serve as the

inner conductor of the 1/2-wave transmission line. The cavity box serves as the outer
conductor. The transmission line is open at the far end and repeats this condition at the
plate of the tube. The line is, in effect, a parallel resonant circuit for the PA tube.

The physical height of the circuit shown (67 in) was calculated for operation down to

88 MHz. To allow adequate clearance at the top of the transmission line and space for

Figure 4.23 The 1/2-wavelength PA cavity in its basic form.

© 1999 CRC Press LLC

input circuitry at the bottom of the assembly, the complete cavity box would have to be
almost 8 ft tall. This is too large for any practical transmitter.

Figure 4.24

shows RF voltage, current, and impedance for the inner conductor of the

transmission line and the anode of the tube. The load impedance at the plate is thou-
sands of ohms. Therefore, the RF current is extremely small, and the RF voltage is ex-
tremely large. In the application of such a circuit, arcing would become a problem. The
high plate impedance also would make amplifier operation inefficient.

The figure also shows an area between the anode and the 1/4-wavelength location

where the impedance of the circuit is 600 to 800

. As noted previously, this value is

ideal for the anode of the PA tube. To achieve this plate impedance, the inner conductor
must be less than a full 1/2 wavelength. The physically foreshortened transmission-line
circuit must, however, be electrically resonated (lengthened) to 1/2 wavelength for
proper operation.

If the line length were changed to operate at a different frequency, the plate imped-

ance also would change because of the new distribution of RF voltage and current on
the new length of line. The problem of frequency change, therefore, is twofold: The
length of the line must be adjusted for resonance, and the plate impedance of the tube
must be kept constant for good efficiency.

To accommodate operation of this system at different frequencies, while keeping

the plate impedance constant, two forms of coarse-tuning and one form of fine-tuning
are built into the 1/2-wave PA cavity.

Figure 4.25

shows the tube and its plate line (inner

Figure 4.24 The distribution of RF current, voltage, and impedance along the inner con-
ductor of a 1/2-wavelength cavity.

© 1999 CRC Press LLC

conductor). The inner conductor is U-shaped to reduce the cavity height. With the mov-
able section (the plate tune control) fully extended, the inner conductor measures 38 in,
and the anode strap measures 7 in. The RF path from the anode strap to the inside of the
tube plate (along the surface because of the skin effect) is estimated to be about 8 in.
This makes the inner conductor maximum length about 53 in. This is too short to be a
1/2 wavelength at 88 MHz, the target low-end operating frequency in this example. The
full length of a 1/2-wave line is 67.1 in at 88 MHz.

The coarse-tuning and fine-tuning provisions of the cavity, coupled with the PA

tube's output capacity, resonate the plate line to the exact operating frequency. In effect,
they electrically lengthen the physically foreshortened line. This process, along with
proper loading, determines the plate impedance and, therefore, the efficiency.

Lengthening the Plate Line

The output capacity of the tube is the first element that electrically lengthens the plate
line. A 1/2-wave transmission line that is too short offers a high impedance that is
both resistive and inductive. The output capacity of the tube resonates this inductance.
The detrimental effects of the tube's output capacity, therefore, are eliminated. The
anode strap and the cavity inner conductor rotary section provide two methods of
coarse frequency adjustment for resonance.

Figure 4.25 The configuration of a practical 1/2-wavelength PA cavity.

© 1999 CRC Press LLC

The anode strap, shown in

Figure 4.26

, has a smaller cross-sectional area than the in-

ner conductor of the transmission line. Therefore, it has more inductance than an equal
length of the inner conductor. The anode-coupling strap acts as a series inductance and
electrically lengthens the plate circuit.

At low frequencies, one narrow anode strap is used. At midband frequencies, one

wide strap is used. The wide strap exhibits less inductance than the narrow strap and
does not electrically lengthen the plate circuit as much. At the upper end of the operat-
ing band, two anode straps are used. The parallel arrangement lowers the total induc-
tance of the strap connection and adds still less electrical length to the plate circuit.

The main section of the plate resonant line, together with the rotary section, func-

tions as a parallel inductance. RF current flows in the same direction through the trans-
mission line and the rotary section. Therefore, the magnetic fields of the two paths add.
When the rotary section is at maximum height, the magnetic coupling between the
main section of the transmission line and the rotary assembly is maximum. Because of
the relatively large mutual inductance provided by this close coupling, the total induc-
tance of these parallel inductors increases. This electrically lengthens the transmission
line and lowers the resonant frequency. The concept is illustrated in

Figure 4.27a

.

When the rotary section is at minimum height, the magnetic coupling between the

two parts of the inner conductor is minimum. This reduced coupling lowers the mutual
inductance, which lowers the total inductance of the parallel combination. The reduced

Figure 4.26 Coarse-tuning mechanisms for a 1/2-wave cavity.

© 1999 CRC Press LLC

inductance allows operation at a higher resonant frequency. This condition is illustrated
in

Figure 4.27b

. The rotary section provides an infinite number of coarse settings for

various operating frequencies.

The movable plate-tune assembly is located at the end of the inner-plate transmis-

sion line. It can be moved up and down to change the physical length of the inner con-
ductor by about 4-3/4 in. This assembly is linked to the front-panel plate-tuning knob,
providing a fine adjustment for cavity resonance.

Figure 4.27 Using the cavity's movable section to adjust for resonance: (

a) the rotary

section at maximum height, (

b) the rotary section at minimum height.

(

a)

(

b)

© 1999 CRC Press LLC

4.3.5 Folded 1/2-Wavelength Cavity

A special case of the 1/2-wavelength PA cavity is shown in

Figure 4.28

. The design

employs a folded 1/2-wave resonator constructed with coaxial aluminum and copper
tubing. This cavity arrangement eliminates the high-voltage blocking capacitor and
high-current shorting contacts of conventional designs by connecting the main trans-
mission-line resonant circuit conductor directly to the anode of the power tube. A
grounded, concentric transmission-line center conductor tunes the cavity with a vari-
able reentrant length inserted into the end of the main conductor opposite the tube.

The main conductor (the fixed portion of the plate line) is insulated from ground and

carries the anode dc potential. High-voltage power is fed at the fundamental-frequency
RF voltage null point, approximately 1/4 wavelength from the anode, for easy RF de-
coupling. A large surface area without sliding contacts results in minimal loss.

Incorporated into the tank design is a second harmonic suppressor. Rather than at-

tenuating the second harmonic after the signal has been generated and amplified, this
design essentially prevents formation of second-harmonic energy by series-LC trap-
ping the second-harmonic waveform at the point where the wave exhibits a high imped-

Figure 4.28 The basic design of a folded 1/2-wavelength cavity.

© 1999 CRC Press LLC

ance (approximately 1/4 wavelength from the anode). The second harmonic will peak
in voltage at the same point that the dc-plate potential is applied.

Plate tuning is accomplished by an adjustable bellows on the center portion of the

plate line, which is maintained at chassis ground potential. Output coupling is accom-
plished with an untuned loop intercepting the magnetic field concentration at the volt-
age null (maximum RF current) point of the main line. The PA loading control varies
the angular position of the plane of the loop with respect to the plate line, changing the
amount of magnetic field that it intercepts. Multiple phosphor-bronze leaves connect
one side of the output loop to ground and the other side to the center conductor of the
output transmission line. This allows for mechanical movement of the loop by the PA
loading control without using sliding contacts. The grounded loop improves immunity
to lightning and to static buildup on the antenna.

4.3.6 Wideband Cavity

The cavity systems discussed so far in this section provide adequate bandwidth for
many applications. Some uses, however, dictate an operating bandwidth beyond what
a conventional cavity amplifier can provide. One method of achieving wider operat-
ing bandwidth involves the use of a double-tuned overcoupled amplifier, shown in

Figure 4.29

. The system includes four controls to accomplish tuning:

·

Primary tune, which resonates the plate circuit and tends to tilt the response and
slide it up and down the bandpass.

·

Coupling, which sets the bandwidth of the PA. Increased coupling increases the
operating bandwidth and lowers the PA efficiency. When the coupling is ad-
justed, it can tilt the response and change the center of the bandpass, necessitating
readjustment of the plate-tune control.

·

Secondary tune, which resonates the secondary cavity and will tilt the response if
so adjusted. The secondary tune control typically can be used to slide the response
up and down the bandpass, similar to the primary tune control.

·

Loading, which determines the value of ripple in the bandpass response. Adjust-
ment of the loading control usually tilts the response and changes the bandwidth,
necessitating readjustment of the secondary tune and coupling controls.

4.3.7 Output Coupling

Coupling is the process by which RF energy is transferred from the amplifier cavity to
the output transmission line. Wideband cavity systems use coupling to transfer energy
from the primary cavity to the secondary cavity. Coupling in tube-type power amplifi-
ers usually transforms a high (plate or cavity) impedance to a lower output (transmis-
sion line) impedance. Both capacitive (electrostatic) and inductive (magnetic) cou-
pling methods are used in cavity RF amplifiers. In some designs, combinations of the
two methods are used.

© 1999 CRC Press LLC

Inductive Coupling

Inductive (magnetic) coupling employs the principles of transformer action. The effi-
ciency of the coupling depends upon three conditions:

·

The cross-sectional area under the coupling loop, compared to the cross-sectional
area of the cavity (see

Figure 4.30

). This effect can be compared to the turns ratio

principle of a transformer.

·

The orientation of the coupling loop to the axis of the magnetic field. Coupling
from the cavity is proportional to the cosine of the angle at which the coupling
loop is rotated away from the axis of the magnetic field, as illustrated in

Figure

4.31

.

·

The amount of magnetic field that the coupling loop intercepts (see

Figure 4.32

).

The strongest magnetic field will be found at the point of maximum RF current in
the cavity. This is the area where maximum inductive coupling is obtained.
Greater magnetic field strength also is found closer to the center conductor of the
cavity. Coupling, therefore, is inversely proportional to the distance of the cou-
pling loop from the center conductor.

Figure 4.29 Double-tuned overcoupled broadband cavity amplifier.

© 1999 CRC Press LLC

In both 1/4- and 1/2-wavelength cavities, the coupling loop generally feeds a 50

transmission line (the load). The loop is in series with the load and has considerable in-
ductance at very high frequencies. This inductance will reduce the RF current that
flows into the load, thus reducing power output. This effect can be overcome by placing
a variable capacitor in series with the output coupling loop. The load is connected to
one end of the coupling loop, and the variable capacitor ties the other end of the loop to

Figure 4.30 The use of inductive coupling in a 1/4-wavelength PA stage.

Figure 4.31 Top view of a cavity showing the inductive coupling loop rotated away from
the axis of the magnetic field of the cavity.

© 1999 CRC Press LLC

ground. The variable capacitor cancels some or all of the loop inductance. It functions
as the PA-stage loading control.

Maximum loop current and output power occurs when the loading capacitor cancels

all of the inductance of the loading loop. This lowers the plate impedance and results in
heavier loading.

Light loading results if the loading capacitance does not cancel all of the loop induc-

tance. The loop inductance that is not canceled causes a decrease in load current and
power output, and an increase in plate impedance.

Capacitive Coupling

Capacitive (electrostatic) coupling, which physically appears to be straightforward,
often baffles the applications engineer because of its unique characteristics.

Figure

4.33

shows a cavity amplifier with a capacitive coupling plate positioned near its cen-

ter conductor. This coupling plate is connected to the output load, which may be a
transmission line or a secondary cavity (for wideband operation). The parameters that
control the amount of capacitive coupling include:

·

The area of the coupling capacitor plate (the larger the area, the greater the cou-
pling)

·

The distance from the coupling plate to the center conductor (the greater the dis-
tance, the lighter the coupling)

Figure 4.32 Cutaway view of the cavity showing the coupling loop rotated away from
the axis of the magnetic field.

© 1999 CRC Press LLC

Maximum capacitive coupling occurs when the coupling plate is at the maximum
voltage point on the cavity center conductor.

To understand the effects of capacitive coupling, observe the equivalent circuit of

the cavity.

Figure 4.34

shows the PA tube, cavity (functioning as a parallel resonant cir-

cuit), and output section. The plate-blocking capacitor isolates the tube's dc voltage
from the cavity. The coupling capacitor and output load are physically in series, but
electrically they appear to be in parallel, as shown in

Figure 4.35

. The resistive compo-

nent of the equivalent parallel circuit is increased by the coupling reactance. The equiv-
alent parallel coupling reactance is absorbed into the parallel resonant circuit. This ex-
plains the need to retune the plate after changing the PA stage coupling (loading). The
coupling reactance may be a series capacitor or inductor.

The series-to-parallel transformations are explained by the following formula:

R

R

X

R

p

s

s

s

=

+

2

2

(4.12)

X

R

X

X

p

s

s

s

=

+

2

2

(4.13)

Where:
R

p

= effective parallel resistance

R

s

= actual series resistance

Figure 4.33 A 1/4-wavelength cavity with capacitive coupling to the output load.

© 1999 CRC Press LLC

X

s

= actual series reactance

X

p

= effective parallel reactance

Figure 4.34 The equivalent circuit of a 1/4-wavelength cavity amplifier with capacitive
coupling.

Figure 4.35 The equivalent circuit of a 1/4-wavelength cavity amplifier showing how se-
ries capacitive coupling appears electrically as a parallel circuit to the PA tube.

© 1999 CRC Press LLC

PA Loading

Proper loading of a cavity PA stage to the output transmission line is critical to de-
pendable operation and optimum efficiency of the overall system. Light coupling pro-
duces light loading and results in a high plate impedance; conversely, heavy coupling
results in heavier loading and a lower plate impedance. Maximum output power, coin-
ciding with maximum efficiency and acceptable dissipation, dictates a specific plate
impedance for a cavity of given design. This plate impedance is also dependent upon
the operating values of dc plate voltage (E

p

) and plate current (I

p

).

Plate impedance dictates the cavity parameters of loaded Q, RF circulating current,

and bandwidth. The relationships can be expressed as follows:

·

Loaded Q is directly proportional to the plate impedance and controls the other
two cavity parameters. Loaded Q = Z

p

/X

l

, where Z

p

= cavity plate impedance and

X

l

= cavity inductive reactance.

·

Circulating current in the cavity is much greater (by a factor of the loaded Q) than
the RF current supplied by the tube. Circulating current = Q

× I

p

, where I

p

= the RF

current supplied to the cavity by the tube.

·

The cavity bandwidth is dependent on the loaded Q and operating frequency.
Bandwidth = F

r

/Q, where F

r

= the cavity resonant frequency.

Heavy loading lowers the PA plate impedance and cavity Q. A lower Q reduces the

cavity RF circulating currents. In some cavities, high circulating currents can cause
cavity heating and premature failure of the plate or screen blocking capacitors. The ef-
fects of lower plate impedance--a by-product of heavy loading--are higher RF and dc
plate currents and reduced RF plate voltage. The instantaneous plate voltage is the re-
sult of the RF plate voltage added to the dc plate voltage. The reduced swing of plate
voltage causes less positive dc screen current to flow. Positive screen current flows only
when the plate voltage swings close to or below the value of the positive screen grid.

Light loading raises the plate impedance and cavity Q. A higher Q will increase the

cavity circulating currents, raising the possibility of component overheating and fail-
ure. The effects of higher plate impedance are reduced RF and dc plate current and in-
creased RF and dc plate voltage excursions. The higher cavity RF or peak dc voltages
may cause arcing in the cavity.

There is one value of plate impedance that will yield optimum output power, effi-

ciency, dissipation, and dependable operation. It is dictated by cavity design and the
values of the various dc and RF voltages and currents supplied to the stage.

Depending on the cavity design, light loading may seem deceptively attractive. The

dc plate voltage is constant (set by the power supply), and the lower dc plate current re-
sulting from light loading reduces the tube's overall dc input power. The RF output
power may change with light loading, depending on the plate impedance and cavity de-
sign, while efficiency will probably increase or, at worst, remain constant. Caution
must be exercised with light loading, however, because of the increased RF voltages
and circulating currents that such operation creates. Possible adverse effects include

© 1999 CRC Press LLC

cavity arcing and overheating of cavity components, such as capacitors. The manufac-
turer's recommendations on PA tube loading should, therefore, be carefully observed.

Despite the many similarities among various cavity designs, each imposes its own

set of operational requirements and limitations. No two cavity systems will tune up in
exactly the same fashion. Given proper maintenance, a cavity amplifier will provide
years of reliable service.

4.3.8 Mechanical Design

The operation of a cavity amplifier can be puzzling because of the nature of the major
component elements. It is often difficult to relate the electrical schematic diagram to
the mechanical assembly that exists within the transmitter. Consider the PA cavity
schematic diagram shown in

Figure 4.36

. The grounded screen stage is of conven-

tional design. Decoupling of the high-voltage power supply is accomplished by C1,
C2, C3, and L1. Capacitor C3 is located inside the PA chimney (cavity inner conduc-
tor). The RF sample lines provide two low-power RF outputs for a modulation moni-
tor or other test instrument. Neutralization inductors L3 and L4 consist of adjustable
grounding bars on the screen grid ring assembly.

Figure 4.37

shows the electrical equivalent of the PA cavity schematic diagram. The

1/4-wavelength cavity acts as the resonant tank for the PA. Coarse-tuning of the cavity

Figure 4.36 Typical VHF cavity amplifier.

© 1999 CRC Press LLC

is accomplished by adjustment of the shorting plane. Fine-tuning is performed by the
PA tuning control, which acts as a variable capacitor to bring the cavity into resonance.
The PA loading control consists of a variable capacitor that matches the cavity to the
load. The assembly made up of L2 and C6 prevents spurious oscillations within the cav-
ity.

Blocking capacitor C4 is constructed of a roll of Kapton insulating material

sandwiched between two circular sections of aluminum. (Kapton is a registered trade-
mark of Du Pont.) PA plate-tuning control C5 consists of an aluminum plate of large
surface area that can be moved in or out of the cavity to reach resonance. PA loading
control C7 is constructed much the same as the PA tuning assembly, with a large-area
paddle feeding the harmonic filter, located external to the cavity. The loading paddle
may be moved toward the PA tube or away from it to achieve the required loading. The
L2-C6 damper assembly actually consists of a 50

noninductive resistor mounted on

the side of the cavity wall. Component L2 is formed by the inductance of the connecting
strap between the plate-tuning paddle and the resistor. Component C6 is the equivalent
stray capacitance between the resistor and the surrounding cavity box.

It can be seen that cavity amplifiers involve as much mechanical engineering as they

do electrical engineering. The photographs of

Figure 4.38

show graphically the level of

complexity that a cavity amplifier may involve.

Figure 4.38a

depicts a VHF power am-

plifier (Philips) with broadband input circuitry.

Figure 4.38b

shows a wideband VHF

amplifier intended for television service (Varian).

Figure 4.38c

illustrates a VHF cavity

amplifier designed for FM broadcast service (Varian).

Figure 4.37 Electrical equivalent of the cavity amplifier shown in

Figure 4.36

.

© 1999 CRC Press LLC

Figure 4.38 VHF cavity amplifiers: (

a) cross-sectional view of a broadband design, (b)

cavity designed for television service, (c, next page) FM broadcast cavity amplifier.

(

b)

(

a)

© 1999 CRC Press LLC

4.4

High-Voltage Power Supplies

Virtually all power grid and microwave vacuum tubes require one or more sources of
high voltage to operate. The usual source of operating power is a single-phase or
multiphase supply operating from ac line current. Most supplies use silicon rectifiers
as the primary ac-to-dc converting device.

4.4.1 Silicon Rectifiers

Rectifier parameters generally are expressed in terms of reverse-voltage ratings and
mean forward-current ratings in a -wave rectifier circuit operating from a 60 Hz sup-
ply and feeding a purely resistive load. The three principle reverse-voltage ratings are:

·

Peak transient reverse voltage (V

rm

). The maximum value of any nonrecurrent

surge voltage. This value must never be exceeded.

·

Maximum repetitive reverse voltage (V

RM(rep)

). The maximum value of reverse

voltage that may be applied recurrently (in every cycle of 60 Hz power). This in-
cludes oscillatory voltages that may appear on the sinusoidal supply.

·

Working peak reverse voltage (V

RM(wkg)

). The crest value of the sinusoidal voltage

of the ac supply at its maximum limit. Rectifier manufacturers generally recom-
mend a value that has a significant safety margin, relative to the peak transient re-
verse voltage (V

rm

), to allow for transient overvoltages on the supply lines.

Figure 4.38

c

(

c)

© 1999 CRC Press LLC

There are three forward-current ratings of similar importance in the application of

silicon rectifiers:

·

Nonrecurrent surge current (I

FM,surge

). The maximum device transient current that

must not be exceeded at any time. I

FM,surge

is sometimes given as a single value, but

more often is presented in the form of a graph of permissible surge-current values
vs. time. Because silicon diodes have a relatively small thermal mass, the poten-
tial for short-term current overloads must be given careful consideration.

·

Repetitive peak forward current (I

FM,rep

). The maximum value of forward-current

reached in each cycle of the 60 Hz waveform. This value does not include random
peaks caused by transient disturbances.

·

Average forward current (I

FM,av

). The upper limit for average load current through

the device. This limit is always well below the repetitive peak forward-current
rating to ensure an adequate margin of safety.

Rectifier manufacturers generally supply curves of the instantaneous forward volt-

age vs. instantaneous forward current at one or more specific operating temperatures.
These curves establish the forward-mode upper operating parameters of the device.

Figure 4.39

shows a typical rectifier application in a bridge circuit.

4.4.2 Operating Rectifiers in Series

High-voltage power supplies (5 kV and greater) often require rectifier voltage ratings
well beyond those typically available from the semiconductor industry. To meet the
requirements of the application, manufacturers commonly use silicon diodes in a se-
ries configuration to give the required working peak reverse voltage. For such a con-
figuration to work properly, the voltage across any one diode must not exceed the
rated peak transient reverse voltage (V

rm

) at any time. The dissimilarity commonly

found between the reverse leakage current characteristics of different diodes of the
same type number makes this objective difficult to achieve. The problem normally is

Figure 4.39 Conventional capacitor input filter full-wave bridge.

© 1999 CRC Press LLC

overcome by connecting shunt resistors across each rectifier in the chain, as shown in

Figure 4.40

. The resistors are chosen so that the current through the shunt elements

(when the diodes are reverse-biased) will be several times greater than the leakage
current of the diodes themselves.

The carrier storage effect also must be considered in the use of a series-connected

rectifier stack. If precautions are not taken, different diode recovery times (caused by
the carrier storage phenomenon) will effectively force the full applied reverse voltage
across a small number of diodes, or even a single diode. This problem can be prevented
by connecting small-value capacitors across each diode in the rectifier stack. The ca-
pacitors equalize the transient reverse voltages during the carrier storage recovery peri-
ods of the individual diodes.

Figure 4.41

illustrates a common circuit configuration for a high-voltage 3-phase

rectifier bank. A photograph of a high-voltage series-connected 3-phase rectifier as-
sembly is shown in

Figure 4.42

.

Figure 4.40 A portion of a high-voltage series-connected rectifier stack.

Figure 4.41 A 3-phase delta-connected high-voltage rectifier.

© 1999 CRC Press LLC

4.4.3 Operating Rectifiers in Parallel

Silicon rectifiers are used in a parallel configuration when a large amount of current
is required from the power supply. Current sharing is the major design problem with a
parallel rectifier assembly because diodes of the same type number do not necessarily
exhibit the same forward characteristics.

Semiconductor manufacturers often divide production runs of rectifiers into toler-

ance groups, matching forward characteristics of the various devices. When parallel di-
odes are used, devices from the same tolerance group must be used to avoid unequal
sharing of the load current. As a margin of safety, designers allow a substantial derating
factor for devices in a parallel assembly to ensure that the maximum operating limits of
any one component are not exceeded.

The problems inherent in a parallel rectifier assembly can be reduced through the

use of a resistance or reactance in series with each component, as shown in

Figure 4.43

.

The buildout resistances (R1 through R4) force the diodes to share the load current
equally. Such assemblies can, however, be difficult to construct and may be more ex-
pensive than simply adding diodes or going to higher-rated components.

Figure 4.42 High-voltage rectifier
a s s e m b l y

f o r

a

3 - p h a s e

delta-connected circuit.

© 1999 CRC Press LLC

4.4.4 Silicon Avalanche Rectifiers

The silicon avalanche diode is a special type of rectifier that can withstand high re-
verse power dissipation. For example, an avalanche diode with a normal forward rat-
ing of 10 A can dissipate a reverse transient of 8 kW for 10 ms without damage. This
characteristic of the device allows elimination of the surge-absorption capacitor and
voltage-dividing resistor networks needed when conventional silicon diodes are used
in a series rectifier assembly. Because fewer diodes are needed for a given applied re-
verse voltage, significant underrating of the device (to allow for reverse-voltage tran-
sient peaks) is not required.

When an extra-high-voltage rectifier stack is used, it is still advisable to install shunt

capacitors--but not resistors--in an avalanche diode assembly. The capacitors are de-
signed to compensate for the effects of carrier storage and stray capacitance in a long
series assembly.

4.4.5 Thyristor Servo Systems

Thyristor control of ac power has become a common method of regulating high-volt-
age power supplies. The type of servo system employed depends on the application.

Figure 4.44

shows a basic single-phase ac control circuit using discrete thyristors. The

rms load current (I

rms

) at any specific phase delay angle (

) is given in terms of the

normal full-load rms current at a phase delay of zero (I

rms-0

):

( )

I

I

rms

rms

=

- +

-

-

1

2

1

2

1 2

sin

(4.14)

The load rms voltage at any particular phase-delay angle bears the same relationship
to the full-load rms voltage at zero phase delay as the previous equation illustrates for
load current. An analysis of the mathematics shows that although the theoretical delay
range for complete control of a resistive load is 0 to 180°, a practical span of 20 to
160° gives a power-control range of approximately 99 to 1 percent of maximum out-
put to the load.

Figure

4.45

illustrates typical phase-control waveforms.

Figure 4.43 Using buildout resistances
to force current sharing in a parallel recti-
fier assembly.

© 1999 CRC Press LLC

The circuit shown in

Figure 4.44

requires a source of gate trigger pulses that must be

isolated from each other by at least the peak value of the applied ac voltage. The two
gate pulse trains must also be phased 180° with respect to each other. Also, the gate
pulse trains must shift together with respect to the ac supply voltage phase when power
throughput is adjusted.

Some power-control systems use two identical, but isolated, gate pulse trains operat-

ing at a frequency of twice the applied supply voltage (120 Hz for a 60 Hz system). Un-
der such an arrangement, the forward-biased thyristor will fire when the gate pulses are
applied to the silicon controlled rectifier (SCR) pair. The reverse-biased thyristor will

Figure 4.44 Inverse-parallel thyristor ac power control: (

a) circuit diagram, (b) voltage

and current waveforms for full and reduced thyristor control angles. The waveforms ap-
ply for a purely resistive load.

(

a)

(

b)

© 1999 CRC Press LLC

not fire. Normally, it is considered unsafe to drive a thyristor gate positive while its an-
ode-cathode is reverse-biased. In this case, however, it may be permissible because the
thyristor that is fired immediately conducts and removes the reverse voltage from the
other device. The gate of the reverse-biased device is then being triggered on a thyristor
that essentially has no applied voltage.

Figure 4.45 Waveforms in an ac circuit using thyristor power control.

© 1999 CRC Press LLC

Inductive Loads

The waveforms shown in

Figure 4.46

illustrate effects of phase control on an induc-

tive load. When inductive loads are driven at a reduced conduction angle, a sharp
transient change of load voltage occurs at the end of each current pulse (or loop). The

Figure 4.46 Voltage and current waveforms for inverse-parallel thyristor power control
with an inductive load: (

a) full conduction, (b) small-angle phase reduction, (c)

large-angle phase reduction.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

transients generally have no effect on the load, but they can be dangerous to proper
operation of the thyristors. When the conducting thyristor turns off, thereby discon-
necting the load from the ac line supply, the voltage at the load rapidly drops to zero.
This rapid voltage change, in effect, applies a sharply rising positive anode voltage to
the thyristor opposing the device that has been conducting. If the thyristor dv/dt
(change in voltage as a function of a change in time) rating is exceeded, the opposing
device will turn on and conduction will take place, independent of any gate drive
pulse.

A common protective approach involves the addition of a resistor-capacitor (RC)

snubber circuit to control the rate of voltage change seen across the terminals of the thy-
ristor pair. Whenever a thyristor pair is used to drive an inductive load, such as a power
transformer, it is critically important that each device fires at a point in the applied
waveform exactly 180° relative to the other. If proper timing is not achieved, the posi-
tive and negative current loops will differ in magnitude, causing a direct current to flow
through the primary side of the transformer. A common trigger control circuit, there-
fore, should be used to determine gate timing for thyristor pairs.

Applications

Several approaches are possible for thyristor power control in a 3-phase ac system.
The circuit shown in

Figure 4.47

consists of essentially three independent, but inter-

locked, single-phase thyristor controllers. This circuit is probably the most common
configuration found in commercial and industrial equipment.

In a typical application, the thyristor pairs feed a power transformer with multitap

primary windings, thereby giving the user an adjustment range to compensate for varia-

Figure 4.47 Modified full-thyristor 3-phase ac control of an inductive delta load.

© 1999 CRC Press LLC

tions in utility company line voltages from one location to another. A common proce-
dure specifies selection of transformer tap positions that yield a power output of 105
percent when nominal utility company line voltages are present. The thyristor
power-control system then is used to reduce the angle of conduction of the SCR pairs as
necessary to cause a reduction in line voltage to the power transformer to yield 100 per-
cent rated power output from the power supply. A servo loop from a sample point at the
load may be used to compensate automatically for line-voltage variations. With such an
arrangement, the thyristors are kept within a reasonable degree of retarded phase opera-
tion. Line voltages will be allowed to sag 5 percent or so without affecting the dc supply
output. Utility supply voltage excursions above the nominal value simply will result in
further delayed triggering of the SCR pairs.

Thyristor control of high-power loads (200 kW and above) typically uses transform-

ers that provide 6- or 12-phase outputs. Although they are more complicated and ex-
pensive, such designs allow additional operational control, and filtering requirements
are reduced significantly.

Figure 4.48

shows a 6-phase boost rectifier circuit. The con-

figuration consists basically of a full-wave 3-phase SCR bridge connected to a
wye-configured transformer secondary. A second bridge, consisting of six diodes, is
connected to low-voltage taps on the same transformer. When the SCRs are fully on,
their output is at a higher voltage than the diode bridge. As a result, the diodes are re-
verse-biased and turned off. When the SCRs are partially on, the diodes are free to con-

Figure 4.48 A 6-phase boost rectifier circuit.

© 1999 CRC Press LLC

duct. The diodes improve the quality of the output waveform during low-voltage (re-
duced conduction angle) conditions. The minimum output level of the supply is deter-
mined by the transformer taps to which the diodes are connected.

A thyristor-driven 3-phase power-control circuit is shown in

Figure 4.49

. A sin-

gle-phase power-control circuit is shown in

Figure 4.50

.

Triggering Circuits

Accurate, synchronized triggering of the gate pulses is a critical element in thyristor
control of a 3-phase power supply. The gate signal must be synchronized properly
with the phase of the ac line that it is controlling. The pulse also must properly match
the phase-delay angle of the gates of other thyristors in the power-control system.
Lack of proper synchronization of gate pulse signals between thyristor pairs can result
in improper current sharing (current hogging) among individual legs of the 3-phase
supply.

The gate circuit must be protected against electrical disturbances that could make

proper operation of the power-control system difficult or unreliable. Electrical isola-
tion of the gate is a common approach. Standard practice calls for the use of gate pulse
transformers in thyristor servo system gating cards. Pulse transformers are fer-
rite-cored devices with a single primary winding and (usually) multiple secondary

Figure 4.49 Thyristor-controlled high-voltage servo power supply.

© 1999 CRC Press LLC

windings that feed, or at least control, the individual gates of a back-to-back thyristor
pair. This concept is illustrated in

Figure 4.51

. Newer thyristor designs may use

optocouplers to achieve the necessary electrical isolation between the trigger circuit
and the gate.

It is common practice to tightly twist together the control leads from the gate and

cathode of a thyristor to the gating card assembly. This practice provides a degree of im-
munity to high-energy pulses that might inadvertently trigger the thyristor gate. The
gate circuit must be designed and configured carefully to reduce inductive and capaci-
tive coupling that might occur between power and control circuits. Because of the high
di/dt (change in current as a function of a change in time) conditions commonly found
in thyristor-controlled power circuits, power wiring and control (gate) wiring must be
separated physically as much as possible. Shielding of gating cards in a metal card cage
is advisable.

Equipment manufacturers use various means to decrease gate sensitivity to transient

sources, including placement of a series resistor in the gate circuit and/or a shunting ca-
pacitor between the gate and cathode. A series resistor has the effect of decreasing gate
sensitivity, increasing the allowable dv/dt of the thyristor and reducing the turn-off
time, which simultaneously increases the required holding and latching currents. The
use of a shunt capacitor between the gate and cathode leads reduces high-frequency
noise components that might be present on the gate lead and increases the dv/dt with-
stand capability of the thyristor.

Figure 4.50 Phase-controlled power supply with primary regulation.

© 1999 CRC Press LLC

Fusing

Current limiting is a basic method of protection for any piece of equipment operated
from the utility ac line. The device typically used for breaking fault currents is either a
fuse or a circuit breaker. Some designs incorporate both components. Semiconductor
fuses often are used in conjunction with a circuit breaker to provide added protection.
Semiconductor fuses operate more rapidly (typically within 8.3 ms) and more predict-
ably than common fuses or circuit breakers. Surge currents caused by a fault can de-
stroy a semiconductor device, such as a power thyristor, before the ac line circuit
breaker has time to act. Manufacturers of semiconductor fuses and thyristors usually
specify in their data sheets the I

2

t ratings of each device. Because the thyristor rating

normally assumes that the device is operating at maximum rated current and maxi-
mum junction temperature (conditions that do not represent normal operation), a
safety factor is ensured.

Control Flexibility

Thyristor servo control of a high-voltage power supply is beneficial to the user for a
number of reasons, including:

·

Wide control over ac input voltages

Figure 4.51 Simplified block diagram of the gating circuit for a phase-control system us-
ing back-to-back SCRs.

© 1999 CRC Press LLC

·

Capability to compensate automatically for line-voltage variations

·

Capability to soft-start the dc supply

Thyristor control circuits typically include a ramp generator that increases the ac line
voltage to the power transformer from zero to full value within 2 to 5 s. This prevents
high surge currents through rectifier stacks and filter capacitors during system
startup.

Although thyristor servo systems are preferred over other power-control approaches

from an operational standpoint, they are not without their drawbacks. The control sys-
tem is complex and can be damaged by transient activity on the ac power line. Conven-
tional power contactors are simple and straightforward. They either make contact or
they do not. For reliable operation of the thyristor servo system, attention must be given
to transient suppression at the incoming power lines.

4.4.6 Polyphase Rectifier Circuits

High-voltage power supplies typically used in vacuum tube circuits incorporate
multiphase rectification of the ac line voltage. Common configurations include 3-, 6-,
and 12-phase, with 3-phase rectification being the most common.

Figure 4.52

illus-

trates four approaches to 3-phase rectification:

·

Three-phase half-wave wye,

Figure 4.52a

. Three half-wave rectifiers are used in

each leg of the secondary wye forming one phase. In such an arrangement, each
diode carries current one-third of each cycle, and the output wave pulses at three
times the frequency of the ac supply. In order to avoid direct-current saturation in
the transformer, it is necessary to employ a 3-phase transformer rather than three
single-phase transformers.

·

Three-phase full-wave bridge,

Figure 4.52b

. Six diodes are used in this circuit to

produce a low ripple output with a frequency of six times the input ac waveform.
It is permissible in this configuration to use three single-phase transformers, if de-
sired.

·

Six-phase star,

Figure 4.52c

. This circuit, also known as a 3-phase diametric con-

figuration, uses six diodes with a transformer secondary configured as a star, as il-
lustrated in the figure. The output ripple frequency is six times the input ac wave-
form.

·

Three-phase double-wye,

Figure 4.52d

. This circuit uses six diodes and a compli-

cated configuration of transformer windings. Note the balance coil (interphase
transformer) in the circuit.

The relative merits of these rectifier configurations are listed in

Table 4.3

.

Polyphase rectifiers are used when the dc power required is on the order of 2 kW or

more. The main advantages of a polyphase power supply over a single-phase supply in-
clude the following:

·

Division of the load current between three or more lines to reduce line losses.

© 1999 CRC Press LLC

Figure 4.52 Basic 3-phase rectifier circuits: (

a) half-wave wye, (b) full-wave bridge, (c)

6-phase star, (

d) 3-phase double-wye.

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

Table 4.3 Operating Parameters of 3-Phase Rectifier Configurations

Parameter

3-Phase

Star

3-Phase

Bridge

6-Phase

Star

3-Phase

Double-Y

Multiplier

1

Rectifier elements

3

6

6

6

Rms dc voltage out-
put

1.02

1.00

1.00

1.00

Average dc voltage
output

Peak dc voltage
output

1.21

1.05

1.05

1.05

Average dc voltage
output

Peak reverse volts
per rectifier

2.09

1.05

2.09

2.42

Average dc voltage
output

2.45

2.45

2.83

2.83

Rms secondary
volts per trans-
former leg

1.41

1.41

1.41

1.41

Rms secondary
volts line-to-line

Average dc output
current per rectifier

0.333

0.333

0.167

0.167

Average dc output
current

Rms current per
rectifier, resistive
load

0.587

0.579

0.409

0.293

Average dc output
current

Rms current per
rectifier, inductive
load

0.578

0.578

0.408

0.289

Average dc output
current

Percent ripple

18.3

4.2

4.2

4.2

Ripple frequency

3

6

6

6

Line frequency

AC line power fac-
tor

0.826

0.955

0.955

0.955

Transformer sec-
ondary rms volts
per leg

2

0.855

0.428

0.740

0.855

Average dc voltage
output

Transformer sec-
ondary rms volts
line-to-line

1.48

0.740

1.48
(max)

1.71
(max, no
load)

Average dc voltage
output

Secondary line cur-
rent

0.578

0.816

0.408

0.289

Average dc output
current

Transformer sec-
ondary
volt-amperes

1.48

1.05

1.81

1.48

DC watts output

Primary line current 0.817

1.41

0.817

0.707

(Avg. load I x sec-
ondary leg V) / pri-
mary line V

1

To determine the value of a parameter in any column, multiply the factor shown by the value

given in this column.

2

For inductive load or large choke input filter

.

© 1999 CRC Press LLC

·

Significantly reduced filtering requirements after rectification because of the low
ripple output of a polyphase rectifier.

·

Improved voltage regulation when using an inductive-input filter. Output voltage
soaring is typically 6 percent or less from full-load to no-load conditions.

·

Greater choice of output voltages from a given transformer by selection of either
a delta or wye configuration.

The main disadvantage of a polyphase system is its susceptibility to phase imbalance.
Resulting operational problems include increased ripple at the output of the supply
and uneven sharing of the load current by the transformer windings.

4.4.7 Power Supply Filter Circuits

A filter network for a high-voltage power supply typically consists of a series induc-
tance and one or more shunt capacitances. Bleeder resistors and various circuit pro-
tection devices are usually incorporated as well. Filter systems can be divided into
two basic types:

·

Inductive input: filter circuits that present a series inductance to the rectifier out-
put

·

Capacitive input: filter circuits that present a shunt capacitance to the rectifier
output

Inductive Input Filter

The inductive input filter is the most common configuration found in high-power RF
equipment. A common circuit is shown in

Figure 4.53

, along with typical current

waveforms. When the input inductance is infinite, current through the inductance is
constant and is carried at any moment by the rectifier anode that has the most positive
voltage applied to it at that instant. As the alternating voltage being rectified passes
through zero, the current suddenly transfers from one anode to another, producing
square current waves through the individual rectifier devices.

When the input inductance is finite (but not too small), the situation changes to that

shown by the solid lines of

Figure 4.53

. The current through the input inductance tends

to increase when the output voltage of the rectifier exceeds the average or dc value, and
to decrease when the rectifier output voltage is less than the dc value. This causes the
current through the individual anodes to be modified as shown. If the input inductance
is too small, the current decreases to zero during a portion of the time between the peaks
of the rectifier output voltage. The conditions then correspond to a capacitor input filter
system.

© 1999 CRC Press LLC

Figure 4.53 Voltage and current waveshapes for an inductive-input filter driven by a
3-phase supply.

© 1999 CRC Press LLC

The output wave of the rectifier can be considered as consisting of a dc component

upon which are superimposed ac voltages (ripple voltages). To a first approximation,
the fluctuation in output current resulting from a finite input inductance can be consid-
ered as the current resulting from the lowest-frequency component of the ripple voltage
acting against the impedance of the input inductance. This assumption is permissible
because the higher-frequency components in the ripple voltage are smaller, and at the
same time encounter higher impedance. Furthermore, in practical filters the shunting
capacitor following the input inductance has a small impedance at the ripple frequency
compared with the reactance of the input inductance. The peak current resulting from a
finite input inductance, therefore, is given approximately by the relation [1]:

I

I

E

E

E

L

f

i

eff

= +

1

1

0

1

(4.15)

Where:
I

f

= peak current with finite input inductance

I

i

= peak current with infinite input inductance

E

1

/E

0

= ratio of lowest-frequency ripple component to the dc voltage in the rectifier

output

R

eff

= effective load resistance

L

1

= reactance of the incremental value of the input inductance at the lowest rip-

ple frequency

This equation is derived as follows:

·

The peak alternating current through the input inductance is approximately

E

L

1

1

·

The average or dc current is E

R

eff

0

·

The peak current with finite inductance is, therefore:

(

)

(

)

E

L

E

R

eff

1

1

0

+

·

The current with infinite inductance is E

R

eff

0

The effective load resistance value consists of the actual load resistance plus filter re-
sistance plus equivalent diode and transformer resistances.

The normal operation of an inductive input filter requires that there be a continuous

flow of current through the input inductance. The peak alternating current flowing
through the input inductance, therefore, must be less than the dc output current of the
rectifier. This condition is realized by satisfying the approximate relation:

L

R

E

E

eff

1

1

0

=

(4.16)

In the practical case of a 60 Hz single-phase full-wave rectifier circuit, the foregoing
equation becomes:

© 1999 CRC Press LLC

L

R

eff

1

1130

=

(4.17)

In a polyphase system, the required value of L

1

is significantly less. The higher the

load resistance (the lower the dc load current), the more difficult it is to maintain a
continuous flow of current and, with a given L

1

, the previous equation will not be sat-

isfied when the load resistance exceeds a critical value.

The minimum allowable input inductance (

L

1

) is termed the critical inductance.

When the inductance is less than the critical value, the filter acts as a capacitor input cir-
cuit. When the dc drawn from the rectifiers varies, it is still necessary to satisfy the

L

1

equation at all times, particularly if good voltage regulation is to be maintained. To ful-
fill this requirement at small load currents without excessive inductance, it is necessary
to place a bleeder resistance across the output of the filter system in order to limit R

eff

to

a value corresponding to a reasonable value of L

1

.

Capacitive Input Filter

When a shunt capacitance rather than a series inductance is presented to the output of
a rectifier, the behavior of the circuit is greatly modified. Each time the positive crest
alternating voltage of the transformer is applied to one of the rectifier anodes, the in-
put capacitor charges up to just slightly less than this peak voltage. The rectifier then
ceases to deliver current to the filter until another anode approaches its peak positive
potential, when the capacitor is charged again. During the interval when the voltage
across the input capacitor is greater than the potential of any of the anodes, the voltage
across the input capacitor drops off nearly linearly with time, because the first filter
inductance draws a substantially constant current from the input capacitor. A typical
set of voltage and current waves is illustrated in

Figure 4.54

.

The addition of a shunt capacitor to the input of a filter produces fundamental

changes in behavior, including the following:

·

The output voltage is appreciably higher than with an inductance input.

·

The ripple voltage is less with a capacitance input filter than with an inductance
input filter.

·

The dc voltage across the filter input drops as the load current increases for the ca-
pacitive input case, instead of being substantially constant as for the inductive in-
put case.

·

The ratio of peak-to-average anode current at the rectifiers is higher in the capaci-
tive case.

·

The utilization factor of the transformer is less with a capacitive input configura-
tion.

Filters incorporating shunt capacitor inputs generally are employed when the

amount of dc power required is small. Inductance input filters are used when the
amount of power involved is large; the higher utilization factor and lower peak current

© 1999 CRC Press LLC

result in important savings in rectifier and transformer costs under these conditions. In-
ductance input systems are almost universally employed in polyphase rectifiers.

4.5

Parameter Sampling Circuits

Most RF system parameters can be reduced to a sample voltage by using voltage di-
viders or current-sense resistors in the circuit to be measured [2]. Remote metering is
accomplished by sending a signal representing the sample voltage to the remote me-
tering point, and then displaying a value representing the original parameter. In prac-
tice, the actual sampling ratio (sample voltage/parameter value) is not particularly im-
portant as long as the sampling ratio is stable and the sample voltage is reasonable

Figure 4.54 Characteristics of a capacitive input filter circuit: (

a) schematic diagram, (b)

voltage waveshape across the input capacitor, (

c) waveshape of current flowing

through the diodes.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

(i.e., low enough to be easily handled, yet high enough to be immune to noise).

Figure

4.55

shows some typical voltage sampling circuits for high-power RF systems.

Figure 4.55a

shows how a voltage divider can be used to sample plate voltage. The

circuit is independent of the transmitter front panel plate voltage meter. A zener clamp-
ing diode is placed across the sample line for safety; if the shunt resistor opened, the full
plate voltage would be present at the sampling output terminals (assuming minimal
loading by the remote indicator). The zener will typically fail in a short-circuit mode,
protecting the external sampling equipment.

The sampling circuit of

Figure 4.55a

is adequate for systems that measure plate volt-

age from a ground reference. In the event that the cathode operates at a potential other
than ground, the sampling circuit of

Figure 4.55b

may be used. The sampling line is bal-

anced, with both terminals floating above ground by 1 k

The differential output volt-

age is independent of the common mode input voltage if all the resistors are precisely
matched.

Several techniques are available for dealing with differential sampling voltages

(where neither side of the sample is grounded). Typical approaches include the follow-
ing:

·

Isolation amplifier. Isolation amplifiers provide isolation between the input and
output by using magnetic, capacitive, optical, or thermal coupling between the in-

Figure 4.55 Plate voltage sampling circuits: (

a) single-ended, (b) differential.

(

a)

(

b)

© 1999 CRC Press LLC

put and output. Some units include feedback techniques to reduce nonlinearities
induced by the coupling method.

·

Differential amplifier. In most cases, electrical isolation between the sampling
circuit and the remote metering equipment is not required. Instead, a differential
voltage must be changed to a single ended voltage (one side grounded). This can
be accomplished using the differential amplifier shown in

Figure 4.56a

or the in-

strumentation amplifier shown in

Figure 4.56b

.

Measurement of plate and/or screen current is typically done by inserting a sam-

pling resistor into the ground return lead of the respective supplies. This approach is
shown in

Figure 4.57

. Notice that the remote plate current metering is taken across R1,

with balancing resistors of 1.2 k

used to lift the sample above the ground reference. A

bidirectional zener diode is placed across the output of the sample terminals for protec-
tion. The plate current meter on the transmitter is sampled across R2, and the over-cur-
rent relay is sampled across R3. Device X1 is a crowbar that will shunt the plate current
to ground in the event of a failure in the R1R3 chain.

The initial tolerance of resistors used as current sense elements or in a voltage di-

vider circuit is usually not particularly critical. An inexact resistance will cause the
sample voltage to be something other than that expected. However, whatever sample
voltage appears, the remote meter can be calibrated to agree with the local meter. The
more important attribute is resistance changes with time and temperature.

Figure 4.56 Differential amplifier circuits: (

a) basic circuit, (b) instrumentation amplifier.

(

a)

(

b)

© 1999 CRC Press LLC

Figure 4.57 Transmitter high-voltage power supply showing remote plate current and voltage sensing circuits.

©

1
999
CRC Press LLC

A carbon composition resistor can commonly have a temperature coefficient of

about

±600 ppm/C° for resistors up to 1 K. This increases to ±1875 ppm/C° for resis-

tors up to 1 M

. If these resistors were used in a common plate voltage sampling circuit,

such as than shown in

Figure 4.57

, the sample voltage could change as much as 20 per-

cent over a 0 C

° to 50 C° temperature range. Metal film resistors are available with

temperature coefficients as low as 15 ppm/C

° . Bulk metal resistors are available with

temperature coefficients of 5 ppm/C

° . If the resistors in a voltage divider are at the

same temperature, the matching between the temperature coefficient of resistance
(TCR) is more important than the actual TCR. Because the TCR matching is generally
better than the TCR of an individual resistor, sampling systems should be designed to
keep the resistors in the network at the same operating temperature.

In differential amplifiers, the matching of resistor values is critical. To insure the re-

sistance values track with temperature, it is common practice to use a resistor network
made using hybrid film techniques or integrated circuit techniques. Because the resis-
tors are made at the same time and of the same materials, the TCRs track closely.

4.6

References

1. Terman, F. E., Radio Engineering, 3rd. Ed., McGraw-Hill, New York, pg. 560,

1947.

2. Hallikainen, Harold, "Transmitter Control Systems," in NAB Engineering Hand-

book, 9th Ed., Jerry C. Whitaker (ed.), National Association of Broadcasters, Wash-
ington, D.C., 1998.

4.7

Bibliography

Crutchfield, E. B. (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, 1992.

Gray, T. S., Applied Electronics, Massachusetts Institute of Technology, 1954.
Heising, R. A., "Modulation in Radio Telephony," Proceedings of the IRE, Vol. 9, no.

3, pg. 305, June 1921.

Heising, R. A., "Transmission System," U.S. Patent no. 1,655,543, January 1928.
Jordan, Edward C. (ed.), Reference Data for Engineers: Radio, Electronics, Com-

puters, and Communications, 7th Ed., Howard W. Sams, Indianapolis, IN, 1985.

Laboratory Staff, The Care and Feeding of Power Grid Tubes, Varian Eimac, San

Carlos, CA, 1984.

Martin, T. L., Jr., Electronic Circuits, Prentice Hall, Englewood Cliffs, NJ, 1955.
Mendenhall, G. N., M. B. Shrestha, and E. J. Anthony, "FM Broadcast Transmitters,"

in NAB Engineering Handbook, 8th Ed., E. B. Crutchfield (ed.), National Associa-
tion of Broadcasters, Washington, DC, pp. 429474, 1992.

Shrestha, Mukunda B., "Design of Tube Power Amplifiers for Optimum FM Transmit-

ter Performance," IEEE Transactions on Broadcasting, Vol. 36, no. 1, IEEE, New
York, pp. 4664, March 1990.

Terman, F. E., and J. R. Woodyard, "A High Efficiency Grid-Modulated Amplifier,"

Proceedings of the IRE, Vol. 26, no. 8, pg. 929, August 1938.

Terman, F. E., Radio Engineering, 3rd. Ed., McGraw-Hill, New York, 1947.

© 1999 CRC Press LLC

Weirather, R. R., G. L. Best, and R. Plonka, "TV Transmitters," in NAB Engineering

Handbook, 8th Ed., E. B. Crutchfield (ed.), National Association of Broadcasters,
Washington, DC, pp. 499551, 1992.

Whitaker, Jerry C., AC Power Systems, 2nd Ed., CRC Press, Boca Raton, FL, 1998.
Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1991.

Woodard, George W., "AM Transmitters," in NAB Engineering Handbook, 8th Ed.,

E. B. Crutchfield (ed.), National Association of Broadcasters, Washington, DC, pp.
353381, 1992.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

5 Applying Vacuum Tube Devices"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

5

Applying Vacuum Tube Devices

5.1

Introduction

Numerous circuits have been developed to meet the varied needs of high-power RF
amplification. Designers have adapted standard vacuum tube devices to specific ap-
plications that require unique performance criteria. Most of these circuits are built
upon a few basic concepts.

5.2

AM Power Amplification Systems

Class C plate modulation is the classic method of producing an AM waveform [1].

Figure 5.1

shows the basic circuit. Triodes and tetrodes may be used as the modulator

or carrier tube. Triodes offer the simplest and most common configuration.

Numerous variations on the basic design exist, including a combination of plate and

screen grid modulation. The carrier signal is applied to the control grid, and the modu-
lating signal is applied to the screen and plate. The plate is fully modulated, and the
screen is modulated 70 to 100 percent to achieve 100 percent carrier modulation. Mod-
ulation of the screen can be accomplished using one of the following methods:

·

Screen voltage supplied through a dropping resistor connected to the
unmodulated dc plate supply, shown in

Figure 5.2a

·

An additional (third) winding on the modulation transformer, illustrated in

Figure

5.2b

·

A modulation choke placed in series with a low-voltage fixed screen supply,
shown in

Figure 5.2c

Depending on the design, screen/plate modulation also may require partial modula-
tion of the control grid to achieve the desired performance characteristics.

During the portion of the modulation cycle when the plate voltage is increased, the

screen current decreases. If the screen is supplied through an impedance, such as the
screen dropping resistance of a modulation choke, the voltage drop in this series imped-
ance becomes less and the screen voltage rises in the desired manner. During the part of
the modulation cycle when the plate voltage is decreased, the screen current increases,

© 1999 CRC Press LLC

causing a greater voltage drop in the screen series impedance, and lowering the voltage
on the screen of the tube. The value of the screen bypass capacitor in the class C stage is
a compromise between good RF bypassing and the shunting effect of this capacitance
on the screen modulation circuit.

5.2.1 Control Grid Modulation

Although not as common as class C plate modulation, a class C RF amplifier may be
modulated by varying the voltage on the control grid of a triode [1]. This approach,
shown in

Figure 5.3

, produces a change in the magnitude of the plate current pulses

and, therefore, variations in the output waveform taken from the plate tank. Both the
carrier and modulation signals are applied to the grid.

The primary benefit of grid modulation is that a modulator is not required. Grid

modulation, however, requires a fixed plate supply voltage that is twice the peak RF
voltage without modulation. The result is higher plate dissipation at lower modulation
levels, including carrier level. The typical plate efficiency of a grid-modulated stage
may range from only 35 to 45 percent at carrier. Grid modulation typically is used only
in systems where plate modulation transformers cannot provide adequate bandwidth
for the intended application.

When grid modulation is used, the screen voltage and grid bias must be taken from

sources with good regulation. This usually means separate low-voltage power supplies.

Figure 5.1 The classic plate-modulated PA circuit for AM applications.

© 1999 CRC Press LLC

5.2.2 Suppressor Grid Modulation

The output of a class C pentode amplifier may be controlled by applying a modulating
voltage, superimposed on a suitable bias, to the suppressor grid to produce an AM
waveform [1]. As the suppressor grid becomes more negative, the minimum instanta-

Figure 5.2 Basic methods of screen and plate modulation: (

a) plate modulation, (b)

plate and screen modulation, (

c) plate modulation using separate plate/screen supplies.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

neous potential at which current can be drawn to the plate is increased. Thus, as mod-
ulation varies the suppressor grid potential, the output voltage changes.

This method of modulating an RF amplifier provides about the same plate effi-

ciency as a grid-modulated stage. The overall operating efficiency, however, is slightly
lower because of increased screen grid losses associated with the design. Linearity of
the circuit is not usually high.

5.2.3 Cathode Modulation

Cathode modulation incorporates the principles of both control grid and plate modu-
lation [1]. As shown in

Figure 5.4

, a modulation transformer in the cathode circuit

varies the grid-cathode potential, as well as the plate-cathode potential. The ratio of
grid modulation to plate modulation is set by adjustment of the tap shown in the fig-
ure. Grid leak bias typically is employed to improve linearity of the stage.

5.2.4 High-Level AM Amplification

The basic tuned-anode vacuum tube power amplifier is described in graphical form in

Figure 5.5

[2]. The tube may be a triode, tetrode, or pentode. Tetrode final amplifiers

are the most common. The RF excitation voltage is applied to the grid of the PA tube,
and the ratio of dc grid bias voltage to peak RF excitation voltage (shown sinusoidal
in

Figure 5.5

) determines the conduction angle

c

of anode current:

c

cc

g

cc

E

E

E

=

-

2 arccos

(5.1)

Figure 5.3 Grid modulation of a class C RF amplifier.

© 1999 CRC Press LLC

Where:
E

g

= exciting grid signal

E

cc

= dc grid bias voltage

The shape of the anode current pulse is determined by the vacuum tube transfer

characteristics and the input waveshape. The pulse of current thus generated is supplied
by the dc power supply E

bb

and passes through the resonant anode tank circuit. The tank

is assumed to have sufficient operating Q to force anode voltage e

p

to be essentially si-

nusoidal and of the same periodic frequency as the RF excitation voltage and resultant
anode current pulse. The instantaneous anode dissipation is the product of instanta-
neous tube anode voltage drop and anode current. The tube transfer characteristic is a
variable, dependent upon device geometry and other factors, including the maximum
drive signal E

g

. The exact shape and magnitude of the current waveform is normally ob-

tained from a load-line plot on the constant-current characteristic tube curves supplied
by the device manufacturer. The resonant anode load impedance is chosen and adjusted
to allow E

p(min)

to be as low as possible without causing excessive screen grid (tetrode

case) or control grid dissipation.

The overall anode efficiency of this circuit can be extended beyond the normal limits

for typical class C amplification by employing a third-harmonic resonator between the
output anode connection and the fundamental resonant circuit. In some cases,
fifth-harmonic resonators are also employed. This technique has the effect of squaring
up the anode voltage waveform e

p

, thus causing the integral of the e

p

× i

p

product (anode

dissipation) to be smaller, resulting in lower dissipation for a given RF power output.
An amplifier employing the third-harmonic anode trap commonly is referred to as a
class C-D stage, suggesting efficiency ranging between conventional class C operation

Figure 5.4 Cathode modulation of a class C RF amplifier.

© 1999 CRC Press LLC

and class D operation. The use of the third-harmonic technique permits efficiencies of
85 percent or more for transmitters of 10 kW power output and above.

Class B Modulator

The most common method of applying low-frequency (typically audio) intelligence
to a high-level amplitude-modulated amplifier is the class B push-pull system illus-
trated in

Figure 5.6

[1]. The vacuum tubes used in such a circuit may be triodes,

tetrodes, or pentodes. The output circuit includes a modulation transformer, audio
coupling capacitor, and dc shunt feed inductor. The capacitor and shunt inductor net-
work is used to prevent unbalanced dc from magnetizing the modulation transformer
core, which would result in poor low-frequency performance. Advanced core materi-
als and improved transformer design have permitted elimination of the coupling ca-

Figure 5.5 Classic vacuum tube class C amplifier: (

a) representative waveforms, (b)

schematic diagram. Conditions are as follows: sinusoidal grid drive, 120° anode current,
resonant anode load. (

After [2].)

(

a)

(

b)

© 1999 CRC Press LLC

pacitor and the shunt reactor in modern systems. The direct current to the modulated
RF amplifier anode, therefore, flows directly through the secondary of the modula-
tion transformer.

Elimination of the coupling capacitor and shunt reactor results in improved perfor-

mance from the modulator circuit. The L and C components effectively form a
three-pole high-pass filter that causes low-frequency transient distortion to be gener-
ated when driven with complex audio waveforms. The simplified circuit constitutes a
single-pole high-pass filter, greatly reducing low-frequency transients.

Even in the improved design, the transient distortion performance of the class B

push-pull modulator can be insufficient for some applications. Complex modulating
waveforms with sharp transitions present special problems. Stray internal winding ca-
pacitances and leakage inductances effectively form a multipole low-pass filter at the
high-frequency end of the audio spectrum. This filter can produce overshoots when
driven with complex waveforms. Such distortion can be reduced by filtering the input
signal, at the cost of somewhat lower high-frequency response.

Negative feedback is used to reduce nonlinear distortion and noise in the push-pull

circuit. Negative feedback, however, usually increases high frequency transient distor-
tion.

Figure 5.6 Class B push-pull modulator stage for a high-level AM amplifier.

© 1999 CRC Press LLC

5.2.5 Pulse Width Modulation

Pulse width modulation (PWM), also known as pulse duration modulation (PDM), is
a common transmission system for modern vacuum tube transmitters.

Figure 5.7

shows the PDM scheme (Harris) applied to an amplitude-modulated PA. The PDM
system works by utilizing a square wave switching system, illustrated in

Figure 5.8

.

The PDM process begins with a signal generator (

Figure 5.9

). A 75 kHz sine wave is

produced by an oscillator and used to drive a square wave generator, resulting in a sim-
ple 75 kHz square wave. The square wave is then integrated, resulting in a triangular
waveform that is mixed with the input audio in a summing circuit. The resulting signal
is a triangular waveform that rides on the incoming signal, typically audio. This com-
posite signal is then applied to a threshold amplifier, which functions as a switch that is
turned on whenever the value of the input signal exceeds a certain limit. The result is a
string of pulses in which the width of the pulse is proportional to the period of time the
triangular waveform exceeds the threshold. The pulse output is applied to an amplifier
to obtain the necessary power to drive subsequent stages. A filter eliminates whatever
transients may exist after the switching process is complete.

The PDM scheme is, in effect, a digital modulation system with the input informa-

tion being sampled at a 75 kHz rate. The width of the pulses contains all the intelli-
gence. The pulse-width-modulated signal is applied to a switch or modulator tube. The
tube is simply turned on, to a fully saturated state, or off in accordance with the instanta-

Figure 5.7 The pulse duration modulation (PDM) method of pulse width modulation.

© 1999 CRC Press LLC

neous value of the pulse. When the pulse goes positive, the modulator tube is turned on
and the voltage across the tube drops to a minimum. When the pulse returns to its mini-
mum value, the modulator tube turns off.

This PDM signal becomes the power supply to the final RF amplifier tube. When the

modulator is switched on, the final amplifier will experience current flow and RF will
be generated. When the switch or modulator tube goes off, the final amplifier current
will cease. This system causes the final amplifier to operate in a highly efficient class D
switching mode. A dc offset voltage to the summing amplifier is used to set the carrier
(no modulation) level of the transmitter.

A high degree of third-harmonic energy will exist at the output of the final amplifier

because of the switching-mode operation. This energy is eliminated by a third-har-
monic trap. The result is a stable amplifier that normally operates in excess of 90 per-
cent efficiency. The power consumed by the modulator and its driver is usually a frac-
tion of a full class B amplifier stage.

Figure 5.8 The principal waveforms of the PDM system.

© 1999 CRC Press LLC

The damping diode shown in

Figure 5.9

is included to prevent potentially damaging

transient overvoltages during the switching process. When the switching tube turns off
the supply current during a period when the final amplifier is conducting, the high cur-
rent through the inductors contained in the PDM filters could cause a large transient
voltage to be generated. The energy in the PDM filter is returned to the power supply by
the damping diode. If no alternative route were established, the energy would return by
arcing through the modulator tube itself.

The PWM system makes it possible to eliminate audio frequency transformers in

the transmitter. The result is wide frequency response and low distortion. Note that
variations on this amplifier and modulation scheme have been used by other manufac-
turers for standard broadcast, shortwave service, and other applications.

5.3

Linear Amplification

The following features are desirable for vacuum tubes used in RF linear amplifier ser-
vice [1]:

·

High power gain

·

Low plate-to-grid capacitance

·

Good efficiency

·

Linear characteristics that are maintained without degradation across the desired
operating band

For linear service, RF amplifiers may be operated in class A, AB

1

, AB

2

, or B modes.

Triode, tetrode, or pentode devices may be used, either grid- or cathode-driven. The
choice of mode, tube, and driving method depends upon the operational specifica-
tions of each application.

Figure 5.9 Block diagram of a PDM waveform generator.

© 1999 CRC Press LLC

5.3.1 Device Selection

The triode tube, having a large plate-to-grid interelectrode capacitance, requires neu-
tralization in grid-driven linear service to prevent oscillation [1]. A triode having a
low amplification factor is suitable for class AB and AB

2

grid-driven operation. The

RF grid excitation voltage for this type of service will be high; grid excursions into
the positive region are normal for class AB

2

. A swamping resistor should be used

across the input tuned circuit to maintain a constant input impedance to the stage and
to provide for stability. With a low value of swamping resistance, the grid current
drawn is only a small part of the total grid load, and the driver load impedance is rela-
tively constant. The swamping resistor improves RF stability by providing a low im-
pedance to ground for regenerative feedback through the plate-to-grid capacitance.

High-amplification-factor triodes perform exceptionally well in circuits where the

grid is grounded and the cathode is driven. Under these conditions, the control grid acts
as a shield between the input and output circuits. Neutralization, therefore, is not nor-
mally required. Zero-bias triodes operate in the class AB

2

mode and require only fila-

ment, plate, and drive power. For optimum linear operation, a tuned circuit is placed in
the cathode RF return path to maintain a sinusoidal waveshape over the drive cycle. The
tuned circuit will reduce the intermodulation distortion produced by the amplifier and
also will reduce drive power requirements.

If the driver and PA stages are located close to each other, the tuned cathode circuit

can be a part of the output circuit of the driver. If, however, the amplifiers are far re-
moved and coupled by a length of coaxial cable, it is recommended that a tuned cathode
circuit with a Q of between 2 and 4 be used.

Most tetrode and pentode amplifiers are designed to be grid-driven to take advan-

tage of the high power gain of these devices.

In all linear amplifier systems, the driver output impedance should be kept low be-

cause of the nonlinear input loading characteristics of the amplifier tube as it ap-
proaches maximum power. The lower the driver amplifier impedance, the smaller the
effect of the nonlinear input loading.

5.3.2 Grid-Driven Linear Amplifier

A linear amplifier utilizing a tetrode or pentode is usually grid-driven to take advan-
tage of the inherent high gain of the tube [1]. Such an amplifier can be driven into the
grid-current region under the proper circumstances. In any case, the input circuit will
be loaded by the tube grid. For the no-grid-current case, the driver will work into the
input conductance loading; for the grid-current case, the driver will work into the in-
put conductance loading plus grid-current loading. It is therefore desirable (and nec-
essary in the grid-current case) to swamp the input circuit with an appropriate
noninductive resistor. The resistor will maintain an almost constant load on the driver
and minimize the effects of any nonlinearity in grid loading.

© 1999 CRC Press LLC

5.3.3 Cathode-Driven Linear Amplifier

The cathode-driven amplifier may use triode, tetrode, or pentode tubes [1]. The drive
signal is applied to the cathode in this class of operation. The cathode-driven ampli-
fier is particularly suitable for high-power stages using high-

µ triodes in the HF and

VHF region. This class of operation normally eliminates the need for neutralization
because the control grid screens the plate from the input circuit. The power gain for
suitable triode class AB cathode-driven amplifiers is on the order of 7 to 20. The ac-
tual tube power gain is approximately equal to the ratio of RF plate voltage to RF
cathode voltage. This relationship is true because the fundamental component of the
plate current is common to the input and output circuits.

Tetrode tubes also can be used in cathode-driven operation. Power gain is consider-

ably higher than that of the triode, on the order of 20 to 50. It is important to note that
screen grid current loads the input circuit, just as control-grid current does.

For an amplifier located some distance from the driver, an improvement in

intermodulation distortion can be realized by tuning the cathode circuit. When the
driver is located close to the amplifier (0.1 wavelength or so) other means may be used
to minimize the nonlinear loading of the cathode-driven stage.

5.3.4 Intermodulation Distortion

When an RF signal with varying amplitude is passed through a nonlinear device,
many new products are produced [1]. (Intermodulation distortion is discussed in de-
tail in Section 10.3.6.) The frequency and amplitude of each component can be deter-
mined mathematically because the nonlinear device can be represented by a power se-
ries expanded about the zero-signal operating point. The example of a typical
two-tone signal serves to summarize this mathematical relationship. Assume that two
equal-amplitude test signals (f

1

= 2.001 MHz, and f

2

= 2.003 MHz) are applied to a lin-

ear amplifier. The output spectrum of the amplifier is shown in

Figure 5.10

. Many of

the distortion products fall outside the passband of the amplifier tuned circuits. If no
impedance exists at the frequencies of the distortion components, then no voltage can
be developed. Further study of this spectrum discloses that no even-order products
fall near the two desired signals. Some odd-order products, however, fall near the de-
sired frequencies and possibly within the passband of the tuned circuits.

The distortion products usually given in tube data sheets are the third- and fifth-or-

der intermodulation distortion products that can fall within the amplifier passband.
Using the same f

1

and f

2

frequencies of the previous example, the frequencies of the

third-order products are:

2f

1

f

2

= 1.999 MHz

2f

2

f

1

= 2.005 Mhz

The frequencies of the fifth-order products are:

© 1999 CRC Press LLC

Figure 5.10

Spectrum at the output of a nonlinear device with an input of two

equal-amplitude sine waves of

f

1

= 2.001 MHz and

f

2

= 2.003 MHz.

© 1999 CRC Press LLC

3f

1

2f

2

= 1.997 MHz

3f

2

2f

1

= 2.007 MHz

These frequencies are well within the passband of a tuned circuit intended to pass
voice frequencies and, therefore, power will be delivered to the antenna at these fre-
quencies. All intermodulation distortion power is wasted and serves no purpose other
than to cause interference to adjacent channels.

Intermodulation distortion in a power amplifier tube is the result of variations of the

transfer characteristics of the device from the ideal case. An ideal transfer characteristic
curve is shown in

Figure 5.11

. Even-order products do not contribute to the

intermodulation distortion problem because they fall outside the amplifier passband.
Therefore, if the transfer characteristic produces an even-order curvature at the
small-signal end of the curve (from point A to point B) and the remaining portion of the
curve (point B to point C) is linear, the tube is considered to have an ideal transfer char-
acteristic. If the operating point of the amplifier is set at point 0 (located midway hori-
zontally between point A and point B), there will be no distortion in a class AB ampli-
fier. However, no tube has this idealized transfer characteristic. It is possible, by manip-
ulation of the electron ballistics within a given tube structure, to alter the transfer char-
acteristic and minimize the distortion products.

5.4

High-Efficiency Linear Amplification

The class B RF amplifier is the classic means of achieving linear amplification of an
AM signal. A significant efficiency penalty, however, is paid for linear operation. The

Figure 5.11 Ideal grid-plate transfer curve for class AB operation.

© 1999 CRC Press LLC

plate efficiency at carrier level in a practical linear amplifier is about 33 percent,
while a high-level (plate)-modulated stage can deliver a basic efficiency of 65 to 80
percent. In a real-world transmitter, power consumed by the modulator stage must
also be taken into account. However, because the output power of the modulator is, at
most, one-third of the total power output, the net efficiency of the plate-modulated
amplifier is still higher than a conventional linear amplifier of comparable power.

Still, linear amplification is attractive to the users of high-power transmitters be-

cause low-level modulation can be employed. This eliminates the need for modulation
transformers and reactors, which--at certain power levels and for particular applica-
tions--may be difficult to design and expensive to construct. To overcome the effi-
ciency penalty of linear amplification, special systems have been designed that take ad-
vantage of the benefits of a linear mode, while not imposing an excessive efficiency
burden.

5.4.1 Chireix Outphasing Modulated Amplifier

The Chireix amplifier employs two output stage tubes. The grids of the tubes are
driven with signals whose relative phase varies with the applied modulating signal.
The output waveforms of the tubes are then applied to a summing network, and fi-
nally to the load.

Figure 5.12

shows a simplified schematic diagram.

Operation of the circuit is straightforward. When the grids of the tubes are driven

with signals that are 180° out of phase, power output will be zero. This corresponds to a
negative AM modulation peak of 100 percent. Full (100 percent) positive modulation
occurs when both grids are driven with signals that are in phase. The AM modulating
signal is produced by varying the relative phase of the drive signals.

The phase-modulated carrier-frequency grid waveforms are generated in a

low-level balanced modulator and amplified to the level required by the PA tubes. Sta-
bility of the exciter in the Chireix system is critical. Any shift in the relative phase of the
exciter outputs will translate to amplitude modulation of the transmitted waveform.

Because of the outphasing method employed to produce the AM signal, the power

factor associated with the output stage is of special significance. The basic modulating
scheme would produce a unity power factor at zero output. The power factor would then
decrease with increasing RF output. To avoid this undesirable characteristic, the output
plate circuits are detuned, one above resonance and one below, to produce a specified
offset. This dephasing, coupled with adjustment of the relative phase of the driving sig-
nals, maintains the power factor within a reasonable range.

There are, however, two major disadvantages to this system. First, the efficiency of

the amplifier is compromised because at all instantaneous levels of modulation except
one, the anode circuits must work into a reactive load. Typical overall efficiency is on
the order of 60 percent. Second, output carrier power adjustment is sensitive to tuning
of any stage in the chain. This problem can be reduced by employing broadband ampli-
fying circuits at lower-level stages.

© 1999 CRC Press LLC

5.4.2 Doherty Amplifier

The Doherty modulated amplifier has seen considerable use in high-efficiency linear
AM transmitters. The system is a true high-efficiency linear amplifier, rather than a
hybrid amplifier/modulator. The Doherty amplifier employs two tubes and a 90° net-
work as an impedance inverter to achieve load-line modification as a function of
power level. The basic circuit is shown in

Figure 5.13

.

Two tubes are used in the Doherty amplifier, a carrier tube (V1) and a peak tube

(V2). The carrier tube is biased class B. Loading and drive for the tube are set to provide

Figure 5.12 Simplified schematic diagram of the Chireix outphasing modulated ampli-
fier: (

a) output circuit, (b) drive signal generation circuit.

(

a)

(

b)

© 1999 CRC Press LLC

maximum linear voltage swing at carrier level. The peak tube is biased class C; at car-
rier level it just begins to conduct plate current. Each tube delivers an output power
equal to twice the carrier power when working into a load impedance of R

. At carrier,

the reflected load impedance at the plate of V1 is equal to 2R

. This is the correct value

of load impedance for carrier-level output at full plate voltage swing. (The imped-
ance-inverting property of the 90

° network is similar to a 1/4-wave transmission line.)

A 90° phase lead network is included in the grid circuit of V1 to compensate for the

90° phase lag produced by the impedance-inverting network in the plate circuit. For
negative modulation swings, the carrier tube performs as a linear amplifier with a load
impedance of 2R

; the peak tube contributes no power to the output.

On positive modulation swings, the peak tube conducts and contributes power to the

R/2 load resistance. This is equivalent to connecting a negative resistance in shunt with
the R/2 load resistance, so that the value seen at the load end of the 90° network in-

Figure 5.13 Doherty high-efficiency AM amplifier: (

a) operating theory, (b) schematic

diagram of a typical system.

(

a)

(

b)

© 1999 CRC Press LLC

creases. This increase is reflected through the network as a decrease in load resistance
at the plate of V1, causing an increase in output current from the tube, and an increase in
output power. The drive levels on the tubes are adjusted so that each contributes the
same power (equal to twice the carrier power) at a positive modulation peak. For this
condition, a load of R

is presented to each tube.

The key aspect of the Doherty circuit is the change in load impedance on tube V1

with modulation. This property enables the device to deliver increased output power at
a constant plate voltage swing. The result is high efficiency and good linearity. Overall
efficiency of the Doherty amplifier ranges from 60 to 65 percent.

A practical application of the Doherty circuit is shown in

Figure 5.13b

. The shunt

reactances of the phase-shift networks are achieved by detuning the related tuned cir-
cuits. The tuned circuits in the grid are tuned above the operating frequency; those in
the plate are tuned below the operating frequency.

The Doherty amplifier has two important advantages in high-power applications.

First, the peak anode voltage at either tube is (approximately) only one-half that re-
quired for an equivalent carrier power high-level pulse width modulation (PWM) or
class B anode-modulated transmitter, thus significantly increasing reliability and us-
able tube life. Second, no large modulation transformer or special filtering components
are used in the final amplifier stages that might cause transient overshoot distortion.

Disadvantages of the Doherty amplifier include nonlinear distortion performance

and increased complexity of tuning. The major sources of nonlinear distortion include
the nonlinearity of the carrier tube at or near the 100 percent negative modulation crest,
and the nonlinearity of the peak tube at or near carrier level when it is just beginning to
conduct. Both sources of distortion may be effectively reduced by using moderate
amounts of overall envelope feedback.

5.4.3 Screen-Modulated Doherty-Type Amplifier

A variation on the basic Doherty scheme can be found in the screen-modulated power
amplifier. The design is unique in that the system does not have to function as a linear
amplifier. Instead, modulation is applied to the screen grids of both the carrier and
peak tubes. The peak tube is modulated upward during the positive half of the modu-
lation cycle, and the carrier tube is modulated downward during the negative half of
the cycle. This technique results in performance improvements over the classic
Doherty design in that RF excitation voltages and modulating voltages are isolated
from each other, thereby eliminating a troublesome source of tuning-vs.-modulation
interaction.

The basic screen modulation system is shown in

Figure 5.14

. The peak and carrier

tubes are biased and driven in quadrature as class C amplifiers from the continuous
wave RF drive source. At carrier level, the screen voltage of the carrier tube is adjusted
so that the carrier device is near anode saturation and delivering approximately 96 per-
cent of the carrier power. The screen voltage of the peak tube is adjusted so that it is just
into conduction, and is supplying the remaining 4 percent (approximately) of carrier
power. The combined anode efficiency at carrier level is greater than 77 percent.

© 1999 CRC Press LLC

Modulation of the carrier occurs when the screen voltage of the peak tube begins to

rise during the positive modulation half-cycle, thus causing the peak tube to supply
more RF current to the output load. This increase of current into the output network
causes the resistance seen by the interanode network to increase and, because of the im-
pedance-inverting characteristics of the 90° interanode network, causes a proportional
decrease in the load impedance presented to the carrier tube.

The carrier-tube resonant anode voltage drop is fully saturated over the entire posi-

tive modulation half-cycle, and therefore is effectively a constant voltage source. The
power output of the carrier tube thus increases during the positive modulation half-cy-
cle, caused by the modulated decreasing impedance at its anode, until both peak and
carrier tubes deliver twice carrier power at the 100 percent positive modulation crest.
During the negative modulation half-cycle, the peak tube is held out of conduction,
while the carrier-tube output voltage decreases linearly to zero output at the 100 percent
modulation trough.

A screen-modulated amplifier designed for MF operation at 1 MW is shown in Fig-

ure 5.15. The carrier tube delivers all of the 1 MW carrier power to the load through the
90° impedance-inverting network. When modulation is applied to the screen of the car-
rier tube (during the negative half-cycle), the carrier tube output follows the modulation
linearly. Because the carrier tube is driven to full swing, its voltage excursion does not
increase during the positive half-cycle. Positive modulation is needed on the screen
only to maintain the full plate swing as the impedance load changes on the carrier tube.

The peak tube normally is operated with a negative screen voltage that maintains

plate current at near cutoff. During the positive peak of modulation, the screen of the
peak tube is modulated upward. As the peak tube is modulated positive, it delivers
power into the output circuit. As the delivered power increases, the load seen by the
interplate network changes as described for the general case.

The advantages of screen modulation are the same as those identified for the

Doherty linear amplifier, plus higher efficiency at all depths of modulation. Further-

Figure 5.14 Screen-modulated Doherty-type circuit.

© 1999 CRC Press LLC

more, the screen modulation system is less sensitive to misadjustment of RF amplifier
tuning.

5.4.4 Terman-Woodyard Modulated Amplifier

The Terman-Woodyard configuration uses the basic scheme of Doherty for achieving
high efficiency (the impedance-inverting property of a 90° phase-shift network).
However, the Terman-Woodyard design also employs grid modulation of both the car-
rier tube and the peak tube, allowing both tubes to be operated class C. The result is an
increase in efficiency over the Doherty configuration.

The basic circuit is shown in

Figure 5.16

. With no modulation, V1 operates as a class

C amplifier, supplying the carrier power, and V2 is biased so that it is just beginning to
conduct. The efficiency at carrier is, therefore, essentially that of a class C amplifier.

During positive modulation swings, V2 conducts. At 100 percent modulation, both

tubes supply equal amounts of power to the load, as in the Doherty design. During nega-
tive modulation swings, V2 is cut off, and V1 performs as a standard grid-modulated
amplifier.

The Terman-Woodyard system rates efficiency as a function of modulation percent-

age for sinusoidal waveforms. Typical efficiency at carrier is 80 percent, falling to a
minimum of 68 percent at 50 percent modulation and 73 percent at 100 percent modu-
lation.

Figure 5.15 Practical implementation of a screen-modulated Doherty-type 1 MW
power amplifier (Continental Electronics).

© 1999 CRC Press LLC

5.4.5 Dome Modulated Amplifier

The Dome high-efficiency amplifier employs three power output tubes driven by dif-
ferent input signals. Modulation is achieved by load-line modification during positive
modulation excursions, and by linear amplification during negative modulation
swings. Load-line modification is achieved by absorption of a portion of the gener-
ated RF power. However, most of the absorbed power is returned to the plate power
supply, rather than being dissipated as heat. High efficiency is the result.

A basic Dome modulated amplifier is shown in

Figure 5.17

. Tube V1 is used in a

plate-modulated configuration, supplying power to the grid of V2 (the power amplifier
tube). The load impedance in the plate circuit of V2 is equivalent to the load impedance
reflected into the primary of transformer T1 (from the antenna) in series with the im-
pedance appearing across the C8 terminals of the 90° phase-shift network consisting of
C8, C9, and L4. The impedance appearing across the C8 terminals of the 90° network is
inversely related to the impedance across the other terminals of the network (in other
words, the effective ac impedance of V3). Thus, with V3 cut off, a short circuit is re-
flected across the C8 terminals of the network. Tube V3 performs as a modulated recti-
fier with intelligence supplied to its grid. This tube is referred to as the modifier.

With no modulation (the carrier condition), tube V3 is biased to cutoff, and the drive

to V2 is adjusted until power output is equal to four times the carrier power (corre-
sponding to a positive modulation peak). Note that all the plate signal voltage of V2 ap-
pears across the primary of transformer T1 for this condition. The bias on V3 is then re-
duced, lowering the ac impedance of the tube and, therefore, reflecting an increasing

Figure 5.16 Terman-Woodyard high-efficiency modulated amplifier.

© 1999 CRC Press LLC

impedance at the C8 terminals of the 90° network. The bias on V3 is reduced until the
operating carrier power is delivered to the antenna.

An amount of power equal to the carrier is thus rectified by V3 and returned to the

plate power supply, except for that portion dissipated on the V3 plate. The drive level to
V2 is finally adjusted so that the tube is just out of saturation.

For positive modulation excursions, V3 grid voltage is driven negative, reaching

cutoff for a positive modulation peak. For negative modulation swings, V2 acts as a lin-
ear amplifier. V3 does not conduct during the negative modulation swing because the
peak RF voltage on the plate is less than the dc supply voltage on the cathode.

Because of the serial loss incurred in the power amplifier tube and in the modifier

tube for energy returned to the power supply, the Dome circuit is not as efficient as the
Doherty configuration. Typical efficiency at carrier ranges from 55 to 60 percent.

5.5

Television Power Amplifier Systems

Although VSBAM is used for a number of varied applications, TV transmission is the
most common. A TV transmitter is divided into two basic subsystems:

·

The visual section, which accepts the video input, amplitude-modulates an RF
carrier, and amplifies the signal to feed the antenna system

Figure 5.17 Dome high-efficiency modulated amplifier.

© 1999 CRC Press LLC

·

The aural section, which accepts the audio input, frequency-modulates a separate
RF carrier, and amplifies the signal to feed the antenna system

The visual and aural signals usually are combined to feed a single radiating antenna.
Different transmitter manufacturers have different philosophies with regard to the de-
sign and construction of a transmitter. Some generalizations are possible, however,
with respect to basic system configurations. Transmitters can be divided into catego-
ries based on the following criteria:

·

Output power

·

Final-stage design

·

Modulation system

5.5.1 System Considerations

When the power output of a TV transmitter is discussed, the visual section is the pri-
mary consideration. Output power refers to the peak power of the visual stage of the
transmitter (peak of sync). The FCC-licensed ERP is equal to the transmitter power
output times feedline efficiency times the power gain of the antenna.

A VHF station can achieve its maximum power output through a wide range of

transmitter and antenna combinations. Reasonable pairings for a high-band VHF sta-
tion range from a transmitter with a power output of 50 kW feeding an antenna with a
gain of 8, to a 30 kW transmitter connected to a gain-of-12 antenna. These combina-
tions assume reasonably low feedline losses. To reach the exact power level, minor ad-
justments are made to the power output of the transmitter, usually by a front-panel
power control.

UHF stations that want to achieve their maximum licensed power output are faced

with installing a very high power transmitter. Typical pairings include a transmitter
rated for 220 kW and an antenna with a gain of 25, or a 110 kW transmitter and a
gain-of-50 antenna. In the latter case, the antenna could pose a significant problem.
UHF antennas with gains in the region of 50 are possible, but not advisable for most in-
stallations because of coverage problems that can result.

The amount of output power required of the transmitter will have a fundamental ef-

fect on system design. Power levels dictate the following parameters:

·

Whether the unit will be of solid-state or vacuum tube design

·

Whether air, water, or vapor cooling must be used

·

The type of power supply required

·

The sophistication of the high-voltage control and supervisory circuitry

·

Whether common amplification of the visual and aural signals (rather than sepa-
rate visual and aural amplifiers) is practical

Tetrodes generally are used for VHF transmitters above 25 kW, and specialized

tetrodes can be found in UHF transmitters at the 15 kW power level and higher. As

© 1999 CRC Press LLC

solid-state technology advances, the power levels possible in a reasonable transmitter
design steadily increase, making solid-state systems more attractive options.

In the realm of UHF transmitters, the klystron (and its related devices) reigns su-

preme. Klystrons use an electron-bunching technique to generate high power--55 kW
from a single tube is not uncommon--at ultrahigh frequencies. They are currently the
first choice for high-power, high-frequency service. Klystrons, however, are not partic-
ularly efficient. A stock klystron with no special circuitry might be only 40 percent effi-
cient. Various schemes have been devised to improve klystron efficiency, the best
known of which is beam pulsing. Two types of pulsing are in common use:

·

Mod-anode pulsing, a technique designed to reduce power consumption of the
device during the color burst and video portion of the signal (and thereby improve
overall system efficiency)

·

Annular control electrode (ACE) pulsing, which accomplishes basically the
same goal by incorporating the pulsing signal into a low-voltage stage of the
transmitter, rather than a high-voltage stage (as with mod-anode pulsing)

Experience has shown the ACE approach--and other similar designs--to provide
greater improvement in operating efficiency than mod-anode pulsing, and better reli-
ability as well.

Several newer technologies offer additional ways to improve UHF transmitter effi-

ciency, including:

·

The inductive output tube (IOT), also known as the Klystrode. This device essen-
tially combines the cathode/grid structure of the tetrode with the drift tube/collec-
tor structure of the klystron. (The Klystrode tube is a registered trademark of
Varian Associates.)

·

The multistage depressed collector (MSDC) klystron, a device that achieves
greater efficiency through a redesign of the collector assembly. A multistage col-
lector is used to recover energy from the electron stream inside the klystron and
return it to the beam power supply.

Improved tetrode devices, featuring higher operating power at UHF and greater effi-
ciency, have also been developed.

A number of approaches may be taken to amplitude modulation of the visual carrier.

Current technology systems utilize low-level intermediate-frequency (IF) modulation.
This approach allows superior distortion correction, more accurate vestigial sideband
shaping, and significant economic advantages to the transmitter manufacturer.

A TV transmitter can be divided into four major subsystems:

·

The exciter

·

Intermediate power amplifier (IPA)

·

Power amplifier

·

High-voltage power supply

Figure 5.18

shows the audio, video, and RF paths for a typical design.

© 1999 CRC Press LLC

5.5.2 Power Amplifier

The power amplifier raises the output energy of the transmitter to the required RF op-
erating level. Tetrodes in TV service are usually operated in the class B mode to ob-
tain reasonable efficiency while maintaining a linear transfer characteristic. Class B

Figure 5.18 Basic block diagram of a TV transmitter. The three major subassemblies
are the exciter, IPA, and PA. The power supply provides operating voltages to all sec-
tions, and high voltage to the PA stage.

© 1999 CRC Press LLC

amplifiers, when operated in tuned circuits, provide linear performance because of
the flywheel effect of the resonance circuit. This allows a single tube to be used in-
stead of two in push-pull fashion. The bias point of the linear amplifier must be cho-
sen so that the transfer characteristic at low modulation levels matches that at higher
modulation levels. Even so, some nonlinearity is generated in the final stage, requir-
ing differential gain correction. The plate (anode) circuit of a tetrode PA usually is
built around a coaxial resonant cavity, providing a stable and reliable tank.

UHF transmitters using a klystron in the final output stage must operate class A, the

most linear but also most inefficient operating mode for a vacuum tube. The basic effi-
ciency of a nonpulsed klystron is approximately 40 percent. Pulsing, which provides
full available beam current only when it is needed (during peak of sync), can improve
device efficiency by as much as 25 percent, depending on the type of pulsing used.

Two types of klystrons are presently in service:

·

Integral-cavity klystron

·

External-cavity klystron

The basic theory of operation is identical for each tube, but the mechanical approach
is radically different. In the integral-cavity klystron, the cavities are built into the kly-
stron to form a single unit. In the external-cavity klystron, the cavities are outside the
vacuum envelope and bolted around the tube when the klystron is installed in the
transmitter.

A number of factors come into play in a discussion of the relative merits of integral-

vs. external-cavity designs. Primary considerations include operating efficiency, pur-
chase price, and life expectancy.

The PA stage includes a number of sensors that provide input to supervisory and

control circuits. Because of the power levels present in the PA stage, sophisticated
fault-detection circuits are required to prevent damage to components in the event of a
problem inside or outside the transmitter. An RF sample, obtained from a directional
coupler installed at the output of the transmitter, is used to provide automatic
power-level control.

The transmitter system discussed in this section assumes separate visual and aural

PA stages. This configuration is normally used for high-power transmitters. A com-
bined mode also may be used, however, in which the aural and visual signals are added
prior to the PA. This approach offers a simplified system, but at the cost of additional
precorrection of the input video signal.

PA stages often are configured so that the circuitry of the visual and aural amplifiers

is identical, providing backup protection in the event of a visual PA failure. The aural
PA can then be reconfigured to amplify both the aural and the visual signals, at reduced
power.

The aural output stage of a TV transmitter is similar in basic design to an FM broad-

cast transmitter. Tetrode output devices generally operate class C, providing good effi-
ciency. Klystron-based aural PAs are used in UHF transmitters.

© 1999 CRC Press LLC

Application Example

A 60 kW transmitter is shown in block diagram form in

Figure 5.19

. A single

high-power klystron is used in the visual amplifier, and another is used in the aural
amplifier. The tubes are driven from solid-state intermediate power amplifier mod-
ules. The transmitter utilizes ACE-type beam control, requiring additional
predistortion to compensate for the nonlinearities of the final visual stage.
Predistortion is achieved by correction circuitry at an intermediate frequency in the
modulator. Both klystrons are driven from the output of a circulator, which ensures a
minimum of driver-to-load mismatch problems.

A block diagram of the beam modulator circuit is shown in

Figure 5.20

. The system

receives input signals from the modulator, which synchronizes ACE pulses to the visual
tube with the video information. The pulse waveform is developed through a pulse am-
plifier, rather than a switch. This permits more accurate adjustments of operating con-
ditions of the visual amplifier.

Although the current demand from the beam modulator is low, the bias is near cath-

ode potential, which is at a high voltage relative to ground. The modulator, therefore,
must be insulated from the chassis. This is accomplished with optical transmitters and
receivers connected via fiber optic cables. The fiber optic lines carry supervisory, gain
control, and modulating signals.

The four-cavity external klystrons will tune to any channel in the UHF-TV band. An

adjustable beam perveance feature enables the effective electrical length of the device
to be varied by altering the beam voltage as a function of operating frequency (see Sec-
tion 6.3). Electromagnetic focusing is used on both tubes. The cavities, body, and gun

Figure 5.19 Schematic diagram of a 60 kW klystron-based TV transmitter.

© 1999 CRC Press LLC

areas of the klystrons are air-cooled. The collectors are vapor-phase-cooled using an
external heat exchanger system.

The outputs of the visual and aural klystrons are passed through harmonic filters to

an RF combiner before being applied to the antenna system.

5.6

FM Power Amplifier Systems

The amplitude of an FM signal remains constant with modulation so that efficient
class B and C amplifiers can be used.

1

Two basic circuit types have evolved:

·

Amplifiers based on a tetrode or pentode in a grid-driven configuration

Figure 5.20 Schematic diagram of the modulator section of a 60 kW TV transmitter.

1

This section was contributed by Geoffrey N. Mendenhall, P. E., Cincinnati, OH, and Warren
B. Bruene, P.E., Dallas, TX.

© 1999 CRC Press LLC

·

Amplifiers based on a high-

µtriode or tetrode device in a cathode-driven configu-

ration (grounded grid)

High-power FM amplifiers operating in the VHF range typically employ resonant
cavity systems. The basic configurations of FM amplifiers are discussed in this sec-
tion, and amplifiers utilizing cavity resonators are discussed in Section 4.3.

5.6.1 Cathode-Driven Triode Amplifier

The characteristics of high-

µ triodes are well adapted to FM power amplifier service.

The grounded-grid circuit is simple, and no screen or grid bias power supplies are re-
quired.

Figure 5.21

shows the basic circuit configuration. In this case, the grid is con-

nected directly to chassis ground. The difference between dc cathode current and dc
plate current is dc grid current. The output tank is a shorted coaxial cavity that is ca-
pacitively loaded by the tube output and stray circuit capacitance. A small capacitor is
used for trimming the tuning, and another small variable capacitor is used for adjust-
ing the loading. A pi network is used to match the 50

input to the tube cathode input

impedance.

Figure 5.21 Cathode-driven triode power amplifier.

© 1999 CRC Press LLC

The triode usually is operated in the class B mode to achieve maximum power gain,

which is on the order of 20 (13 dB). The device can be driven into class C operation by
providing negative grid bias or positive cathode bias. This increases the plate effi-
ciency, but also requires increased drive power.

Most of the drive into a grounded-grid amplifier is fed through the tube and appears

at the output. This increases the apparent efficiency so that the efficiency factor given
by the transmitter manufacturer may be higher than the actual plate efficiency of the
tube. The true plate efficiency is determined from the following:

(

)

(

)

Eff

P

I

E

P

p

O

p

p

in

=

×

+

×

(5.2)

Where:
Eff

p

= actual plate efficiency

P

o

= measured output power

I

p

= dc plate input current to final stage

E

p

= dc plate input voltage to final stage

P

in

= RF input drive power

= fraction of grid-to-plate transfer (0.9 typical)

Because most of the drive power is fed through the tube, any changes in loading of the
output circuit also will affect the input impedance and the driver stage.

Because there is RF drive voltage on the cathode (filament), some means of decoup-

ling must be used to block the signal from the filament transformer. One method em-
ploys high-current RF chokes; another feeds the filament power through the input tank
circuit inductor.

5.6.2 Grounded-Grid vs. Grid-Driven Tetrode

Tetrodes also may be operated in the grounded-grid configuration by placing both the
control grid and the screen grid at RF ground. Higher efficiency and gain can be
achieved by placing negative bias on the control grid while placing a positive voltage
on the screen grid of a cathode-driven tetrode.

The input capacitance of a tetrode in a grounded-grid configuration is much less

than in a grid-driven configuration, and the input impedance is lower, providing wider
bandwidth. The approximate input capacitances of some common tube types are listed
in

Table 5.1

.

The typical drive power requirements of a high-gain tetrode, as a function of plate

voltage, for a 5 kW power amplifier operating at 100 MHz are as follows:

·

Grounded-grid configuration: 340 W input at 4.5 kV plate voltage, 280 W at 5.2
kV

·

Grid-driven configuration: 190 W at 4.5 kV plate voltage, 140 W at 5.2 kV

© 1999 CRC Press LLC

There are several tradeoffs in performance between the grounded-grid and the

grid-driven modes of a tetrode PA with respect to gain, efficiency, amplitude band-
width, phase bandwidth, and incidental AM under equivalent operating conditions:

·

When the PA is driven into saturation, its bandwidth is limited by the output cav-
ity bandwidth in the grounded-grid amplifier. The PA bandwidth in the
grid-driven amplifier is limited by the input circuit Q.

·

Ouput bandwidth under saturation can be improved in either configuration by re-
ducing the plate voltage. This involves a tradeoff in efficiency with a smaller volt-
age swing. The bandwidth improvement can be obtained with a loss of PA gain
and efficiency.

·

A grounded-grid saturated PA improves bandwidth over a grid-driven saturated
PA at the expense of amplifier gain. Best performance for FM operation is ob-
tained when the amplifier is driven into saturation where little change in output
power occurs with increasing drive power. Maximum efficiency also occurs at
this point.

·

The phase linearity in the 0.5 dB bandwidth of the amplifier is better in a
grid-driven configuration. The class C grounded-grid PA exhibits a more nonlin-
ear phase slope within the passband, yet has a wider amplitude bandwidth. This
phenomenon is the result of interaction of the input and output circuits because
they are effectively connected in series in the grounded-grid configuration. The
neutralized grid-driven PA provides more isolation between these networks, so
they behave more like independent filters.

5.6.3 Grid-Driven Tetrode/Pentode Amplifiers

RF generators using tetrode amplifiers throughout usually have one less stage than
those using triodes. Because tetrodes have higher power gain, they are driven into
class C operation for high plate efficiency. Countering these advantages is the re-
quirement for neutralization, along with screen and bias power supplies.

Figure 5.22

Table 5.1 Approximate Input Capacitance of Common Power Grid Tubes Used in
Grounded-Grid and Grounded-Cathode Service

Tube Type

Gounded Grid

Grounded Cathode

4CX3000A

67

140

4CX3500A

59

111

4CX5000A

53

115

4CX15,000A

67

161

4CX20,000A/8990

83

190

© 1999 CRC Press LLC

shows a schematic of a grid-driven tetrode amplifier. In this example, the screen is op-
erated at dc ground potential, and the cathode (filament) is operated below ground by
the amount of screen voltage required (grounded-screen operation). The advantage of
this configuration is that stability problems resulting from undesired resonances in
the screen bypass capacitors are eliminated. With this arrangement, however, the
screen supply must be capable of handling the combined plate and screen currents.

With directly heated tubes, it is necessary to use filament bypass capacitors. During

grounded-screen operation, these bypass capacitors must have a higher breakdown
voltage rating (relative to grounded-grid operation) because they will have the dc
screen voltage across them. The filament transformer must have additional insulation
to withstand the dc screen voltage. The screen power supply provides a negative voltage
in series with the cathode-to-ground, and must have the additional capacity to handle
the sum of the plate and screen currents. A coaxial cavity is used in the output circuit so
that the circulating current is spread over large surfaces, keeping the losses low. This
cavity is a shorted 1/4-wavelength transmission line section that resonates the tube out-
put capacitance. The length is preset to the desired carrier frequency, and a small-value
variable capacitor is used to trim the system to resonance. (Amplifiers utilizing cavity
resonator systems are discussed in Section 4.3.)

Pentode amplifiers have even higher gain than their tetrode counterparts. The circuit

configuration and bias supply requirements for the pentode are similar to the tetrode's
because the third (suppressor) grid is tied directly to ground. The additional isolating

Figure 5.22 Grid-driven grounded-screen tetrode power amplifier.

© 1999 CRC Press LLC

effect of the suppressor grid eliminates the need for neutralization in the pentode ampli-
fier.

5.6.4 Impedance Matching into the Grid

The grid circuit usually is loaded (swamped) with added resistance. The purpose of
this resistance is to broaden the bandwidth of the circuit by lowering the circuit Q and
to provide a more constant load to the driver. The resistance also makes neutralization
less critical so that the amplifier is less likely to become unstable with varying output
circuit loading.

Cathode and filament lead inductance from inside the tube, through the socket and

filament bypass capacitors to ground, can increase grid driver requirements. This effect
is the result of RF current flowing from grid to filament through the tube capacitance
and then through the filament lead inductance to ground. An RF voltage is developed
on the filament that, in effect, causes the tube to be partly cathode-driven. This undesir-
able extra drive power requirement can be minimized by series resonating the cathode
return path with the filament bypass capacitors or by minimizing the cath-
ode-to-ground inductance by using a tube socket incorporating thin-film dielectric
sandwich capacitors for coupling and bypassing.

High-power grid-driven class C amplifiers require a swing of several hundred RF

volts on the grid. To develop this high voltage swing, the input impedance of the grid
must be increased by the grid input matching circuit. Because the capacitance between
the grid and the other tube elements may be 100 pF or more, the capacitive reactance at
100 MHz, for example, will be very low unless the input capacitance is parallel reso-
nated with an inductor. Bandwidth can be maximized by minimizing any additional cir-
cuit capacitance and using a portion of the tube input capacitance as part of the imped-
ance-transformation network.

Figure 5.23

illustrates two popular methods of resonat-

ing and matching into the grid of a high-power tube. Both schemes can be analyzed by
recognizing that the desired impedance transformation is produced by an equivalent L
network.

In

Figure 5.23a

, a variable inductor (L

in

) is used to raise the input reactance of the

tube by bringing the tube input capacitance (C

in

) almost to parallel resonance. Parallel

resonance is not reached because a small amount of parallel capacitance (C

p

) is re-

quired by the equivalent L network to transform the high impedance (Z

in

) of the tube

down to a lower value through the series matching inductor (L

s

). This configuration has

the advantage of providing a low-pass filter by using part of the tube's input capaci-
tance to form C

p

.

Figure 5.23b

uses variable inductor L

in

to take the input capacitance (C

in

) past paral-

lel resonance so that the tube's input impedance becomes slightly inductive. The vari-
able series matching capacitor (C

s

) forms the rest of the equivalent L network. This con-

figuration constitutes a high-pass filter.

© 1999 CRC Press LLC

Interstage Coupling

The IPA output circuit and the final amplifier input circuit often are coupled by a co-
axial transmission line. Impedance matching usually is accomplished at either end by
one of the configurations shown in

Figure 5.24

. The circuits shown, except for 5.24d,

require some interactive adjustment of the tuning and loading elements to provide a
satisfactory impedance match for each operating frequency and RF drive level. The
circuit in

Figure 5.24d

utilizes multiple LC sections, with each section providing a

small step in the total impedance transformation. This technique provides a broad-
band impedance match without adjustment, thereby improving transmitter stability,
ease of operation, and maintainability. A single grid resonating control is sufficient to
tune and match the 50

driver impedance to the high input impedance of the grid

over a relatively wide band of frequencies and RF power levels.

The transmission line matching problem is eliminated in some transmitter designs

by integrating an IPA stage, using one or more tubes, into the grid circuit of the final
amplifier by having the plate of the IPA and the grid of the final tube share a common
tuned circuit. This technique has the advantage of simplicity by not transforming the
impedance down to 50

and then back up to the grid impedance level.

Solid-state RF power devices possess a low load impedance at the device output ter-

minal, so an impedance transformation that goes through the 50

intermediate imped-

ance of a coaxial cable is necessary to couple these devices into the relatively high im-
pedance of the final amplifier grid.

Figure 5.23 Input matching circuits: (

a) inductive input matching, (b) capacitive input

matching.

(

a)

(

b)

© 1999 CRC Press LLC

5.6.5 Neutralization

Most cathode-driven amplifiers using grounded-grid triodes do not require neutral-
ization. It is necessary that the grid-to-ground inductance, both internal and external
to the tube, be kept low to maintain this advantage. Omission of neutralization will al-
low a small amount of interaction between the output circuit and the input circuit
through the plate-to-filament capacitance. This effect is not particularly noticeable
because of the large coupling between the input and output circuits through the elec-
tron beam of the tube. Cathode-driven tetrodes have higher gain than triodes and,
therefore, often require some form of neutralization.

Grid-driven high-gain tetrodes need accurate neutralization for best stability and

performance. Self-neutralization can be accomplished simply by placing a small
amount of inductance between the tube screen grid and ground. This inductance is usu-
ally in the form of several short adjustable-length straps. The RF current flowing from
plate to screen in the tube also flows through this screen lead inductance. This develops
a small RF voltage on the screen, of the opposite phase, which cancels the voltage fed
back through the plate-to-grid capacitance. This method of lowering the self-neutraliz-
ing frequency of the tube works only if the self-neutralizing frequency of the

Figure 5.24 Interstage RF coupling circuits: (

a) variable capacitance matching, (b) vari-

able tank matching, (

c) variable pi matching, (d) fixed broadband imped-

ance-transformation network.

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

tube/socket combination is above the desired operating frequency before the induc-
tance is added. Special attention must be given to minimizing the inductances in the
tube socket by integrating distributed bypass capacitors into the socket and cavity deck
assembly. Pentodes normally do not require neutralization because the suppressor grid
effectively isolates the plate from the grid.

5.7

Special-Application Amplifiers

The operating environment of tubes used in commercial, industrial, and military ap-
plications usually is characterized by widely varying supply voltages, heavy and vari-
able vibration, and significant changes in loading. Scientific applications are no less
demanding. Research projects span a wide range of powers and frequencies. Typical
uses for power vacuum tubes as of this writing include:

·

Super proton synchrotron delivering 2.4 MW at 200 MHz

·

Fusion reactor using 33 MW of high-frequency heating in the 25 to 60 MHz band

·

Plasma research using 12 MW in the 60 to 120 MHz range

·

Fusion research using a 500 kW ion cyclotron resonance heating generator oper-
ating at 30 to 80 MHz

It can be seen from these examples that heat dissipation is a major consideration in the
design of an amplifier or oscillator for scientific uses.

5.7.1 Distributed Amplification

Specialized research and industrial applications often require an amplifier covering
several octaves of bandwidth. At microwave frequencies, a traveling wave tube
(TWT) may be used. At MF and below, multiple tetrodes usually are combined to
achieve the required bandwidth. A distributed amplifier consists of multiple tetrodes
and lumped-constant transmission lines. The lines are terminated by load resistances
with magnitudes equal to their characteristic impedances. A cyclical wave of current
is present on the output transmission line circuit, with each tube contributing its part
in proper phase. A typical configuration for a distributed amplifier includes 8 to 16
tubes. The efficiency of such a design is low.

5.7.2 Radar

The exciter stage of a radar set comprises oscillators, frequency multipliers, and mix-
ers. The signals produced depend on whether the transmitter output device is operated
as a power amplifier or an oscillator.

Transmitters using power oscillators such as magnetrons determine the radio fre-

quency by tuning of the device itself. In a conventional (noncoherent) radar system, the
only frequency required is the local oscillator (LO). The LO differs from the magne-
tron frequency by an intermediate frequency, and this difference usually is maintained

© 1999 CRC Press LLC

with an automatic frequency control (AFC) loop.

Figure 5.25

shows a simple magne-

tron-based radar system with two methods of tuning:

·

Slaving the magnetron to follow the stable local oscillator (STALO)

·

Slaving the STALO to follow the magnetron

If the radar must use coherent detection (such as in Doppler applications), a second

oscillator, called a coherent oscillator (COHO), is required. The COHO operates at the
intermediate frequency and provides a reference output for signal processing circuits.

The synchronizer circuits in the exciter supply timing pulses to various radar sub-

systems. In a simple marine radar, this may consist of a single multivibrator that trig-
gers the transmitter. In a larger system, 20 to 30 timing pulses may be required. These
may turn the beam current on and off in various transmitter stages, start and stop RF
pulse time attenuators, start display sweeps, and perform numerous other functions.
Newer radar systems generate the required timing signals digitally. A digital synchro-
nizer is illustrated in

Figure 5.26

.

Modulator

A radar system RF amplifier usually centers on one of two microwave devices: a
crossed-field tube or linear-beam tube. Both are capable of high peak output power at
microwave frequencies. To obtain high efficiency from a pulsed radar transmitter, it is
necessary to cut off the current in the output tube between pulses. The modulator per-

Figure 5.25 Two common approaches to automatic frequency control of a magnetron
oscillator.

© 1999 CRC Press LLC

forms this function. Some RF tubes include control electrodes or grids to achieve the
same result. Three common types of modulators are used in radar equipment:

·

Line-type modulator (

Figure 5.27

). This common radar modulator is used most

often to pulse a magnetron. Between pulses, a charge is stored in a pulse-forming
network (PFN). A trigger signal fires a thyratron tube, short-circuiting the input
to the PFN, which causes a voltage pulse to appear at the primary of transformer
T1. The PFN components are chosen to produce a rectangular pulse at the magne-
tron cathode, with the proper voltage and current to excite the magnetron to oscil-
lation. An advantage of this design is its simplicity. A drawback is the inability to
electronically change the width of the transmitted pulse.

·

Active-switch modulator (

Figure 5.28

). This system permits pulse width varia-

tion, within the limitations of the energy stored in the high-voltage power supply.
A switch tube controls the generation of RF by completing the circuit path from
the output tube to the power supply, or by causing stored energy to be dumped to
the output device. The figure shows the basic design of an active-switch modula-
tor and three variations on the scheme. The circuits differ in the method of cou-
pling power supply energy to the output tube (capacitor-coupled, trans-
former-coupled, or a combination of the two methods).

Figure 5.26 Digital synchronizer system for radar applications.

© 1999 CRC Press LLC

·

Magnetic modulator (

Figure 5.29

). This design is the simplest of the three modu-

lators discussed. No thyratron or switching device is used. Operation of the mod-
ulator is based on the saturation characteristics of inductors L1, L2, and L3. A
long-duration low-amplitude pulse is applied to L1, which charges C1. As C1 ap-
proaches its fully charged state, L2 saturates and the energy in C1 is transferred in
a resonant fashion to C2. This process continues to the next stage (L3 and C3).
The transfer time is set by selection of the components to be about one-tenth that
of the previous stage. At the end of the chain, a short-duration high-amplitude
pulse is generated, exciting the RF output tube.

Because the applications for radar vary widely, so do antenna designs. Sizes range

from less than 1 ft to hundreds of feet in diameter. An antenna intended for radar appli-
cations must direct radiated power from the transmitter to the azimuth and elevation co-
ordinates of the target. It also must serve as a receive antenna for the echo.

Figure 5.27 A line-type modulator for radar.

© 1999 CRC Press LLC

Figure 5.28 Active-switch modulator circuits: (

a) direct-coupled system, (b) capaci-

tor-coupled, (

c) transformer-coupled, (d) capacitor- and transformer-coupled.

Figure 5.29 A magnetic modulator circuit.

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

5.8

References

1. Laboratory Staff, The Care and Feeding of Power Grid Tubes, Varian Eimac, San

Carlos, CA, 1984.

2. Woodard, G. W., "AM Transmitters," in NAB Engineering Handbook, 8th Ed., E. B.

Crutchfield (ed.), National Association of Broadcasters, Washington, DC, pg. 356,
1992.)

5.9

Bibliography

Chireix, Henry, "High Power Outphasing Modulation," Proceedings of the IRE, Vol.

23, no. 11, pg. 1370, November 1935.

Crutchfield, E. B., (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, 1992.

Doherty, W. H., "A New High Efficiency Power Amplifier for Modulated Waves,"

Proceedings of the IRE, Vol. 24, no. 9, pg. 1163, September 1936.

Fink, D., and D. Christiansen, Electronic Engineers' Handbook, 3rd Ed.,

McGraw-Hill, New York, 1989.

Gray, T. S., Applied Electronics, Massachusetts Institute of Technology, 1954.
Heising, R. A., "Modulation in Radio Telephony," Proceedings of the IRE, Vol. 9, no.

3, pg. 305, June 1921.

Heising, R. A., "Transmission System," U.S. Patent no. 1,655,543, January 1928.
High Power Transmitting Tubes for Broadcasting and Research, Philips Technical

Publication, Eindhoven, the Netherlands, 1988.

Honey, J. F., "Performance of AM and SSB Communications," Tele-Tech., September

1953.

Jordan, Edward C., Reference Data for Engineers: Radio, Electronics, Computers, and

Communications, 7th Ed., Howard W. Sams, Indianapolis, IN, 1985.

Martin, T. L., Jr., Electronic Circuits, Prentice Hall, Englewood Cliffs, NJ, 1955.
Mina and Parry, "Broadcasting with Megawatts of Power: The Modern Era of Efficient

Powerful Transmitters in the Middle East," IEEE Transactions on Broadcasting,
Vol. 35, no. 2, IEEE, Washington, D.C., June 1989.

Pappenfus, E. W., W. B. Bruene, and E. O. Schoenike, Single Sideband Principles and

Circuits, McGraw-Hill, New York, 1964.

Ridgwell, J. F., "A New Range of Beam-Modulated High-Power UHF Television

Transmitters," Communications & Broadcasting, no. 28, Marconi Communica-
tions, Clemsford Essex, England, 1988.

Skolnik, M. I. (ed.), Radar Handbook, McGraw-Hill, New York, 1980.
Terman, F. E., and J. R. Woodyard, "A High Efficiency Grid-Modulated Amplifier,

Proceedings of the IRE, Vol. 26, no. 8, pg. 929, August 1938.

Terman, F. E., Radio Engineering, 3rd Ed., McGraw-Hill, New York, 1947.
Weldon, J. O., "Amplifiers," U. S. Patent no. 2,836,665, May 1958.
Whitaker, Jerry C., AC Power Systems, 2nd Ed., CRC Press, Boca Raton, FL, 1998.
Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1991.

Woodard, George W., "AM Transmitters," in NAB Engineering Handbook, 9th Ed.,

Jerry C. Whitaker (ed.), National Association of Broadcasters, Washington, D.C.,
pp. 353-381, 1998.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

6 Microwave Power Tubes"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

6

Microwave Power Tubes

6.1

Introduction

Microwave power tubes span a wide range of applications, operating at frequencies
from 300 MHz to 300 GHz with output powers from a few hundred watts to more than
10 MW. Applications range from the familiar to the exotic. The following devices are
included under the general description of microwave power tubes:

·

Klystron, including the reflex and multicavity klystron

·

Multistage depressed collector (MSDC) klystron

·

Inductive output tube (IOT)

·

Traveling wave tube (TWT)

·

Crossed-field tube

·

Coaxial magnetron

·

Gyrotron

·

Planar triode

·

High-frequency tetrode

·

Diacrode

This wide variety of microwave devices has been developed to meet a broad range of
applications. Some common uses include:

·

UHF-TV transmission

·

Shipboard and ground-based radar

·

Weapons guidance systems

·

Electronic countermeasure (ECM) systems

·

Satellite communications

·

Tropospheric scatter communications

© 1999 CRC Press LLC

·

Fusion research

As new applications are identified, improved devices are designed to meet the

needs. Microwave power tube manufacturers continue to push the limits of frequency,
operating power, and efficiency. Microwave technology, therefore, is an evolving sci-
ence.

Figure 6.1

charts device type as a function of operating frequency and power out-

put.

Two principal classes of microwave vacuum devices are in common use today:

·

Linear-beam tubes

·

Crossed-field tubes

Each class serves a specific range of applications. In addition to these primary
classes, some power grid tubes also are used at microwave frequencies.

6.1.1 Linear-Beam Tubes

In a linear-beam tube, as the name implies, the electron beam and the circuit elements
with which it interacts are arranged linearly. The major classifications of linear-beam
tubes are shown in

Figure 6.2

. In such a device, a voltage applied to an anode acceler-

ates electrons drawn from a cathode, creating a beam of kinetic energy. Power supply
potential energy is converted to kinetic energy in the electron beam as it travels to-
ward the microwave circuit. A portion of this kinetic energy is transferred to micro-
wave energy as RF waves slow down the electrons. The remaining beam energy is ei-
ther dissipated as heat or returned to the power supply at the collector. Because elec-
trons will repel one another, there usually is an applied magnetic focusing field to
maintain the beam during the interaction process. The magnetic field is supplied ei-
ther by a solenoid or permanent magnets.

Figure 6.3

shows a simplified schematic of

a linear-beam tube.

6.1.2 Crossed-Field Tubes

The magnetron is the pioneering device of the family of crossed-field tubes. The fam-
ily tree of this class of devices is shown in

Figure 6.4

. Although the physical appear-

ance differs from that of linear-beam tubes, which are usually circular in format, the
major difference is in the interaction physics that requires a magnetic field at right an-
gles to the applied electric field. Whereas the linear-beam tube sometimes requires a
magnetic field to maintain the beam, the crossed-field tube always requires a mag-
netic focusing field.

Figure 6.5

shows a cross section of the magnetron, including the magnetic field ap-

plied perpendicular to the cathode-anode plane. The device is basically a diode with the
anode composed of a plurality of resonant cavities. The interaction between the elec-
trons emitted from the cathode and the crossed electric and magnetic fields produces a
series of space-charge spokes that travel around the anode-cathode space in a manner
that transfers energy to the RF signal supported by the multicavity circuit. The mecha-
nism is highly efficient.

© 1999 CRC Press LLC

Figure 6.1 Microwave power tube type as a function of frequency and output power.

©

1
999
CRC Press LLC

Figure 6.2 Types of linear-beam microwave tubes.

Figure 6.3 Schematic diagram of a linear-beam tube.

© 1999 CRC Press LLC

Crossed-Field Amplifiers

Figure 6.6

shows the family tree of the crossed-field amplifier (CFA). The configura-

tion of a typical present-day distributed emission amplifier is similar to that of the
magnetron except that the device has an input for the introduction of RF energy into
the circuit. Current is obtained primarily by secondary emission from the negative

Figure 6.4 Types of crossed-field microwave tubes.

Figure 6.5 Magnetron electron path looking down into the cavity with the magnetic field
applied.

© 1999 CRC Press LLC

electrode that serves as a cathode throughout all or most of the interaction space. The
earliest versions of this tube type were called amplitrons.

The CFA is deployed in radar systems operating from UHF to the Ku-band, and at

power levels up to several megawatts. In general, bandwidth ranges from a few percent
to as much as 25 percent of the center frequency.

6.2

Grid Vacuum Tubes

The physical construction of a vacuum tube causes the output power and available
gain to decrease with increasing frequency. The principal limitations faced by
grid-based devices include the following:

·

Physical size. Ideally, the RF voltages between electrodes should be uniform, but
this condition cannot be realized unless the major electrode dimensions are sig-
nificantly less than 1/4 wavelength at the operating frequency. This restriction
presents no problems at VHF, but as the operating frequency increases into the
microwave range, severe restrictions are placed on the physical size of individual
tube elements.

·

Electron transit time. Interelectrode spacing, principally between the grid and the
cathode, must be scaled inversely with frequency to avoid problems associated
with electron transit time. Possible adverse conditions include: 1) excessive load-
ing of the drive source, 2) reduction in power gain, 3) back-heating of the cathode
as a result of electron bombardment, and 4) reduced conversion efficiency.

Figure 6.6 Family tree of the distributed emission crossed-field amplifier (CFA).

© 1999 CRC Press LLC

·

Voltage standoff. High-power tubes operate at high voltages. This presents sig-
nificant problems for microwave vacuum tubes. For example, at 1 GHz the
grid-cathode spacing must not exceed a few mils. This places restrictions on the
operating voltages that may be applied to the individual elements.

·

Circulating currents. Substantial RF currents may develop as a result of the inher-
ent interelectrode capacitances and stray inductances/capacitances of the device.
Significant heating of the grid, connecting leads, and vacuum seals may result.

·

Heat dissipation. Because the elements of a microwave grid tube must be kept
small, power dissipation is limited.

Still, a number of grid-based vacuum tubes find applications at high frequencies. For
example, planar triodes are available that operate at several gigahertz, with output
powers of 1 to 2 kW in pulsed service. Efficiency (again for pulsed applications)
ranges from 30 to 60 percent, depending on the frequency.

6.2.1 Planar Triode

A cross-sectional diagram of a planar triode is shown in

Figure 6.7

. The envelope is

made of ceramic, with metal members penetrating the ceramic to provide for connec-
tion points. The metal members are shaped either as disks or as disks with cylindrical
projections.

The cathode is typically oxide-coated and indirectly heated. The key design objec-

tive for a cathode is high emission density and long tube life. Low-temperature emitters
are preferred because high cathode temperatures typically result in more evaporation
and shorter life.

The grid of the planar triode is perhaps the greatest design challenge for tube manu-

facturers. Close spacing of small-sized elements is needed, at tight tolerances. Good
thermal stability also is required, because the grid is subjected to heating from currents
in the element itself, plus heating from the cathode and bombardment of electrons from
the cathode.

The anode, usually made of copper, conducts the heat of electron bombardment to

an external heat sink. Most planar triodes are air-cooled.

Planar triodes designed for operation at 1 GHz and above are used in a variety of cir-

cuits. The grounded-grid configuration is most common. The plate resonant circuit is
cavity-based, using waveguide, coaxial line, or stripline. Electrically, the operation of
the planar triode is much more complicated at microwave frequencies than at low fre-
quencies.

Figure 6.8a

compares the elements at work for a grounded-grid amplifier op-

erating at low frequencies and

Figure 6.8b

compares the situation at microwave fre-

quencies. The equivalent circuit is made more complex by:

·

Stray inductance and capacitance of the tube elements

·

Effects of the tube contact rings and socket elements

·

Distributed reactance of cavity resonators and the device itself

© 1999 CRC Press LLC

·

Electron transit-time effects, which result in resistive loading and phase shifts

Reasonable gains of 5 to 10 dB may be achieved with a planar triode. Increased gain

is available by cascading stages. Interstage coupling may consist of waveguide or coax-
ial-line elements. Tuning is accomplished by varying the cavity inductance or capaci-
tance. Additional bandwidth is possible by stagger tuning of cascaded stages.

6.2.2 High-Power UHF Tetrode

New advancements in vacuum tube technology have permitted the construction of
high-power UHF transmitters based on tetrodes. Such devices are attractive because
they inherently operate in a relatively efficient class AB mode. UHF tetrodes operat-
ing at high power levels provide essentially the same specifications, gain, and effi-
ciency as tubes operating at lower powers. The anode power supply is much lower in
voltage than the collector potential of a klystron- or IOT-based system (8 kV is com-
mon). Also, the tetrode does not require a focusing magnet system.

Figure 6.7 Cross section of a 7289 planar triode.

© 1999 CRC Press LLC

Efficient removal of heat is the key to making a tetrode practical at high power lev-

els. Such devices typically use water or vapor-phase cooling. Air cooling at such levels
is impractical because of the fin size that would be required. Also, the blower for the
tube would have to be quite large, reducing the overall transmitter ac-to-RF efficiency.

Another drawback inherent in tetrode operation is that the output circuit of the de-

vice appears electrically in series with the input circuit and the load [1]. The parasitic
reactance of the tube elements, therefore, is a part of the input and output tuned circuits.
It follows, then, that any change in the operating parameters of the tube as it ages can af-
fect tuning. More importantly, the series nature of the tetrode places stringent limita-
tions on internal element spacings and the physical size of those elements in order to
minimize the electron transit time through the tube vacuum space. It is also fair to point
out, however, the tetrode's input-to-output circuit characteristic has at least one advan-
tage: power delivered to the input passes through the tube and contributes to the total
power output of the transmitter. Because tetrodes typically exhibit low gain compared

Figure 6.8 Grounded-grid equivalent circuits: (

a) low-frequency operation, (b) micro-

wave-frequency operation. The cathode-heating and grid-bias circuits are not shown.

(

a)

(

b)

© 1999 CRC Press LLC

to a klystron-based device, significant power can be required at the input circuit. The
pass-through effect, therefore, contributes to the overall operating efficiency of the
transmitter.

The expected lifetime of a tetrode in UHF service is usually shorter than a klystron

of the same power level. Typical lifetimes of 8,000 to 15,000 hours have been reported.
Intensive work, however, has led to products that offer higher output powers and ex-
tended operating lifetime, while retaining the benefits inherent in tetrode devices.

6.2.3 Diacrode

The Diacrode (Thomson) is a promising adaptation of the high-power UHF tetrode.

The operating principle of the Diacrode is basically the same as that of the tetrode. The
anode current is modulated by an RF drive voltage applied between the cathode and the
power grid. The main difference is in the position of the active zones of the tube in the
resonant coaxial circuits, resulting in improved reactive current distribution in the elec-
trodes of the device.

Figure 6.9

compares the conventional tetrode with the Diacrode. The Diacrode in-

cludes an electrical extension of the output circuit structure to an external cavity [2].
The small dc-blocked cavity rests on top of the tube, as illustrated in

Figure 6.10

.

The cavity is a quarter-wave transmission line, as measured from the top of the cav-

ity to the vertical center of the tube. The cavity is short-circuited at the top, reflecting an
open circuit (current minimum) at the vertical center of the tube and a current maxi-
mum at the base of the tube, like the conventional tetrode, and a second current maxi-
mum above the tube at the cavity short-circuit.

Figure 6.9 Cutaway view of the tetrode (

left) and the Diacrode (right). Note that the RF

current peaks above and below the Diacrode center while on the tetrode there is only
one peak at the bottom. (

After [2].)

© 1999 CRC Press LLC

With two current maximums, the RF power capability of the Diacrode is double that

of the equivalent tetrode, while the element voltages remain the same. All other proper-
ties and aspects of the Diacrode are basically identical to the TH563 high-power UHF
tetrode (Thomson), upon which the Diacrode is patterned.

Some of the benefits of such a device, in addition to the robust power output avail-

able, is the low high-voltage requirements (low relative to a klystron/IOT-based system,
that is), small size, and simple replacement procedures. On the downside, there is little
installed service lifetime data at this writing because the Diacrode is relatively new to
the market.

6.3

Klystron

The klystron is a linear-beam device that overcomes the transit-time limitations of a
grid-controlled tube by accelerating an electron stream to a high velocity before it is
modulated. Modulation is accomplished by varying the velocity of the beam, which
causes the drifting of electrons into bunches to produce RF space current. One or
more cavities reinforce this action at the operating frequency. The output cavity acts
as a transformer to couple the high-impedance beam to a low-impedance transmission
line. The frequency response of a klystron is limited by the impedance-bandwidth
product of the cavities, but may be extended through stagger tuning or the use of mul-
tiple-resonance filter-type cavities.

The klystron is one of the primary means of generating high power at UHF and

above. Output powers for multicavity devices range from a few thousand watts to 10

Figure 6.10 The elements of the Diacrode, including the upper cavity. Double current,
and consequently, double power, is achieved with the device because of the current
peaks at the top and bottom of the tube, as shown. (

After [2].)

© 1999 CRC Press LLC

MW or more. The klystron provides high gain and requires little external support cir-
cuitry. Mechanically, the klystron is relatively simple. It offers long life and requires
minimal routine maintenance.

6.3.1 Reflex Klystron

The reflex klystron uses a single-cavity resonator to modulate the RF beam and ex-
tract energy from it. The construction of a reflex klystron is shown in

Figure 6.11

. In

its basic form, the tube consists of the following elements:

·

A cathode

·

Focusing electrode at cathode potential

·

Coaxial line or reentrant-type cavity resonator, which also serves as an anode

·

Repeller or reflector electrode, which is operated at a moderately negative poten-
tial with respect to the cathode

The cathode is so shaped that, in relation to the focusing electrode and anode, an elec-
tron beam is formed that passes through a gap in the resonator, as shown in the figure,
and travels toward the repeller. Because the repeller has a negative potential with re-
spect to the cathode, it turns the electrons back toward the anode, where they pass
through the anode gap a second time. By varying the applied voltage on the reflector

Figure 6.11 Schematic representation of a reflex klystron.

© 1999 CRC Press LLC

electrode, phasing of the beam can be varied to produce the desired oscillating mode
and to control the frequency of oscillation.

The variation of position with time for electrons in the anode-repeller space is illus-

trated in

Figure 6.12

. Path a corresponds to an electron that emerges from the anode

with a velocity corresponding to the anode voltage. This electron follows a parabolic
path, as shown, determined by the electric field in the anode-repeller space.

Operation of the reflex klystron can best be understood by examining the movement

of electrons inside the device. Assume that oscillations exist in the resonator so that an
alternating voltage develops across the gap. Assume further that the electron corre-
sponding to path a passed through the gap at the instant that this alternating voltage
across the gap was zero and becoming negative. An electron passing through the gap
just before electron a will encounter an accelerating voltage across the gap and, there-
fore, will emerge from the anode with greater velocity than the first or reference elec-
tron. This second electron, accordingly, penetrates farther toward the repeller against
the retarding field and, as a result, takes longer to return to the anode. Consequently,
this electron follows path b, as shown in

Figure 6.12

, and tends to arrive at the anode on

its return path at the same time as the reference electron because its earlier start is more
or less compensated for by increased transit time. In a similar manner, an electron pass-
ing through the anode gap slightly later than the reference electron will encounter a
negative or retarding field across the gap, and so will emerge from the anode with less
velocity than the electron that follows path a. This third electron will then follow trajec-
tory c and return to the anode more quickly than electron a. Electron c, therefore, tends
to return to the anode at about the same time as electron a, because the later start of elec-
tron c is compensated for by the reduced transit time.

Figure 6.12 Position-time curves of electrons in the anode-repeller space, showing the
tendency of the electrons to bunch around the electron passing through the anode at the
time when the alternating gap voltage is zero and becoming negative.

© 1999 CRC Press LLC

This variation with time of the velocity of electrons emerging from the anode is

termed velocity modulation. The effect of this phenomenon can be seen in

Figure 6.12

to cause a bunching of electrons about the electron that passed through the gap when the
resonator voltage was zero and becoming negative. This bunching causes the electrons
that are returned toward the anode by the repeller to pass through the anode gap in
bursts or pulses, one each cycle. When these pulses pass through the gap at such a time
that the electrons in the pulse are slowed as a result of the alternating voltage existing
across the gap at the instant of their return passage, energy will be delivered to the oscil-
lations in the resonator, thereby assisting in maintaining the oscillations. This condition
corresponds to a transit time N from the resonator toward the repeller and back to the
resonator of approximately:

N

n

= +

3

4

(6.1)

Where:
n = an integer (including zero)

The transit time in the anode-repeller space in any particular case depends upon the
following:

·

The anode voltage

·

Repeller voltage

·

Geometry of the anode-repeller space

The extent of the bunching action that takes place when the transit time of the refer-
ence electron has the correct value for sustaining oscillations is determined by the fol-
lowing:

·

The amplitude of the alternating voltage across the resonator gap in relation to the
anode and repeller voltage

·

The geometry of the repeller space

The reflex klystron typically includes a grid to concentrate the electric field so that it

may efficiently couple to the electron beam. Such a device is illustrated in

Figure 6.13

.

The reflex klystron may be used as a local oscillator, low-power FM transmitter, or

test signal source. Reflex tubes are used primarily from 4 to 40 GHz. Power outputs of 1
W or less are common.

The reflex tube is the only klystron in which beam feedback is used to produce out-

put energy. In klystrons with more than one cavity, the electron beam passes through
each cavity in succession.

6.3.2 The Two-Cavity Klystron

The two-cavity klystron operates on the same bunching principle as the reflex kly-
stron, but incorporates two cavities connected by a drift tube.

Figure 6.14

shows a

cross section of a classic device. The heater/cathode element (shown as A in the fig-

© 1999 CRC Press LLC

ure) produces an electron beam in conjunction with a focusing electrode. The route
taken by the electrons is as follows:

·

The beam passes through grid elements D in the side of a reentrant cavity resona-
tor (the buncher).

·

The beam then passes through the drift tube, which is at the same electrical poten-
tial as the buncher.

·

Finally, the beam enters a second resonator termed the collector, which is pro-
vided with a grid E.

The cathode and its associated focusing electrode are maintained at a high negative
potential with respect to the remaining part of the structure, all of which is at the same
dc potential. The entire arrangement illustrated in the figure is enclosed in a vacuum.

The operational principles of the two-cavity klystron are similar in nature to those of

the reflex klystron. Assume, first, that oscillations exist in the buncher so that an alter-
nating voltage is present across the gap D. When this voltage is zero but just becoming
positive, an electron passing through the buncher travels through the grids D, down the
drift tube, and into the collector resonator with unchanged velocity. However, an elec-
tron that passes through the buncher slightly later receives acceleration while passing
through, because of the positive alternating field that it encounters between grids D,
and enters the drift tube with increased velocity. This later electron, therefore, tends to
overtake the earlier electron. Similarly, an electron that arrives at the buncher slightly

Figure 6.13 Schematic cross section of a reflex klystron oscillator.

© 1999 CRC Press LLC

earlier than the first (reference electron) encounters a field between grids D that op-
poses its motion. Hence, this early electron enters the drift tube with reduced velocity
and tends to drop back and be overtaken by the reference electron.

As a result of these actions, the electrons bunch together as they travel down the drift

tube. This effect is more pronounced at certain distances from the buncher. If the col-
lector is located at a distance where the bunching is pronounced, the electrons enter the
element in pulses, one pulse per cycle.

With proper adjustment, the amount of power required to produce the bunching ef-

fect is relatively small compared with the amount of energy delivered by the electron
beam to the collector. As a result, the klystron operates as an amplifying device.

The Two-Cavity Klystron Oscillator

The two-cavity klystron oscillator is designed for applications requiring moderate
power (up to 100 W), stable frequency output, and low sideband noise. The device has
a coupling iris on the wall between the two cavities. The tube can be frequency-modu-
lated by varying the cathode voltage about the center of the oscillating mode. Al-
though it is more efficient and powerful than the reflex klystron, the two-cavity kly-

Figure 6.14 Cross section of a classic two-cavity klystron oscillator.

© 1999 CRC Press LLC

stron requires more modulator power. The two-cavity klystron typically is used in
Doppler radar systems.

The Two-Cavity Klystron Amplifier

Similar in design to the two-cavity oscillator, the two-cavity klystron amplifier pro-
vides limited power output (10 W or less) and moderate gain (about 10 dB). A driving
signal is coupled into the input cavity, which produces velocity modulation of the
beam. After the drift space, the density-modulated beam induces current in the output
resonator. Electrostatic focusing of the beam is common.

The two-cavity klystron finds only limited applications because of its restrictions on

output power and gain. For many applications, solid-state amplifiers are a better choice.

6.3.3 The Multicavity Klystron

The multicavity klystron is an important device for amplifying signals to high power
levels at microwave frequencies [3]. Each cavity tuned to the operating frequency
adds about 20 dB gain to the 10 dB gain offered by the basic two-cavity klystron am-
plifier. Overall gains of 60 dB are practical. Cavities may be tuned to either side of
resonance to broaden the operating bandwidth of the device. Klystrons with up to
eight cavities have been produced. Operating power for continuous wave klystrons
ranges up to 1 MW per device, and as much as 50 MW per device for pulsed applica-
tions.

The primary physical advantage of the klystron over a grid-based power tube is that

the cathode-to-collector structure is virtually independent of transit-time effects.
Therefore, the cathode can be made large and the electron beam density kept low.

The operating frequency of a klystron may be fixed--determined by the mechanical

characteristics of the tube and its cavities--or tunable. Cavities are tuned mechanically
using one of several methods, depending on the operating power and frequency. Tuning
is accomplished by changing the physical dimensions of the cavities using one or more
of the following techniques:

·

Cavity wall deformation, in which one wall of the cavity consists of a thin dia-
phragm that is moved in and out by a tuning mechanism. About 3 percent fre-
quency shift may be accomplished using this method, which varies the induc-
tance of the cavity.

·

Movable cavity wall, in which one wall of the cavity is moved in or out by a tuning
mechanism. About 10 percent frequency shift is possible with this approach,
which varies the inductance of the cavity.

·

Paddle element, in which an element inside the cavity moves perpendicularly to
the beam and adds capacitance across the interaction gap. A tuning range of about
25 percent is provided by this approach.

·

Combined inductive-capacitive tuning, which uses a combination of the previous
methods. Tuning variations of 35 percent are possible.

© 1999 CRC Press LLC

Each of these tuning methods may be used whether the cavity is inside or outside the
vacuum envelope of the tube. Generally speaking, however, tubes that use external
cavities provide more adjustment range, usually on the order of 35 percent. Band-
width may be increased by stagger tuning of the cavities, at the expense of gain.

High conversion efficiency requires the formation of electron bunches, which oc-

cupy a small region in velocity space, and the formation of interbunch regions with low
electron density. The latter is particularly important because these electrons are phased
to be accelerated into the collector at the expense of the RF field. Studies show that the
energy loss as a result of an electron accelerated into the collector may exceed the en-
ergy delivered to the field by an equal but properly phased electron. Therein lies a key in
improving the efficiency of the klystron: Recover a portion of this wasted energy.

Klystrons are cooled by air or liquid for powers up to 5 kW. Tubes operating in ex-

cess of 5 kW are usually water- or vapor-cooled.

Operating Principles

A high-velocity electron beam emitted from the cathode passes through the anode and
into the RF interaction region, as shown in

Figure 6.15

[3]. An external magnetic field

is employed to prevent the beam from spreading as it passes through the tube. At the
other end of the device, the electron beam impinges on the collector electrode, which
dissipates the beam energy and returns the electron current to the beam power supply.

Figure 6.15 Principal elements of a klystron.

© 1999 CRC Press LLC

The RF interaction region, where amplification occurs, contains resonant cavities

and field-free drift spaces. The first resonant cavity encountered by an electron in the
beam (the input cavity) is excited by the microwave signal to be amplified, and an alter-
nating voltage of the signal frequency is developed across the gap. This action can be
best explained by drawing an analogy between a resonant cavity and a conventional
parallel resonant LC circuit (see

Figure 6.16

). The cavity gap corresponds to the capaci-

tor, and the volume of the cavity to the inductor. If the cavity is of the correct physical
dimensions, it will resonate at the desired microwave frequency. At resonance, opposite
sides of the gap become alternately positive and negative at a frequency equal to the mi-
crowave input signal frequency.

An electron passing through the gap when the voltage across the gap is zero contin-

ues with unchanged velocity along the drift tube toward the next cavity gap; this elec-
tron is the reference electron. An electron passing through the same gap slightly later is
accelerated by the positive field at the gap. This electron speeds up and tends to over-
take the reference electron ahead of it in the drift tube. However, an electron that passes
through the gap slightly ahead of the reference electron encounters a negative field and
is slowed down. This electron tends to fall back toward the following reference elec-
tron. As a result of passing the alternating field of the input-cavity gap, the electrons
gradually bunch together as they travel down the drift tube, as illustrated in

Figure 6.17

.

Because electrons approach the input-cavity gap with equal velocities and emerge with
different velocities, which are a function of the microwave signal, the electron beam is
said to be velocity-modulated. As the electrons travel down the drift tube, bunching de-
velops, and the density of electrons passing a given point varies cyclically with time.
This bunching is identical in nature to the action in a two-cavity klystron, discussed in
Section 6.3.2.

The modulation component of the beam current induces current in each of the fol-

lowing cavities. Because each cavity is tuned near resonance, the resulting increase in
field at each gap produces successively better-defined electron bunches and, conse-
quently, amplification of the input signal. The RF energy produced in this interaction
with the beam is extracted from the beam and fed into a coaxial or waveguide transmis-
sion line by means of a coupling loop in the output cavity. The dc beam input power not
converted to RF energy is dissipated in the collector.

Figure 6.16 Klystron cavity: (

a) physical arrangement, (b) equivalent circuit.

(

a)

(

b)

© 1999 CRC Press LLC

Emission

The electron gun section of a klystron, shown in

Figure 6.18

, consists of the following

elements [3]:

·

A heater

·

An emitter

·

A beam-forming focusing electrode

·

A modulating anode

When the emitter temperature is raised to the proper value by the heater, electrons are
released from the emitter surface. The electrons are accelerated toward the modulat-
ing anode, which is at a positive potential with respect to the emitter. As the electrons
travel between the emitter and the modulating anode, they are formed into a beam by
the lens action of the focusing electrode and modulating anode.

Figure 6.19

shows

how this lens is formed.

All cathodes have a specific optimum range of operating temperature. The tempera-

ture of the cathode must be sufficiently high to prevent variations in heater power from
affecting the electron emission current (beam current) in the klystron. However, the
temperature of the emitting surface must not be higher than necessary because exces-
sive temperature can shorten emission life.

Figure 6.20

shows beam current (emission

current) as a function of the emitter temperature, which varies directly with heater
power. When the heater voltage (E

f1

) is too low, the emitter will not be hot enough to

Figure 6.17 Bunching effect of a multicavity klystron.

© 1999 CRC Press LLC

produce the desired beam current. In addition, even small variations in heater voltage
will change the beam current significantly. With the proper heater voltage (E

f2

), con-

stant beam current will be maintained even with minor variations in heater voltage. The
same is true for a higher heater voltage value (E

f3

), but in this case the emitter tempera-

ture is greater than that needed for the desired beam current. Reduced tube life will re-

Figure 6.18 Diode section of a klystron electron gun.

Figure 6.19 Beam forming in the diode section of a klystron electron gun.

© 1999 CRC Press LLC

sult. The correct value of heater voltage and/or heater current is included in the data
shipped with each klystron.

Modulating Anode

Because the modulating anode is electrically isolated from both the cathode and kly-
stron body (the RF structure, between polepieces) the voltage applied to it provides a
convenient means for controlling beam current independently of the beam voltage ap-
plied between the cathode and body, as shown in

Figure 6.21

. When the cathode is op-

erated in the space-charge-limited region, E

f2

and E

f3

of

Figure 6.20

, the emission cur-

rent will be a specific function of the applied voltage:

I

k E

b

=

3

2

(6.2)

Where:
I

b

= beam current in amps

E = beam potential in volts

The constant k is a function of the geometry of the cathode-anode structure, and is
termed perveance. Because the modulating anode is physically positioned between
the RF structure (body) and the cathode, even if the full beam voltage is maintained
between cathode and body, the actual beam current into the tube may be reduced at
will by biasing the modulating anode to any voltage between cathode and body.

Fig-

ure 6.22

shows the relationship between beam current and voltage described in the

previous equation.

Figure 6.20 Klystron beam-current variation as a function of emitter temperature.

© 1999 CRC Press LLC

Magnetic Field

Beam focusing is used in multielement klystrons to keep the electron beam uniformly
small. Focusing may be accomplished by one or more electrostatic lenses or external
magnetic fields placed parallel to the beam. Magnetic focusing is the most common
method. Permanent magnets are used at operating powers of about 5 kW or less. Elec-
tromagnets are used at higher powers.

Electromagnetic coils typically are placed around the klystron to develop the mag-

netic field along the axis of the RF circuit. This field controls the size of the electron
beam and keeps it aligned with the drift tubes.

Figure 6.23

illustrates the beam-forming

portion at the cathode end of the klystron and RF section, where the magnetic field is
developed between two cylindrical disks (polepieces). The electron beam in this illus-
tration is shown traveling two paths. One path shows the beam spreading out to points
A; the other path shows the beam confined by the magnetic field to a constant size
throughout the distance between polepieces. The beam spreads toward points A when
the magnetic field is inadequate.

Figure 6.24

shows the magnetic field pattern of a typical solenoid used for klystrons.

When direct current passes through the magnetic coils, a magnetic field is generated
along the axis of the tube. The strength of this field can be controlled by changing the
current flow through the magnetic coils. The shape of the field is determined by
polepiece geometry and winding distribution inside the solenoid.

Figure 6.25

illus-

trates the field pattern and the shape of the beam for a properly adjusted field.

Figure 6.26

demonstrates how the beam of a klystron is distorted when an external

magnetic material is placed near the RF circuit of the tube. Electrons in the beam will
follow the bent magnetic field lines and may strike the walls of the drift tube. Klystron
damage can result. Magnetic materials such as screwdrivers, wrenches, bolts, and nuts
must not be left near the magnetic circuit or near the cathode or collector. Magnetic
tools must not be used to tune a klystron.

Figure 6.21 Modulating-anode electrode in a multicavity klystron.

© 1999 CRC Press LLC

Figure 6.23 The effect of a magnetic field on the electron beam of a klystron.

Figure 6.22 Beam-current variation as a function of modulating-anode voltage.

© 1999 CRC Press LLC

RF Structure

The RF structure of a klystron amplifier consists of tunable resonant circuits (cavi-
ties) positioned along the axis of the electron beam. The electron beam traveling
through the cavities provides the necessary coupling between each of the RF circuits.

The cavities of a klystron are high-frequency parallel resonant circuits constructed

so that they provide an RF voltage across the capacitive component (gap), which inter-
acts with the dc beam.

Figure 6.27a

illustrates the polarity near the drift-tube tips within

a cavity excited by an alternating voltage of microwave signal frequency.

Figure 6.27b

Figure 6.24 Field pattern of a klystron electromagnet.

Figure 6.25 Field pattern and beam shape in a properly adjusted magnetic field.

© 1999 CRC Press LLC

is the equivalent circuit of a simple cavity. To achieve circuit resonance, the inductive
and capacitive reactances of each of the components must be equal. The reactance of
each of the components shown in the figure can be measured separately. However, the
reactances of the components within a klystron cavity are difficult to determine, be-
cause they cannot be measured individually. Therefore, the regions of voltage maxima
or minima are used to define each component of a klystron cavity in the following way:

·

The capacitance of a cavity is developed across the gap at the drift tubes where the
voltage is at maximum.

·

The inductance of a cavity is located in the outer volume of the cavity where the
voltage is at minimum.

Figure 6.26 Distortion of the field pattern and beam shape due to magnetic material in
the magnetic field.

Figure 6.27 Klystron cavity: (

a) physical element, (b) equivalent electric circuit.

(

a)

(

b)

© 1999 CRC Press LLC

By defining each component of a cavity in these terms, it is easy to visualize changes
in the volume as changes in inductance, and changes affecting the gap as changes in
capacitance.

The resonant frequency of each cavity can be adjusted to the operating frequency of

the RF generator. This can be accomplished in one of two ways:

·

Change the inductance by changing the volume of the cavity.

·

Change the capacitance of the drift-tube gaps.

Figure 6.28

illustrates how the capacitance of the cavity gap can be modified by at-

taching a post to a thin-wall diaphragm with a paddle close to the drift-tube gap. In (a),
the mechanical configuration of a cavity with this type of tuning is illustrated; (b)
shows the equivalent circuit capacitance formed between the paddle and the drift tubes
at the gap. Moving the paddle away from the drift tubes decreases the gap capacitance
and increases the resonant frequency of the cavity.

Figure 6.29

shows a schematic diagram of the equivalent circuits of a four-cavity

klystron. Circuit 1 is the input, and circuit 4 is the output.

Figure 6.30a

illustrates mag-

netic-loop coupling, where the RF energy is fed through a coaxial line with its center
conductor inserted into the klystron cavity. The end of the center conductor is formed
into a loop. This forms a simple one-turn transformer that couples RF energy into or out
of the cavity through a coaxial transmission line.

Figure 6.30b

shows the equivalent cir-

cuit. The transformer formed by the loop and cavity is an impedance-matching device
between the transmission line and the cavity.

Klystron cavities may be externally loaded to improve their instantaneous band-

width characteristics. These loads lower the Q of the cavities slightly and thereby in-
crease the bandwidth of the klystron.

Figure 6.28 Capacitance-tuned klystron cavity: (

a) physical element, (b) equivalent cir-

cuit.

(

a)

(

b)

© 1999 CRC Press LLC

Phased Electron Operation

By properly phasing the second-harmonic fields of a klystron, a favorable electron
density distribution pattern can be established at the output gap. The result is the gen-
eration of additional RF energy.

A phase-space diagram for a high-power klystron is shown in

Figure 6.31

. The

curves represent a plot of the relative phase of the reference electrons as a function of
axial distance along the tube. Electrons having negative slope have been decelerated.
Electrons having positive slope have been accelerated with respect to a nonaccelerated
electron parallel to the axis. The diagram shows how the electrons are nicely grouped at
the output cavity gap while the interbunch regions are relatively free of electrons.

This interaction can be viewed another way, as shown in

Figure 6.32

, which plots the

normalized RF beam currents as a function of distance along the tube. The curves show
that the fundamental component of the plasma wave has a negative slope at the third
gap. This normally would be a poor condition, but because of the drift of the interbunch

Figure 6.29 Schematic equivalent circuit of a four-cavity klystron.

Figure 6.30 Klystron loop coupling: (

a) mechanical arrangement, (b) equivalent electric

circuit.

(

a)

(

b)

© 1999 CRC Press LLC

electrons, the fundamental current peaks at nearly 1.8 times the dc beam current. The
theoretical limit for perfect bunching in a delta function is 2. The second harmonic of
the plasma wave also peaks at the output gap, which adds to the conversion efficiency.

Types of Devices

Klystrons can be classified according to the following basic parameters:

Figure 6.31 Plot of the relative phase of the reference electrons as a function of axial
distance in a high-efficiency klystron.

Figure 6.32 Plot of the normalized RF beam currents as a function of distance along the
length of a high-efficiency klystron.

© 1999 CRC Press LLC

·

Power operating level. Klystrons are available ranging from a few hundred watts
to more than 10 MW.

·

Operating frequency. Klystrons typically are used over the frequency range of
300 MHz to 40 GHz.

·

Number of cavities. The number of resonant cavities may range from one to five
or more. Furthermore, the cavities may be integral or external to the vacuum en-
velope of the device.

The klystron is a true linear amplifier from zero signal level up to 2 to 3 dB below

saturated output.

Figure 6.33

shows a typical transfer characteristic for a klystron. RF

modulation is applied to the input drive signal. Amplitude modulation typically is lim-
ited to the linear portion of the gain transfer characteristic (class A operation). The re-
sult is low efficiency, because the beam power is always on. For applications requiring
frequency modulation, the drive power is set for saturated output.

Pulse modulation of the klystron may be obtained by applying a negative rectangular

voltage, instead of a dc voltage, to the cathode. The RF drive, set to a saturation value,
usually is pulsed on for a slightly shorter time than the beam pulse.

Because of the high power levels typically used at UHF frequencies, device effi-

ciency is a critical parameter. Klystrons usually are rated in terms of saturated effi-
ciency, determined by dividing the saturated RF output power by the dc input power.

Saturated efficiency governs the maximum peak-of-sync efficiency available when

beam-pulsing techniques are employed for UHF-TV service. Peak-of-sync efficiency
is the commonly used figure of merit (FOM) expression, defined as the peak-of-sync
output power divided by the dc input power.

Figure 6.33 Typical gain, output power, and drive requirements for a klystron.

© 1999 CRC Press LLC

Table 6.1

lists typical operating parameters for an integral-cavity klystron.

6.3.4 Beam Pulsing

Beam pulsing is a common method of improving the efficiency of a broadband linear
klystron amplifier. Depending on the transmitted waveform, efficiency may be
boosted by 25 percent or more. This technique typically is used in UHF-TV transmit-
ters to reduce visual klystron beam dissipation during video portions of the transmis-
sion. Sync pulsing, as the technique is commonly known, is accomplished by chang-
ing the operating point of the tube during the synchronizing interval, when peak
power is required, and returning it to a linear transfer characteristic during the video
portion of the transmission.

This control is accomplished through the application of a voltage to an electrode

placed near the cathode of the klystron. Biasing toward cathode potential increases the
beam current, and biasing toward ground (collector potential) decreases beam current.

In the composite TV waveform, video information occupies 75 percent of the ampli-

tude, and sync occupies the remaining 25 percent. The tip of sync represents the peak
power of the transmitted waveform. Black (the blanking level) represents 56 percent of
the peak power. If the blanking level could be made to represent 100 percent modula-
tion and the sync pulsed in, as in a radar system, efficiency would be increased signifi-
cantly. Unfortunately, the color-burst signal extends 50 percent into the sync region,
and any attempt to completely pulse the sync component would distort the color-burst
reference waveform. Sync pulsing is, therefore, limited to 12.5 percent above black to
protect color-burst. Two common implementations of beam pulsing can be found:

Table 6.1 Typical Operating Parameters for an Integral-Cavity Klystron for UHF-TV
Service (

Courtesy of Varian.)

Parameter

Typical Value

Operating frequency (visual)

519 MHz

Output power, peak-of-sync

64 kW

Drive power, peak-of-sync

15 W

Gain, peak-of-sync

36 dB

Efficiency, saturated

55%

Bandwidth, 1 dB

6 MHz

Beam voltage

24.5 kV

Beam current

4.8 A

Body current

10 MA dc

Modulating-anode voltage

17 kV dc

Modulating-anode current

0.5 mA dc

Focusing current

30 A dc

Load VSWR

> 1.1:1

Collector temperature

130°C

© 1999 CRC Press LLC

·

Modulating-anode (mod-anode) pulsing

·

Annular beam control electrode (ACE) pulsing

The mod-anode system (first discussed in Section 6.3.3) utilizes an additional elec-
trode after the cathode to control beam power. The ACE-type tube operates on a simi-
lar principle, but the annular ring is placed close to the cathode so that the ring en-
closes the electron beam. Because of the physical design, pulsing with the ACE-type
gun is accomplished at a much lower voltage than with a mod-anode device. The ACE
element, in effect, grid-modulates the beam.

In theory, the amount of beam-current reduction achievable and the resulting effi-

ciency improvement are independent of whether mod-anode or ACE-type pulsing are
used. In practice, however, differences are noted. With existing mod-anode pulsers, an
efficiency improvement of about 19 percent in beam current over nonpulsed operation
may be achieved. A beam reduction of 30 to 35 percent may be achieved through use of
an ACE-equipped tube. The effect of ACE voltage on beam current is shown in

Figure

6.34

for an external-cavity klystron.

The ACE electrode, positioned in the gun assembly, is driven by a negative-directed

narrowband video signal of a few hundred volts peak. The annular ring varies the beam
density through a pinching action that effectively reduces the cross-sectional area of
the stream of electrons emitted by the cathode. The klystron thus operates in a
quasi-class-AB condition rather than the normal class A (for linear TV service).

A peak-of-sync FOM for an integral-cavity UHF-TV klystron without ACE or

equivalent control is 0.67 to 0.68. Through the use of an ACE-type tube, the FOM may
be increased to 0.80. Similar improvements in efficiency can be realized for exter-
nal-cavity klystrons. A typical switching-type mod-anode pulser is shown in

Figure

6.35

.

Pulsing is not without its drawbacks, however. The greater the pulsing, the greater

the precorrection required from the modulator. Precorrection is needed to compensate
for nonlinearities of the klystron transfer characteristic during the video modulation pe-
riod. Level-dependent RF phase precorrection also may be required. Switching be-
tween different klystron characteristics produces phase modulation of the visual car-
rier. If not corrected, this phase modulation may result in intercarrier "buzz" in the re-
ceived audio. These and other considerations limit the degree of pulsing that may be
achieved on a reliable basis.

6.3.5 Integral vs. External Cavity

In an integral-cavity klystron, the resonant cavities are located within the vacuum en-
velope of the tube. In an external-cavity klystron, the cavities are located outside the
envelope in a mechanical assembly that wraps around the drift tube.

Fundamentally, klystron theory applies equally to integral- and external-cavity

tubes. In both cases, a velocity-modulated electron beam interacts with multiple reso-
nant cavities to provide an amplified output signal. The resonant cavity interaction gap
and drift length requirements for optimum performance--including conversion effi-
ciency--are independent of whether the tuning mechanism is inside or outside the vac-

© 1999 CRC Press LLC

Figure 6.35 Block diagram of a switching mod-anode pulser.

Figure 6.34 Klystron beam current as a function of annular control electrode voltage.

© 1999 CRC Press LLC

uum envelope. High-efficiency integral- and external-cavity klystrons have been de-
signed to provide maximum conversion performance consistent with signal bandwidth
requirements. The saturated conversion efficiency is essentially identical for integral-
or external-cavity klystrons with equal numbers of resonant cavities.

Number of Cavities

The relative benefits of integral- vs. external-cavity klystrons can be debated at
length. Discussion typically centers on operating efficiency, life expectancy, and re-
placement device cost. In a comparison of the efficiency of integral- vs. external-cav-
ity klystrons for UHF-TV applications, however, the question really boils down to the
number of cavities (four or five) used in the device. Four-cavity external tubes are
standard. It is usually not practical to produce high-power five-cavity external tubes
for mechanical and electrical (voltage standoff) reasons.

Integral devices permit the addition of a fifth cavity to the design for two primary

reasons. First, the device itself is mechanically more robust. The addition of a fifth cav-
ity to an external klystron increases the length and weight of the device. Because of the
size of the drift tube in an external device designed for UHF-TV frequencies, the addi-
tional weight is difficult to support. Second, because the cavities are enclosed in a vac-
uum in the integral design, voltage standoff problems are greatly reduced.

The tuning mechanism of an integral-cavity klystron is enclosed in a rigid steel

shell. In the external-cavity unit, the tuning mechanism is in air. Within the region of the
tuning mechanism, RF fields can reach high levels, especially at the high end of the
UHF-TV band. The highest energy field occurs in the area of the last cavity. Because of
the high dielectric properties of a vacuum, dielectric breakdown is less of a problem in
the integral design, which places the tuning mechanism within a vacuum envelope.
This situation is of particular concern at the high end of the UHF band, where the spac-
ing of tuning elements is closer.

Four-cavity integral-type klystrons are manufactured, but for power outputs of 30

kW or less. The 60 kW integral klystron, typically used for UHF-TV broadcasting, is
produced as a five-cavity design.

The physics of the integral- and external-cavity klystrons are essentially the same. In

a comparison of integral and external units of like design--that is, with the same num-
ber of cavities--performance should be identical. The two units follow the same laws
of physics and use basically the same components up to the beam stick. It is when man-
ufacturers take advantage of the relative merits of each design that differences in per-
formance are realized.

Efficiency

For the sake of comparison, the data presented in this section will assume:

·

The power level is 60 kW

·

All integral-cavity devices utilize five cavities

·

All external-cavity devices utilize four cavities

© 1999 CRC Press LLC

The five-cavity integral S-tuned klystron is inherently at least 20 percent more effi-

cient than a four-cavity tube. (S-tuning refers to the method of stagger tuning the cavi-
ties.) The five-cavity klystron generally is specified by the manufacturer for a mini-
mum efficiency of 52 percent (saturated efficiency). Typical efficiency is 55 percent.
Four-cavity devices are characteristically specified at 42 percent minimum and 45 per-
cent typical.

This efficiency advantage is possible because the fifth cavity of the integral design

permits tuning patterns that allow maximum transfer of RF energy while maintaining
adequate bandpass response. Tighter bunching of electrons in the beam stick, a func-
tion of the number of cavities, also contributes to the higher efficiency operation. In ac-
tuality, the fifth cavity allows design engineers to trade gain for efficiency. Still, the
five-cavity tube has significantly more gain than the four-cavity device. The five-cav-
ity unit, therefore, requires less drive, which simplifies the driving circuit. A five-cav-
ity klystron requires approximately 25 W of drive power, while a four-cavity tube needs
as much as 90 W for the same power output.

Under pulsed operation, approximately the same reduction in beam current is real-

ized with both integral- and external-cavity klystrons. In a comparison of peak-of-sync
FOM, the efficiency differences will track. There is, fundamentally, no reason that one
type of klystron should pulse differently than the other.

Performance Tradeoffs

It is the designer's choice whether to build a transmitter with the klystron cavities lo-
cated inside or outside the vacuum envelope. There are benefits and drawbacks to
each approach.

When the cavity resonators are a part of the tube, the device becomes more compli-

cated and more expensive. However, the power generating system is all together in one
package, which simplifies installation significantly.

When the resonator is separate from the tube, as in an external-cavity device, it can

be made with more compliance (greater room for adjustment). Consequently, a single
device may be used over a wider range of operating frequencies. In terms of UHF-TV, a
single external-cavity device may be tuned for operation over the entire UHF-TV band.
This feature is not possible if the resonant cavities are built into the device. To cover the
entire UHF-TV band, three integral-cavity tubes are required. The operational divi-
sions are:

·

Channels 14 to 29 (470 to 566 MHz)

·

Channels 30 to 51 (566 to 698 MHz)

·

Channels 52 to 69 (698 to 746 MHz)

This practical limitation to integral-cavity klystron construction may be a drawback

for some facilities. For example, it is not uncommon for group operations to share one
or more spare klystrons. If the facilities have operating frequencies outside the limits of
a single integral device, it may be necessary to purchase more than one spare. Also,

© 1999 CRC Press LLC

when the cavities are external, the resonators are in air and can be accessed to permit
fine adjustments of the tuning stages for peak efficiency.

The advantages of tube changing are significant with an integral device. Typical

tube change time for an integral klystron is 1 hour, as opposed to 4 to 6 hours for an ex-
ternal device. The level of experience of the technician is also more critical when an ex-
ternal-type device is being changed. Tuning procedures must be carefully followed by
maintenance personnel to avoid premature device failure.

6.3.6 MSDC Klystron

Developmental work on the multistage depressed collector (MSDC) klystron began in
the mid-1980s. The project

1

produced a working tube capable of efficiency in UHF

service that had been impossible with previous klystron-based technology.

The MSDC device may be used in a number of varied applications. NASA originally

became involved in the project as a way to improve the efficiency of satellite transmit-
ters. With limited power available onboard a space vehicle, efficient operation is criti-
cally important. Such transmitters traditionally operate in a linear, inefficient mode.
UHF-TV broadcasters were interested in the MSDC because it promised to reduce the
huge operating costs associated with high-power operation.

Experimentation with depressed collector klystrons dates back to at least the early

1960s. Early products offered a moderate improvement in efficiency, but at the price of
greater mechanical and electrical complexity. The MSDC design, although mechani-
cally complex, offers a significant gain in efficiency.

Theory of Operation

MSDC tubes have been built around both integral-cavity and external-cavity kly-
strons. The devices are essentially identical to a standard klystron, except for the col-
lector assembly. Mathematical models provided researchers with detailed information
on the interactions of electrons in the collector region. Computer modeling also pro-
vided the basis for optimization of a beam-reconditioning scheme incorporated into
the device. Beam reconditioning is achieved by including a transition region between
the RF interaction circuit and the collector under the influence of a magnetic field. It
is interesting to note that the mathematical models made for the MSDC project trans-
lated well into practice when the actual device was constructed.

From the electrical standpoint, the more stages of a multistage depressed collector

klystron, the better. The tradeoff, predictably, is increased complexity and, therefore,
increased cost for the product. There is also a point of diminishing returns that is
reached as additional stages are added to the depressed collector system. A four-stage
device was chosen for TV service because of these factors. As more stages are added
(beyond four), the resulting improvement in efficiency is proportionally smaller.

1

A joint effort of the National Aeronautics and Space Administration (NASA), several
UHF-TV transmitter manufacturers, Varian Associates, and other concerns.

© 1999 CRC Press LLC

Figure 6.36

shows the mechanical configuration of the four-stage MSDC klystron.

Note the "V" shape that was found, through computer modeling, to provide the best
capture performance, minimizing electron feedback. A partially assembled collector
assembly is shown in

Figure 6.37

.

Because the MSDC device is identical to a conventional klystron except for the col-

lector, efficiency improvement techniques used for klystrons can be incorporated into
the MSDC. ACE-type pulsing commonly is used to improve efficiency in TV applica-
tions.

Figure 6.38

illustrates the effects of ACE voltage on beam current.

Electron Trajectories

The dispersion of electrons in the multistage collector is the key element in recovering
power from the beam and returning it to the power supply. This is the mechanism that
permits greater operating efficiency from the MSDC device.

Figure 6.39

illustrates the dispersion of electrons in the collector region during a car-

rier-only operating mode. Note that there is little dispersion of electrons between stages
of the MSDC. Most are attracted to electrode 4, the element at the lowest potential
(6.125 kV), referenced to the cathode.

Figure 6.36 Mechanical design of the multistage depressed collector assembly. Note
the "V" shape of the 4-element system.

© 1999 CRC Press LLC

Figure 6.40

shows collector electron trajectories at 25 percent saturation. The elec-

trons exhibit predictable dispersion characteristics during the application of modula-
tion, which varies the velocity of the electrons. This waveform is a reasonable approxi-
mation of average modulation for a typical video image.

Figure 6.37 A partially assembled MSDC collector. (

Courtesy of Varian.)

Figure 6.38 Beam current as a function of annular control electrode voltage for an
MSDC klystron.

© 1999 CRC Press LLC

Figure 6.41

shows electron trajectories at 50 percent saturation, approximately the

blanking level. Note the increased number of electrons attracted to electrodes 2 and 3,
the higher-potential electrodes (referenced to the cathode).

Figure 6.42

illustrates electron dispersion at 90 percent saturation, approximately

the level of sync in a video waveform. As the modulation level is increased, more elec-
trons are attracted to the higher-voltage electrodes. The dramatic increase in electron
capture by electrode 1, the highest-potential element of the device (at a voltage of 24.5
kV, referenced to the cathode), can be observed.

The electrons, thus, sort themselves out in a predictable manner. Notice the arc that

is present on many electron traces. The electrons penetrate the electrostatic field of the
collector, then are pulled back to their respective potentials.

A savings in power is realized because the electrostatic forces set up in the MSDC

device slow down the electrons before they contact the copper collector electrode. The
heat that would be produced in the collector is, instead, returned to the power supply in

Figure 6.39 Collector electron trajectories for the carrier-only condition. Note that
nearly all electrons travel to the last electrode (4), producing electrode current

I

4

.

Figure 6.40 Collector trajectories at 25 percent saturation. With the application of mod-
ulation, the electrons begin to sort themselves out.

© 1999 CRC Press LLC

the form of electric energy. In theory, peak efficiency would occur if the electrons were
slowed down to zero velocity. In practice, however, that is not possible.

Figure 6.43

shows the distribution of collector current as a function of drive power.

With no RF drive, essentially all current goes to electrode 4, but as drive is increased, I

4

drops rapidly as collector current is distributed among the other elements. Note that the
current to electrode 5 (cathode potential) peaks at about 10 percent of beam current.
This suggests that the secondary yield of the collector surfaces is within acceptable lim-
its.

Inserted between the klystron and the collector assembly is a refocusing electromag-

net that controls the electron beam as it enters the collector region.

Figure 6.41 Collector electron trajectories at 50 percent saturation, approximately the
blanking level. The last three electrodes (2, 3, and 4) share electrons in a predictable
manner, producing currents

I

2

,

I

3

, and

I

4

.

Figure 6.42 Collector trajectories at 90 percent saturation, the sync level. Note the sig-
nificant increase in the number of electrons attracted to the first electrode, producing

I

1

.

© 1999 CRC Press LLC

Mechanical Construction

The completed MSDC assembly is shown in

Figure 6.44

with the collector shield par-

tially removed to allow visibility of the collector elements. The collector of the
four-stage MSDC design actually is composed of five elements mounted between ce-
ramic rings for electrical insulation. The fifth electrode is at cathode potential.

Cooling for the MSDC is, not surprisingly, more complicated than for a conven-

tional klystron. The tradeoff, however, is that there is less heat to remove because of the
higher efficiency of the device. Water cooling is provided on each electrode of the
MSDC tube.

Figure 6.45

illustrates the overall mechanical design of an external-cavity version of

the MSDC tube, including placement of the device in its cavity bay.

Although the MSDC has not been in service for a sufficient length of time to com-

pletely characterize the product life expectancy, researchers believe that the MSDC de-
sign will have little, if any, effect on the lifetime of the klystron. The electron beam is es-
sentially unchanged. The tube is identical to a conventional integral- or external-cavity
klystron except in collector assembly.

Figure 6.43 The distribution of electrode current as a function of drive power. Note the
significant drop in

I

4

as drive power is increased.

I

5

is the electrode at cathode potential.

© 1999 CRC Press LLC

MSDC Power Supply

Design criteria for the collector power supply system provide a mixed bag of require-
ments. The critical parameter is the degree of regulation between the cathode and an-
ode. The relative differences between the elements of the collector are not, in most ap-
plications, significant. Consequently, the bulk of the power supplied to the tube does
not need to be well regulated. This is in contrast with conventional klystron operation,

Figure 6.44 The MSDC collector assembly with
the protective shield partially removed. (

Courtesy

of Varian.)

Figure 6.45 Mechanical construction of an external-cavity MSDC tube: (

a) device, (b)

device in cavity assembly.

(

a)

(

b)

© 1999 CRC Press LLC

in which the entire beam power supply must be regulated. This factor effectively de-
creases the amount of power that must be regulated to 1 to 2 percent of the dc input,
offsetting to some extent the additional cost involved in constructing multiple sup-
plies to facilitate the 4-stage MSDC design. Two approaches can be taken to collector
power supply design:

·

Parallel arrangement, shown in

Figure 6.46

·

Series arrangement, shown in

Figure 6.47

Note that, in both cases, the collector electrodes are stepped at a 6.125 kV potential
difference for each element.

Device Performance

The efficiency improvement of the MSDC klystron over the conventional klystron is
impressive. For TV applications, the MSDC--in pulsed operation--is more than
twice as efficient as a conventional klystron.

The bandpass performance of the device is another critical parameter.

Figure 6.48

charts power output as a function of frequency and RF drive at full power. (Drive power
is charted from 0.5 to 16 W.)

Figure 6.49

charts power output as a function of frequency

Figure 6.46 Parallel configuration of the MSDC power supply. Each supply section has
an output voltage that is an integral multiple of 6.125 kV.

© 1999 CRC Press LLC

and RF drive with the tube in a beam-pulsing mode. (Drive power is charted from 0.5 to
32 W.) Note that the traces provide good linearity over a 6 MHz bandwidth. Gain, as a
function of frequency and power, is essentially constant and undisturbed.

Table 6.2

lists typical operating parameters for an external-cavity MSDC device.

Applying the MSDC Klystron

A block diagram of a typical system is shown in

Figure 6.50

. The 60 kW TV transmit-

ter incorporates two external-cavity MSDC klystrons, one for the visual and another
for the aural. Design of the transmitter is basically identical to a non-MSDC system,
with the exception of the power supply and cooling system. The efficiency available
from the MSDC makes further device improvements subject to the law of diminishing
returns, as mentioned previously. Support equipment begins to consume an increas-
ingly large share of the power budget as the output device efficiency is improved.
Tuning of the MSDC klystron is the same as for a conventional klystron, and the same
magnetic circuit is typically used.

The power supply arrangement for the example MSDC system is shown in

Figure

6.51

. A series beam supply was chosen for technical and economic reasons. Although

the current to each collector varies widely with instantaneous output level, the total cur-
rent stays within narrow limits. Using a single transformer for all supplies, therefore,
minimizes the size of the iron core required. A 12-pulse rectifier bank provides low rip-

Figure 6.47 Series configuration of the MSDC power supply, which uses four identical
6.125 kV power supplies connected in series to achieve the needed voltages.

© 1999 CRC Press LLC

Figure 6.48 Device bandwidth as a function of frequency and drive power. Beam volt-
age is 24.5 kV, and beam current is 5.04 A for an output power of 64 kW. These traces
represent the full-power test of the MSDC device.

Figure 6.49 Device bandwidth in the beam-pulsing mode as a function of frequency
and drive power. Beam voltage is 24.5 kV, and beam current is 3.56 A with an output
power of 34.8 kW.

© 1999 CRC Press LLC

ple and reduces the need for additional filtering. The size and complexity of the recti-
fier stack are increased little beyond a normal beam supply because the total potential
of the four supplies is similar to that of a normal klystron transmitter (24.5 to 27.5 kV).

The collector stages of the MSDC device are water-cooled by a single water path

that loops through each electrode element. Because high voltage is present on the indi-
vidual elements, purity of the water is critical to proper operation. A two-stage system
is used with a water-to-water plate heat exchanger separating the primary and second-
ary systems.

6.4

Klystrode/Inductive Output Tube (IOT)

The Klystrode tube is the result of a development program started in 1980

2

with

UHF-TV in mind as a primary application. The basic concept of the Klystrode dates
back to the late 1930s,

3

but it was not until the early 1980s that serious engineering ef-

fort was put into the tube to make it a viable product for high-power UHF service. The
fundamental advantage of the Klystrode, also known as the inductive output tube

Table 6.2 Typical Operating Parameters for an External-Cavity MSDC Klystron for
UHF-TV service (

Courtesy of Varian.)

Parameter

Typical Value

Operating frequency (visual)

519 MHz

Output power, peak-of-sync

64 kW

Drive power, peak-of-sync

20 W

Gain, peak-of-sync

35 dB

Figure of merit

1.3

1

Bandwidth, 1 dB

6 MHz

Beam voltage

24.5 kV

Beam current

5.3 A

Body current

50 mA dc

Modulating-anode voltage

19.5 kV dc

Modulating-anode current

0.5 mA dc

Focusing current

11 A dc

Ion pump voltage

3.2 kV dc

Load VSWR

> 1.1:1

Refocusing coil current

7 A dc

Refocusing coil voltage

10 V dc

1

Figure of merit

is equal to the quotient of peak-of-sync output power and the average dc beam

input power.

2

Varian/Eimac

3

Andrew Haeff, 1938

© 1999 CRC Press LLC

Figure 6.51 Simplified schematic diagram of the power supply for an MSDC klystron
transmitter. A single high-voltage transformer with multiple taps is used to provide the
needed collector voltage potentials.

Figure 6.50 Simplified block diagram of a 60 kW (TVT) MSDC transmitter. The aural
klystron may utilize a conventional or MSDC tube at the discretion of the user.

© 1999 CRC Press LLC

(IOT), is its ability to operate class B. The result is higher efficiency when compared
with a conventional klystron.

6.4.1 Theory of Operation

As its name implies, the Klystrode tube is a hybrid between a klystron and a tetrode.
The high reliability and power-handling capability of the klystron is due, in part, to
the fact that electron beam dissipation takes place in the collector electrode, quite sep-
arate from the RF circuitry. The electron dissipation in a tetrode is at the anode and
the screen grid, both of which are an inherent part of the RF circuit and, therefore,
must be physically small at UHF frequencies. The tetrode, however, has the advantage
that modulation is produced directly at the cathode by a grid so that a long drift space
is not required to produce density modulation. The Klystrode/IOT has a similar ad-
vantage over the klystron--high efficiency in a small package [4]. The Klystrode tube
is a registered trademark of Varian. The IOT is the generic description for the device,
and it will be used in this discussion.

The IOT is shown schematically in

Figure 6.52

. The electron beam is formed at the

cathode, density-modulated with the input RF signals by a grid, then accelerated
through the anode aperture. In its bunched form, the beam drifts through a field-free re-
gion, then interacts with the RF field in the output cavity. Power is extracted from the
beam in the same way as in a klystron. The input circuit resembles that of a typical UHF
power grid tube. The output circuit and collector resemble those in a klystron.

A production version of a 60 kW device is shown in

Figure 6.53

. Double-tuned cavi-

ties are used to obtain the required operating bandwidth. The load is coupled at the sec-
ond cavity, as shown in

Figure 6.54

. This arrangement has proved to be an attractive

way to couple power out of the device because no coupling loop or probe is required in
the primary cavity, which can be a problem at the high end of the UHF band.

Because the IOT provides both beam power variation during sync pulses (as in a

pulsed klystron) and variation of beam power over the active modulating waveform, it is
capable of high efficiency. The device provides full-time beam modulation as a result
of its inherent structure and class B operation.

6.4.2 Electron Gun

Many design elements from the klystron have been borrowed for use in the IOT [4]. A
barium aluminate cathode is used for the gun, taken from a conventional klystron to-
gether with its heater structure. The methods used to support the cathode, heater, and
necessary heat shields in the klystron gun have been retained for use in the IOT. Fig-
ure 6.55 shows the construction of the IOT gun and output cavity.

In the IOT, the RF input voltage is applied between the cathode and a grid that allows

extra electrons to be drawn from the cathode into a low-quiescent-current electron
beam according to the instantaneous RF voltage appearing between the grid and cath-
ode. The resulting density-modulated beam is then passed into the klystron-like RF
output interaction region of the tube.

© 1999 CRC Press LLC

The grid is clamped in place in front of the cathode, supported on a metal cylinder,

and isolated from the cathode by a ceramic insulator. A second ceramic insulator sup-
ports the complete gridded electron gun at the correct distance from the grounded an-

Figure 6.53 A 60 kW Klystrode tube mounted in
its support stand with the output cavity attached.
(

Courtesy of Varian.)

Figure 6.52 Simplified schematic diagram of the Klystrode tube.

© 1999 CRC Press LLC

ode. This ceramic insulator completes the vacuum envelope and holds off the full beam
voltage of approximately 30 kV. The IOT gun operates at cathode-to-anode voltages ap-
proximately 50 percent higher than those of conventional klystron guns. Stray capaci-
tance in the grid support structure has been minimized to reduce losses.

The grid-to-cathode space of the electron gun forms the end of a long and complex

RF transmission line from the RF input connector of the input cavity system. This de-
sign has a significant effect upon the final frequency range of the IOT input cavity. Be-
cause the grid-to-cathode distance is crucial to the physics of the device, the grid re-
quires a rigid support structure to maintain proper spacing at high operating tempera-
tures.

6.4.3 Grid Structure

A pyrolytic graphite grid is used in the IOT because of its strength and its ability to
maintain desired specifications at elevated temperatures [4]. Pyrolytic graphite has a
unique advantage over other common grid materials in that its strength increases as
the temperature climbs to 2500°C and above, whereas the strength of pure metals uni-
versally decreases as the temperature increases. This allows the designer to produce a
thin grid, with fine grid wires that may be accurately positioned and will retain their
position and shape when raised to operating temperatures of approximately 1000°C.

During production of the grid, a hydrocarbon, typically methane, is fed into a

low-pressure chamber containing a graphite rod of the correct form to produce the re-
quired graphite shell. A shell typically consists of a graphite cylinder with a closed,
shaped end. The cold gas is passed into a hot zone in the reactor, which is heated by an
RF eddy current system from outside the vessel. An ordered carbon structure, as op-
posed to an amorphous structure, is required. To provide the needed processing, the
temperature of the reactor is set to approximately 2000°C at a pressure of 10 torr. The
graphite shells produced in this way have a layered structure with anisotropic properties
and are physically durable.

Figure 6.54 A close-up view of the
double-tuned output cavity of a 60 kW
Klystrode tube. (

Courtesy of Varian.)

© 1999 CRC Press LLC

The shells are then machined to the desired shape and size, and holes are cut into the

resulting grid back using a laser under computer control. The process of producing a
pyrolytic graphite grid is illustrated in

Figure 6.56

.

6.4.4 Input Cavity

Although similar, the Klystrode and the IOT are not identical. There are both subtle
and significant differences in each design. The greatest departure can be seen in the
configuration of the input cavity [4]. The design of the IOT is shown in

Figure 6.57

. A

cylindrical resonant cavity containing an annular sliding tuning door is used. The cav-
ity is folded at the IOT electron gun end to make contact with the tube via an RF
choke structure. Because both the cathode and grid operate at beam potential (30 kV),
it is necessary to maintain the body of the cavity and its tuning mechanism at ground

Figure 6.55 Overall structure of the IOT gun and output cavity. (

After [4].)

© 1999 CRC Press LLC

potential by using RF chokes to prevent leakage of RF energy, while holding off the
full beam voltage.

At a given point on the grid connection within the RF choke, an insulated high-volt-

age cable is attached and exits so that the 30 kV dc can be supplied to the grid. This ca-
ble is screened and fitted with RF chokes to prevent UHF energy from passing into the
beam power supply. RF input energy is fed via a coaxial cable to the input cavity, which
is excited by means of a loop antenna carried by the annular tuning door.

Cooling air for the IOT electron gun and grid connection is fed through an insulated

air pipe down the inside of the input cavity.

The tuning door of the cavity is driven by three two-stage tuning screws coupled to

the outside via a rubber-toothed belt and bevel-gear drive. A mechanical turns-counter
is provided to relate cavity door position with operating frequency.

6.4.5 Output Cavity

The primary output cavity is clamped around the output ceramic of the IOT, as on a
conventional klystron [4]. The output cavity, shown in

Figure 6.58

, contains an RF

coupling loop that may be rotated about a horizontal axis to adjust the degree of cou-
pling through a short transmission line section into a secondary cavity via a door-
knob-type antenna. The secondary cavity contains a dome structure adjusted in size
so that the cavity can be made to cover the required frequency band. An output cou-
pler, of standard klystron design incorporating a loop antenna, connects the secondary
cavity of the RF output feeder system via a standard output interface. This design pro-
vides for instantaneous bandwidths of 8 MHz or greater.

Because RF voltages on the order of the IOT beam voltage can be expected in the

output system, RF arc detectors are fitted into both cavities.

Cooling of the output cavities is accomplished by means of filtered forced air. Air

enters the primary cavity, passes over the output ceramic and coupling loop, then exits

Figure 6.56 Pyrolytic graphite grid: (

a) shell, (b) grid blank, (c) grid cut.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

via holes in the coupling hub assembly into the secondary cavity. After cooling the sec-
ondary cavity, some of the air exits via the contact fingers on the tuning doors, while a
small portion traverses the output loop and coupler, exiting at a stub-pipe fitting for that
purpose.

Typical operating parameters for the IOT are shown in

Table 6.3

.

6.4.6 Application Considerations

An active crowbar circuit is included in most applications of the Klystrode/IOT to
protect the pyrolytic graphite grid in the event of an arc condition inside the tube [4].
The crowbar provides the added benefit of reducing the amount of gas generated in
the tube during an arc. A block diagram of a typical crowbar is shown in

Figure 6.59

.

The response time of the crowbar is typically less than 10

µs. The peak current per-

missible through the discharge tube may be 3000 A or greater.

Because of the IOT's class B operation, the response of the power supply to a vary-

ing load is an important design parameter. If a standard klystron beam supply were used
in an IOT-equipped RF generator, performance under varying modulation levels could

Figure 6.57 IOT input cavity. (

After [4].)

© 1999 CRC Press LLC

be unacceptable. Consider the application of a bounce (black-to-white) signal to a TV
transmitter. Beam current for a typical IOT 60 kW transmitter would change from ap-
proximately 400 mA to 2 A. The effect with a conventional klystron supply would be
ringing of about 20 percent on the beam voltage. The IOT power supply, therefore, must
be designed for tight transient regulation.

Table 6.3 Typical Operating Parameters of the IOT in TV Service

Parameter

Typical Value

Beam voltage

28 kV

Mean beam current

1.15 A

Peak beam current

2.50 A

Grid bias

72 V

Grid current

1 mA

Body current

6 mA

Input power

399 W

Output power

40.3 kW

Figure 6.58 IOT output cavity. (

After [4].)

© 1999 CRC Press LLC

The IOT does not have the same hard saturation characteristic as a klystron. The

IOT's high-power transfer curve flattens more slowly than a klystron's and continues to
increase with increasing drive power, requiring a different approach to linearity correc-
tion.

6.4.7 Continuing Research Efforts

Several variations on the basic Klystrode/IOT theme have been developed, including
air-cooled devices operating at 30 kW and above. Air cooling is practical at these
power levels because of the improved efficiency that class B operation provides.

Research also is being conducted to extend the operating power of the IOT to 500

kW or more. Designed for scientific research applications, such devices offer numer-
ous benefits over conventional klystron technology. The IOT is much smaller than a
klystron of similar power, and it requires less support circuitry. Because of the im-
proved efficiency, power supply requirements are reduced, and device cooling is sim-
plified.

Figure 6.59 Simplified block diagram of the crowbar circuit designed into the Klystrode
SK-series (Comark) transmitters. The circuit is intended to protect the tube from poten-
tially damaging fault currents.

© 1999 CRC Press LLC

6.5

Constant Efficiency Amplifier

Because of the similarity between the spent electron beam in an IOT and that of a kly-
stron or traveling-wave tube, it is possible to consider the use of a multistage de-
pressed collector on an IOT to improve the operating efficiency [5]. This had been
considered by Priest and Shrader [6] and by Gilmore [7], but the idea was rejected be-
cause of the complexity of the multistage depressed collector assembly and because
the IOT already exhibited fairly high efficiency. Subsequent development by Symons
[5, 8] has led to a prototype device (at this writing). An inductive output tube, modi-
fied by the addition of a multistage depressed collector, has the interesting property of
providing linear amplification with (approximately) constant efficiency.

6.5.1 Theory of Operation

Figure 6.60

shows a schematic representation of the constant efficiency amplifier

(CEA) [5]. The cathode, control grid, anode and output gap, and external circuitry are
essentially identical with those of the IOT amplifier. Drive power introduced into the
input cavity produces an electric field between the control grid and cathode, which
draws current from the cathode during positive half-cycles of the input RF signal. For

Figure 6.60 Schematic overview of the MSDC IOT or

constant efficiency amplifier. (Af-

ter [5].)

© 1999 CRC Press LLC

operation as a linear amplifier, the peak value of the current--or more accurately, the
fundamental component of the current--is made (as nearly as possible) proportional
to the square root of the drive power, so that the product of this current and the voltage
it induces in the output cavity will be proportional to the drive power.

Following the output cavity is a multistage depressed collector in which several typi-

cal electron trajectories are shown. These are identified by the letters a through e. The
collector electrodes are connected to progressively lower potentials between the anode
potential and the cathode potential so that more energetic electrons penetrate more
deeply into the collector structure and are gathered on electrodes of progressively lower
potentials.

In considering the difference between an MSDC IOT and an MSDC klystron, it is

important to recognize that in a class B device, no current flows during the portion of
the RF cycle when the grid voltage is below cutoff and the output gap fields are acceler-
ating. As a result, it is not necessary to have any collector electrode at a potential equal
to or below cathode potential. At low output powers, when the RF output gap voltage is
just equal to the difference in potential between the lowest-potential collector electrode
and the cathode, all the current will flow to that electrode. Full class B efficiency is thus
achieved under these conditions.

As the RF output gap voltage increases with increased drive power, some electrons

will have lost enough energy to the gap fields so they cannot reach the lowest potential
collector, and so current to the next-to-the-lowest potential electrode will start increas-
ing. The efficiency will drop slightly and then start increasing again until all the current
is just barely collected by the two lowest-potential collectors, and so forth.

Maximum output power is reached when the current delivered to the output gap is

sufficient to build up an electric field or voltage that will just stop a few electrons. At
this output power, the current is divided between all of the collector electrodes and the
efficiency will be somewhat higher than the efficiency of a single collector, class B am-
plifier. Computer simulations have demonstrated that it is possible to select the collec-
tor voltages so as to achieve very nearly constant efficiency from the MSDC IOT device
over a wide range of output powers [5].

The challenge of developing a multistage depressed collector for an IOT is not quite

the same as that of developing a collector for a conventional klystron [8]. It is different
because the dc component of beam current rises and falls in proportion to the square
root of the output power of the tube. The dc beam current is not constant as it is in a kly-
stron (or a traveling-wave tube for that matter). As a result, the energy spread is low be-
cause the output cavity RF voltage is low at the same time that the RF and dc beam cur-
rents are low. Thus, there will be small space-charge forces, and the beam will not
spread as much as it travels deep into the collector toward electrodes having the lowest
potential. For this reason, the collector is likely to be rather long and thin when com-
pared to the multistage depressed collector for a conventional klystron, as described
previously.

Figure 6.61

charts the calculated efficiency for an IOT with six depressed collector

stages, set at 0.1, 0.2, 0.3, 0.45, 0.7, and 1.0 times the beam voltage (compared to that of
a conventional IOT).

© 1999 CRC Press LLC

6.6

Traveling Wave Tube

The traveling wave tube (TWT) is a linear-beam device finding extensive applications
in communications and research. Power levels range from a few watts to 10 MW. Gain
ranges from 40 to 70 dB for small drive signals. The TWT consists of four basic ele-
ments:

·

Electron gun. The gun forms a high-current-density beam of electrons that inter-
act with a wave traveling along the RF circuit to increase the amplitude of the RF
signal. In a typical application, electrons are emitted from a cathode and con-
verged to the proper beam size by focusing electrodes.

·

RF interaction circuit. The RF wave is increased in amplitude as a result of inter-
action with the electron beam from the gun. The fundamental principle on which
the TWT operates is that an electron beam, moving at approximately the same ve-
locity as an RF wave traveling along a circuit, gives up energy to the RF wave.

·

Magnetic electron beam focusing system. The beam size is maintained at the
proper dimensions through the interaction structure by the focusing system. This
may be accomplished by using either a permanent magnet or an electromagnetic
focusing element.

·

Collector. The electron beam is received at the collector after it has passed
through the interaction structure. The remaining beam energy is dissipated in the
collector.

Figure 6.61 Calculated efficiency of an MSDC IOT compared to an unmodified IOT.
(

After [5].)

© 1999 CRC Press LLC

Figure 6.62

shows the basic elements of a TWT.

The primary differences between types of TWT devices involve the RF interaction

structure employed. In

Figure 6.62

, the interaction structure is a helix. A variety of

other structures may be employed, depending on the operating power and frequency.
Three common approaches are used to provide the needed magnetic beam focusing. Il-
lustrated in

Figure 6.63

, they are:

·

Electromagnetic focusing, used primarily on high-power tubes, where tight beam
focusing is required.

·

Permanent-magnet focusing, used where the interaction structure is short.

·

Periodic permanent-magnet focusing, used on most helix TWT and coupled-cav-
ity tubes. The magnets are arranged with alternate polarity in successive cells
along the interaction region.

6.6.1 Theory of Operation

The interaction structure acts to slow the RF signal so that it travels at approximately
the same speed as the electron beam. Electrons enter the structure during both posi-
tive and negative portions of the RF cycle. Electrons entering during the positive por-
tion are accelerated; those entering during the negative portion are decelerated. The
result is the creation of electron bunches that produce an alternating current superim-
posed on the dc beam current. This alternating current induces the growth of an RF
circuit wave that encourages even tighter electron bunching.

One or more severs are included to absorb reflected power that travels in a backward

direction on the interaction circuit. This reflected power is the result of a mismatch be-
tween the output port and the load. Without the sever, regenerative oscillations could
occur.

Figure 6.62 Basic elements of a traveling wave tube.

© 1999 CRC Press LLC

At a given frequency, a particular level of drive power will result in maximum

bunching and power output. This operating point is referred to as saturation.

Interaction Circuit

The key to TWT operation lies in the interaction element. Because RF waves travel at
the speed of light, a method must be provided to slow down the forward progress of
the wave to approximately the same velocity as the electron beam from the cathode.
The beam speed of a TWT is typically 10 to 50 percent the speed of light, correspond-
ing to cathode voltages of 4 to 120 kV. Two mechanical structures commonly are used
to slow the RF wave:

·

Helix circuit. The helix is used where bandwidths of an octave or more are re-
quired. Over this range the velocity of the signal carried by the helix is basically
constant with frequency. Typical operating frequencies range from 500 MHz to
40 GHz. Operating power, however, is limited to a few hundred watts. TWTs in-
tended for higher-frequency operation may use a variation of the helix, shown in

Figure 6.64

. The ring-loop and ring-bar designs permit peak powers of hundreds

of kilowatts. The average power, however, is about the same as that of a conven-
tional helix because the structure used to support the interaction circuit is the
same.

Figure 6.63

Magnetic focusing for a TWT: (

a) solenoid-type, (b) perma-

nent-magnet-type, (

c) periodic permanent-magnet structure.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

·

Coupled-cavity circuit. The coupled-cavity interaction structure permits opera-
tion at high peak and average power levels, and moderate bandwidth (10 percent
being typical). TWTs using coupled-cavity structures are available at operating
frequencies from 2 to 100 GHz. The basic design of a coupled-cavity interaction
circuit is shown in

Figure 6.65

. Resonant cavities, coupled through slots cut in the

cavity end walls, resemble a folded waveguide. Two basic schemes are used: the
cloverleaf and the single-slot space harmonic circuit.

The cloverleaf, also known as the forward fundamental circuit, is used primarily on

high-power tubes. The cloverleaf provides operation at up to 3 MW peak power and 5
kW average at S-band frequencies. The single-slot space harmonic interaction circuit is
more common than the cloverleaf. The mechanical design is simple, as shown in the
figure. The single-slot space harmonic structure typically provides peak power of up to
50 kW and average power of 5 kW at X-band frequencies.

Pulse Modulation

The electron beam from the gun may be pulse-modulated using one of four methods:

·

Cathode pulsing. The cathode is pulsed in a negative direction with respect to the
grounded anode. This approach requires the full beam voltage and current to be
switched.

·

Anode pulsing. This approach is similar to cathode pulsing, except that the full
beam voltage is switched between cathode potential and ground. The current
switched, however, is only that value intercepted on the anode. Typically, the in-
tercepted current is a few percent of the full beam potential.

Figure 6.64 Helix structures for a TWT: (

a) ring-loop circuit, (b) ring-bar circuit.

(

a)

(

b)

© 1999 CRC Press LLC

·

Focus electrode pulsing. If the focus electrode, which normally operates at or
near cathode potential, is biased negatively with respect to the cathode, the beam
will be turned off. The voltage swing required is typically one-third of the full
cathode voltage. This approach is attractive because the focus electrode draws es-
sentially no current, making implementation of a switching modulator relatively
easy.

·

Grid pulsing. The addition of a grid to the cathode region permits control of beam
intensity. The voltage swing required for the grid, placed directly in front of the
cathode, is typically 5 percent of the full beam potential.

Electron Gun

The electron gun of a TWT is a device that supplies the electron beam to the tube [9].
A schematic diagram of a generic electron gun is given in

Figure 6.66

. The device

consists of a hot cathode heated by an electric heater, a negatively biased focusing
electrode or focuser, and a positively biased accelerating anode. The cross-sectional
view given in the figure can be a two-dimensional or three-dimensional coaxial struc-
ture [10, 11].

An axially symmetrical solid cylindrical electron beam is produced by the gun struc-

ture shown in

Figure 6.66

if the structure is axially cylindrically symmetrical. If the

middle of the hot cathode is made nonemitting and only the edge of the cathode is emit-
ting, the cathode becomes an annular cathode [9]. The annular cathode produces a hol-
low beam. The annular electron beam can be used to reduce beam current for a given
microwave output power.

If the gun structure shown in

Figure 6.66

is two dimensional, then a ribbon-shaped

electron beam is produced. A ribbon-shaped beam is used for a TWT of a two-dimen-
sional structure.

Figure 6.65 Coupled-cavity interaction structures: (

a) forward fundamental circuit or

"cloverleaf," (

b) single-slot space harmonic circuit.

(

a)

(

b)

© 1999 CRC Press LLC

If the angle of the focusing electrode against the axis of the electron beam is 67.5

°

and the anode is also tilted forward to produce a rectilinear flow (electron flow parallel
to the z axis in

Figure 6.66

), then such an electron gun is termed the Pierce gun.

In practice, the hot cathode surface is curved as shown in

Figure 6.67

to increase the

electron emitting surface and to obtain a high-density electron beam.

Beam Focusing

Electrons in an electron beam mutually repel each other by the electron's own cou-
lomb force because of their negative charge [9]. In addition, the electron beam usually
exists in proximity to the positively biased slow-wave structure, as shown in

Figure

6.68

. Therefore, the electron beam tends to diverge. The process of confining the

electron beam within the desired trajectory against the mutual repulsion and diverg-
ing force from the slow-wave structure is termed electron beam focusing.

The electron beam in a TWT is usually focused by a dc magnetic flux applied paral-

lel to the direction of the electron beam, which is coaxial to the slow-wave transmission
line. Variations on this basic technique include:

·

Brillouin flow, where the output of the electron gun is not exposed to the focusing
magnetic flux [12].

·

Immersed flow, where the electron gun itself is exposed to and unshielded from
the focusing para-axial longitudinal magnetic flux [12].

·

Generic flow, where the electron gun is not shielded from the focusing magnetic
flux, and focusing flux is not para-axia. (In other words, neither Brillouin nor im-
mersed flow.)

Collector Assembly

Various configurations are used for the collector assembly of a TWT.

Figure 6.69

shows a selection of the more common, including [9]:

Figure 6.66 Generic TWT electron gun
structure. (

From [9]. Used with permis-

sion.)

© 1999 CRC Press LLC

·

Plate collector

·

Cone collector

·

Curved cone

·

Cylinder collector

·

Depressed potential cylinder

·

Two-stage collector

·

Three-stage collector

Cooling options include conduction, air, and water.

Figure 6.68 Generic configuration of a traveling wave tube. (

From [9]. Used with per-

mission.)

Figure 6.67 Cross-sectional view of a TWT
electron gun with a curved hot cathode.
(

From [9]. Used with permission.)

© 1999 CRC Press LLC

Cooling of a low-power TWT is accomplished by clamping the tube to a metal base-

plate, mounted in turn on an air- or liquid-cooled heat sink. Coupled-cavity tubes below
1 kW average power are convection-cooled by circulating air over the entire length of
the device. Higher-power coupled-cavity tubes are cooled by circulating liquid over the
tube body and collector.

6.6.2 Operating Efficiency

Efficiency is not one of the TWT's strong points. Early traveling wave tubes offered
only about 10 percent dc-to-RF efficiency. Wide bandwidth and power output are
where the TWT shines. TWT efficiency may be increased in two basic ways: (1) col-
lector depression for a single-stage collector, or (2) use of a multistage collector.

Collector depression refers to the practice of operating the collector at a voltage

lower than the full beam voltage. This introduces a potential difference between the in-
teraction structure and the collector, through which electrons pass. The amount by
which a single-stage collector can be depressed is limited by the remaining energy of
the slowest electrons. In other words, the potential reduction can be no greater than the
amount of energy of the slowest electrons, or they will turn around and reenter the inter-
action structure, causing oscillations.

By introducing multiple depressed collector stages, still greater efficiency can be

realized. This method provides for the collection of the slowest electrons at one collec-
tor potential, while allowing those with more energy to be collected on other stages that
are depressed still further. This approach, similar to the MSDC design discussed previ-
ously, is illustrated in

Figure 6.70

.

Figure 6.69 Cross-sectional view of various collector configurations for a TWT. (

From

[9].

Used with permission.)

© 1999 CRC Press LLC

6.6.3 Operational Considerations

Although traveling wave tubes offer numerous benefits to the end user, they are not
without their drawbacks.

Intermodulation Distortion

TWTs are susceptible to intermodulation (IM) distortion when multiple carriers are
introduced, as in double illuminated or multiple SCPC (single channel per carrier)
satellite transponders. These IM products may be found at frequencies that are dis-
placed from the fundamental carriers by the difference in frequency between them.
When multiple carriers are present, the potential for IM exists. This potential is re-
duced by operating the TWT below saturation. Power must be reduced (backed off) in
proportion to the number of carriers and their relative power.

Second-Harmonic Content

Because of the wide bandwidth and nonlinear operating characteristics under satura-
tion conditions, a TWT may generate significant second-harmonic energy. It is not
uncommon to measure second-harmonic energy at the output of a TWT that is down
only 10 dB from the operating carrier.

Figure 6.70 Power supply configuration for a multistage depressed collector TWT.

© 1999 CRC Press LLC

Reduction of harmonic content usually involves injecting a coherent harmonic sig-

nal with controlled phase and amplitude along with the fundamental carrier so that they
interact, minimizing harmonic energy at the output of the device.

AM/PM Conversion

AM/PM conversion is the change in phase angle between the input and output signals
as the input varies in amplitude. The root cause of this distortion in a TWT centers on
the reduction of electron beam velocity as the input signal level increases. This causes
a greater energy exchange between the electron beam and the RF wave. At a level 20
dB below the input power required for saturation, AM/PM conversion is negligible.
At higher levels, AM/PM distortion may increase.

Phase Variation

When the velocity of the electron beam in the TWT is changed, the phase of the out-
put signal also will vary. The primary causes of beam velocity variations include
changes in one or more of the following:

·

Cathode temperature

·

Grid voltage

·

Anode voltage

·

Cathode voltage

The TWT power supply must be well regulated (to less than 0.2 percent) to prevent
beam velocity changes that may result in output signal phase variations.

6.7

Crossed-Field Tubes

A crossed-field microwave tube is a device that converts dc into microwave energy
using an electronic energy-conversion process. These devices differ from beam tubes
in that they are potential-energy converters, rather than kinetic-energy converters.
The term crossed field is derived from the orthogonal characteristics of the dc electric
field supplied by the power source and the magnetic field required for beam focusing
in the interaction region. This magnetic field typically is supplied by a perma-
nent-magnet structure. Such devices also are referred to as M-tubes.

Practical devices based on the crossed-field principles fall into two broad catego-

ries:

·

Injected-beam crossed-field tubes. The electron stream is produced by an elec-
tron gun located external to the interaction region, similar to a TWT. The concept
is illustrated in

Figure 6.71

.

·

Emitting-sole tubes. The electron current for interaction is produced directly
within the interaction region by secondary electron emissions, which result when
some electrons are driven to the negative electrode and allowed to strike it. The

© 1999 CRC Press LLC

negative electrode is formed using a material capable of producing significant
numbers of secondary-emission electrons. The concept is illustrated in

Figure

6.72

.

6.7.1 Magnetron

The magnetron encompasses a class of devices finding a wide variety of applications.
Pulsed magnetrons have been developed that cover frequency ranges from the low
UHF band to 100 GHz. Peak power from a few kilowatts to several megawatts has
been obtained. Typical overall efficiencies of 30 to 40 percent may be realized, de-
pending on the power level and operating frequency. CW magnetrons also have been
developed, with power levels of a few hundred watts in a tunable tube, and up to 25
kW or more in a fixed-frequency device. Efficiencies range from 30 percent to as
much as 70 percent.

The magnetron operates electrically as a simple diode. Pulsed modulation is ob-

tained by applying a negative rectangular voltage waveform to the cathode with the an-
ode at ground potential. Operating voltages are less critical than for beam tubes;
line-type modulators often are used to supply pulsed electric power. The physical struc-
ture of a conventional magnetron is shown in

Figure 6.73

.

High-power pulsed magnetrons are used primarily in radar systems. Low-power

pulsed devices find applications as beacons. Tunable CW magnetrons are used in ECM
(electronic countermeasures) applications. Fixed-frequency devices are used as micro-
wave heating sources.

Figure 6.71 Linear injected-beam microwave tube.

© 1999 CRC Press LLC

Tuning of conventional magnetrons is accomplished by moving capacitive tuners or

by inserting symmetrical arrays of plungers into the inductive portions of the device.
Tuner motion is produced by a mechanical connection through flexible bellows in the
vacuum wall. Tuning ranges of 10 to 12 percent of bandwidth are possible for pulsed
tubes, and as much as 20 percent for CW tubes.

Operating Principles

Most magnetrons are built around a cavity structure of the type shown in

Figure 6.74

.

The device consists of a cylindrical cathode and anode, with cavities in the anode that
open into the cathode-anode space--the interaction space--as shown. Power can be
coupled out of the cavities by means of a loop or a tapered waveguide.

Cavities, together with the spaces at the ends of the anode block, form the resonant

system that determines the frequency of the generated oscillations. The actual shape of
the cavity is not particularly important, and various types are used, as illustrated in Fig-
ure 6.75. The oscillations associated with the cavities are of such a nature that alternat-
ing magnetic flux lines pass through the cavities parallel to the cathode axis, while the
alternating electric fields are confined largely to the region where the cavities open into
the interaction space. The most important factors determining the resonant frequency
of the system are the dimensions and shape of the cavities in a plane perpendicular to
the axis of the cathode. Frequency also is affected by other factors such as the end space
and the axial length of the anode block, but to a lesser degree.

Figure 6.72 Reentrant emitting-sole crossed-field amplifier tube.

© 1999 CRC Press LLC

Figure 6.73 Conventional magnetron structure.

(

a)

(

b)

Figure 6.74 Cavity magnetron oscillator: (

a) cutaway view, (b) cross section view per-

pendicular to the axis of the cathode.

© 1999 CRC Press LLC

The magnetron requires an external magnetic field with flux lines parallel to the axis

of the cathode. This field usually is provided by a permanent-magnet or electromagnet.

The cathode is commonly constructed as a cylindrical disk.

Coaxial Magnetron

The frequency stability of a conventional magnetron is affected by variations in the
load impedance and by cathode-current fluctuations. Depending on the extent of
these two influences, the magnetron occasionally may fail to produce a pulse. The co-
axial magnetron minimizes these effects by using the anode geometry shown in Fig-
ure 6.76. Alternate cavities are slotted to provide coupling to a surrounding coaxial
cavity.

The oscillating frequency is controlled by the combined vane system and the reso-

nant cavity. Tuning may be accomplished through the addition of a movable end plate in
the cavity, as shown in

Figure 6.77

.

Frequency-Agile Magnetron

Tubes developed for specialized radar and ECM applications permit rapid tuning of
the magnetron. A conventional device may be tuned using one of the following meth-
ods:

·

A rapidly rotating capacitive element. Tubes of this type are referred to as
spin-tuned magnetrons.

·

A hydraulic-driven tuning mechanism. Tubes of this type are referred to as me-
chanically tuned magnetrons.

Electronic tuning of magnetrons is also possible, with tuning rates as high as several
megahertz per microsecond.

(

a)

(

b)

(

c)

Figure 6.75 Cavity magnetron oscillator anode: (

a) hole-and-slot type, (b) slot type, (c)

vane type.

© 1999 CRC Press LLC

Figure 6.76 Structure of a coaxial magnetron.

Figure 6.77 Structure of a tunable coaxial magnetron.

© 1999 CRC Press LLC

Linear Magnetron

Although the most common types of magnetrons are radial in nature, the linear mag-
netron and inverted magnetron may also be used, depending upon the application
[13]. A cross-sectional view of a linear magnetron is given in

Figure 6.78

. Shown in

the figure is the O-type linear magnetron, in which the electron beam emitted from
the electron gun is focused by a longitudinally applied dc magnetic flux density (B),
as in the case of the traveling wave tube.

As shown in the figure, a number of slots are included in the basic structure. These

slots are cut 1/4-wavelength deep, functioning as quarter-wave cavity resonators. This
structure forms a series of microwave cavity resonators coupling to an electron beam,
in a similar manner to the multicavity klystron. The velocity modulated electrons are
bunched, and the tightly bunched electrons produce amplified microwave energy at the
output cavity, which is coupled to an external circuit. The linear magnetron typically of-
fers high gain, but narrow frequency bandwidth.

6.7.2 Backward Wave Oscillator

In a traveling wave tube, if the microwave signal to be amplified is propagating in the
slow-wave structure backwardly to the direction of the electron beam, the device is
termed a backward wave oscillator (BWO) [13]. Microwaves traveling in a backward
direction carry positive feedback energy toward the electron gun and yield stronger
velocity modulation and bunching. Thus, the system is inherently an oscillator rather
than a stable amplifier. The input is typically terminated by an impedance-matched
reflectionless termination device. The oscillation frequency is a function of the speed
of the electrons and the time constant of the feedback mechanism. The speed of elec-
tron motion is controlled by the anode voltage.

An M-type radial BWO is shown in

Figure 6.79

. The direction of electron pole mo-

tion and the direction of microwave propagation along the annular reentrant type

Figure 6.78 Cross-sectional view of a linear magnetron. (

From [13]. Used with permis-

sion.)

© 1999 CRC Press LLC

slow-wave structure are opposite each other. It should be noted that the depths of the
slits cut in the inner surface of the anode is very shallow--much less than 1/4-wave-
length deep. In other words, the slits are not in resonance; they are not cavity resonators,
as is the case of a magnetron. Rather, the slits are nonresonating, as in the case of a
TWT. In the M-type radial BWO, the electron beam is focused by a magnetic flux den-
sity applied perpendicular to the beam, as seen from

Figure 6.79

.

An M-type radial BWO is sometimes termed the Carcinotron, a trade name. A key

feature of the Carcinotron is its wide voltage tunability over a broad frequency range.

6.7.3 Strap-Fed Devices

A radial magnetron can be configured so that every other pole of the anode resonators
are conductively tied for microwave potential equalization, as shown in

Figure 6.80a

[13]. These conducting tie rings are termed straps; the technique of using strap rings
is termed strapping. Strapping ensures good synchronization of microwaves in the
magnetron resonators with the rotation of electron poles.

The technique of strapping is extended and modified for an M-type radial BWO, as

shown in

Figure 6.80b

. Strapping rings tie every other pole of the radial slow-wave

structure, as in the case of a strapped radial magnetron, but the strapping rings are no
longer reentrant. Microwave energy to be amplified is fed to the strap at one end, and
the amplified output is extracted from the other end. This type of electron tube is
termed a strap-fed device.

If an M-type radial BWO is strapped, usually it does not start oscillation by itself.

But, if microwave energy is fed through the strap from the outside using an external mi-
crowave power source to the microwave input, then the oscillation starts--and even if

Figure 6.79 Functional schematic of the M-type radial BWO. (

From [13]. Used with per-

mission.)

© 1999 CRC Press LLC

the exciter source is turned off, the oscillation continues. This type of M-type radial
BWO is termed the platinotron [14].

In a platinotron, if the output of the tube is fed back to the input through a high-Q

cavity resonator, it becomes a self-starting oscillator. The oscillation frequency is stabi-
lized by the high-Q cavity resonator. This type of high-power frequency stabilized
strapped radial BWO is termed the stabilotron [14]. The operating powers are at kilo-
watt and megawatt levels.

Performance of the platinotron depends on, among other things, the design of the

slow-wave structure. For example, the interdigital slow-wave structure as shown in Fig-
ure 6.80 has a limited power handling capability and frequency bandwidth. Design of a
slow-wave structure with greater power handling capacity and stability, with broader
frequency bandwidth, is possible. For example, instead of an anode with an interdigital
slow-wave structure, the anode could be made of an annular open conducting duct,
loaded with a number of pairs of conducting posts across the open duct. Strapping is
done at every other tip of the pairs of conducting posts. This type of strapping loads the
slow-wave structure, stabilizing it and preventing oscillation. The structure of the an-
ode with an annular duct and pairs of posts increases the power handling capability.
This type of loaded radial BWO is termed the amplitron [14, 10]. The amplitron is capa-
ble of amplifying high-power microwave signals with pulses and continuous waves. It
is used for long-range pulsed radar transmitter amplifiers and industrial microwave
heating generators. The operating power levels range from kilowatt to megawatt levels.

Figure 6.80 Strap-fed devices: (

a) strapped radial magnetron, (b) nonreentrant strap-

ping of a BWO. (

From [13]. Used with permission.)

(

a)

(

b)

© 1999 CRC Press LLC

6.7.4 Gyrotron

The gyrotron is a cyclotron resonance maser.

4

The device includes a cathode, collec-

tor, and circular waveguide of gradually varying diameter. Electrons are emitted at the
cathode with small variations in speed. The electrons then are accelerated by an elec-
tric field and guided by a static magnetic field through the device. The nonuniform
induction field causes the rotational speed of the electrons to increase. The linear ve-
locity of the electrons, as a result, decreases. The interaction of the microwave field
within the waveguide and the rotating (helical) electrons causes bunching similar to
the bunching within a klystron. A decompression zone at the end of the device per-
mits decompression and collection of the electrons.

The power available from a gyrotron is 100 times greater than that possible from a

classic microwave tube at the same frequency. Power output of 100 kW is possible at
100 GHz, with 30 percent efficiency. At 300 GHz, up to 1.5 kW may be realized, but
with only 6 percent efficiency.

Theory of Operation

The trajectory of an electron in an electron beam focused by a longitudinally applied
magnetic field is a helix [13]. If the electron velocity, electron injection angle, and ap-
plied longitudinal magnetic flux density are varied, then an electron beam of helical
form with different size and pitch will be formed. A coil-shaped electron beam will be
produced by adjusting the acceleration voltage, applied magnetic flux density, and the
electron injection angle to the focusing magnetic field. The coil of the electron beam
can be a simple single coil, or--depending on the adjustment of the aforementioned
three parameters--it can be an electron beam of a double coil, or a large coil made of
thin small coils. In the case of the double-coil trajectory, the large coil-shaped trajec-
tory is termed the major orbit and smaller coil trajectory is termed the minor orbit.

If a single coil-shaped electron beam is launched in a waveguide, as shown in

Figure

6.81

, then microwaves in the waveguide will interact with the helical beam. This type of

vacuum tube is termed the helical beam tube [14]. In this class of device, a single-coil
helical beam is launched into a TE

10

mode rectangular waveguide. Inside the wave-

guide, microwaves travel from right to left and the helical beam travels in an opposite
direction. Therefore, the microwave-electron beam interaction is of the backward wave
type. If the microwave frequency, the focusing magnetic flux density B, and the acceler-
ation voltage V

a

are properly adjusted, this device will function as a backward wave am-

plifier. Electrons in the helical beam interact with the transverse microwave electric
fields and are velocity modulated at the left-hand side of the waveguide as the beam en-
ters into the waveguide. The velocity modulated electrons in the helical beam are
bunched as they travel toward the right. If the alternating microwave electric field syn-

4

Maser is an acronym for microwave amplification by simulated emission of radiation. Maser
is a general class of microwave amplifiers based on molecular interaction with
electromagnetic radiation.

© 1999 CRC Press LLC

chronizes its period and phase with the helical motion of bunched electrons so that the
electrons always receive retardation from microwave transverse electric fields, then the
electrons lose their kinetic energy and the microwave signal gains in electric field en-
ergy according to the principle of kinetic energy conservation. Thus, the amplified mi-
crowave power emerges at the waveguide output at the left (because the microwaves
travel backward).

In

Figure 6.81

, if the microwave input port and the output port are interchanged with

each other, then the system becomes a forward wave amplifier. Such a forward wave
amplifier is termed a peniotron [14].

If the electron gun is modified to incorporate a side emitting cathode and the wave-

guide is changed to TE

11

mode oversized circular waveguide, as shown in

Figure 6.82a

,

the gyrotron is formed. In this device, both ends of the waveguide are open and there are
sufficient reflections in the waveguide for positive feedback. The gyrotron is, thus, a
forward wave oscillator.

A double-coil helical beam gyrotron is shown in

Figure 6.82b

. The device is formed

by readjusting the anode voltage and the focusing flux density so that the electron beam
is made into a double helical coil (as shown in the figure), and operating the oversized
circular waveguide in the TE

01

mode. In the TE

01

mode, the microwave transverse elec-

tric fields exist as concentric circles. Therefore, the tangential electric fields interact
with electrons in the small coil trajectory. The alternating tangential microwave electric
fields are made to synchronize with the tangential motion of electrons in the minor
coil-shaped trajectory. Thus, electron velocity modulation takes place near the cathode
and bunching takes place in the middle of the tube. Microwave kinetic energy transfer
takes place as the beam approaches the right. The focusing magnetic flux density B is
applied only in the interaction region. Therefore, if the electron beam comes out of the
interaction region, it is defocused and collected by the anode waveguide (as depicted in

Figure 6.82b

). If the circular waveguide is operated in an oversized TE

11

mode, with the

double-coil helical beam, then the device is referred to as a tornadotron [14]. Micro-
wave-electron interaction occurs between the parallel component of tangential motion
of the small helical trajectory and the TE

11

mode microwave electric field. If the phase

of microwave electric field decelerates bunched electrons, then the lost kinetic energy
of the bunched electrons is transferred to the microwave signal and oscillation begins.

Figure 6.81 Basic structure of a
helical beam tube. (

From [13].

Used with permission.)

© 1999 CRC Press LLC

Gyrotron Design Variations

The gyrotron exists in a number of design variations, each optimized for a particular
feature or application [13, 15, 16].

When the gyrotron circular waveguide is split as shown in

Figure 6.83

, the tube is

termed the gyroklystron amplifier [13]. Both waveguides resonate to the input fre-
quency, and there are strong standing waves in both waveguide resonators. The input
microwave signal to be amplified is fed through a side opening to the input waveguide
resonator. This is the buncher resonator, which functions in a manner similar to the kly-
stron. The buncher resonator imparts velocity modulation to gyrating electrons in the
double helical coil-shaped electron beam. There is a drift space between the buncher
resonator and the catcher resonator at the output. While drifting electrons bunch and
bunched electrons enter into the output waveguide catcher resonator, electron speed is
adjusted in such a manner that electrons are decelerated by the resonating microwave
electric field. This lost kinetic energy in bunched electrons is transformed into micro-
wave energy and microwaves in the catcher resonator are, thus, amplified. The ampli-
fied power appears at the output of the tube.

If the gyrotron waveguide is an unsplit one-piece waveguide that is imped-

ance-matched and not resonating, as shown in

Figure 6.84

, the tube is termed the

gyrotron traveling wave tube amplifier [13]. In this tube, the input microwaves are fed
through an opening in the waveguide near the electron gun. Microwaves in the wave-

(

a)

(

b)

Figure 6.82 Functional schematic diagram of the gyrotron: (

a) single coiled helical

beam gyrotron, (

b) electron trajectory of double-coil helical beam gyrotron. (From [13].

Used with permission.)

© 1999 CRC Press LLC

guide are amplified gradually as they travel toward the output port by interacting with
the double-coiled helical electron beam, which is velocity-modulated and bunched.
There are no significant standing waves in the waveguide. Microwaves grow gradually
in the waveguide as they travel toward the output port as a result of interaction with elec-
trons.

If the electron gun of the gyrotron is moved to the side of the waveguide and micro-

wave power is extracted from the waveguide opening in proximity to the electron gun,
as shown in

Figure 6.85

, then the device is termed a gyrotron backward oscillator [13].

The principle involved is similar to the backward wave oscillator, and the process of ve-
locity modulation, drifting, bunching, and catching is similar to that of the klystron.
Microwave energy induced in the waveguide travels in both directions, but the circuit is
adjusted to emphasize the waves traveling in a backward direction. The backward
waves become the output of the tube and, at the same time, carry the positive feedback
energy to the electrons just emitted and to be velocity-modulated. The system, thus,
goes into oscillation.

Figure 6.83 Basic structure of the gyroklystron amplifier. (

From [13]. Used with permis-

sion.)

Figure 6.84 Basic structure of the gyroklystron traveling wave tube amplifier. (

From

[13].

Used with permission.)

© 1999 CRC Press LLC

If the gyrotron waveguide is split into two again, but this time the input side wave-

guide is short and the output side waveguide is long, as shown in

Figure 6.86

, then the

tube is termed a gyrotwystron amplifier [13]. This device is a combination of the
gyroklystron and gyrotron traveling wave tube amplifier, thus the name gyrotwystron
amplifier. The input side waveguide resonator is the same as the input resonator wave-
guide of a gyroklystron. There are strong standing waves in the input bunched-wave-
guide resonator. There is no drift space between the two waveguides. The output side
waveguide is a long impedance-matched waveguide and there is no microwave stand-
ing wave in the waveguide (a traveling-wave waveguide). As microwaves travel in the
waveguide, they interact with bunched electrons and the microwaves grow as they
move toward the output port.

6.8

Other Microwave Devices

There are a number of variations on the basic microwave devices outlined previously
in this chapter. Some find widespread usage, others have little commercial interest.

Figure 6.85 Basic structure of the gyrotron backward oscillator. (

From [13]. Used with

permission.)

Figure 6.86 Basic structure of the gyrotwystron amplifier. (

From [13]. Used with permis-

sion.)

© 1999 CRC Press LLC

Among the important specialty device classifications are quasiquantum devices and
modified klystrons.

6.8.1 Quasiquantum Devices

In a gyrotron, if electrons are accelerated by extremely high voltage, they become rel-
ativistic [13]. That is, the mass of an electron becomes a function of the velocity as,

( )

m

m

c

v

=

-

0

2

1

(6.3)

Where:
m = the relativistic mass of an electron
m

0

= the static mass of an electron

c = the speed of light in vacuum
v = the speed of the electron in question

The energy states of electrons are specified by a set of quantum numbers, and the
transition of energy states occurs only between those numbers so described. A set of
quantum numbers specifies the high-energy orbit and low-energy orbit of an electron.

One device based on this principle is the free electron laser, or the ubitron [13]. A

schematic diagram of the device is given in

Figure 6.87

. A high-speed relativistic elec-

tron beam is emitted from an electron gun focused by a longitudinally applied dc mag-
netic flux density B and is periodically deflected by magnetic means. The repetitious
deflections create among relativistic electrons high-energy states and low-energy
states. If the high-energy electrons in the deflection waveguides are stimulated by the
resonance frequency of the waveguide resonator, then the downward transition (the
transition of electrons in a high-energy state to a low-energy state) occurs. Microwave
emissions result because of the energy transitions at the stimulation frequency.

The ubitron operates at millimeter or submillimeter wave frequencies, and typically

at high power levels.

6.8.2 Variations on the Klystron

Because the multiple cavities of a conventional klystron are difficult to fabricate at
millimeter wavelengths, variations on the basis design have been produced [13]. In-
stead of using multiple cavity resonators, the cavities can instead be combined into a
single cavity. Such a design can be accomplished through the use of ladder-shaped
grids, as shown in

Figure 6.88a

by the dashed

lines with the single cavity. This type of

klystron is referred to as the laddertron [14]. Because of the ladder structure, a stand-
ing wave will exist on on the ladder line. Velocity modulation results near the electron
gun and bunching begins near the middle of the electron stream. Catching is done in
the ladder structure near the collector. Because this device is a one-cavity resonator,

© 1999 CRC Press LLC

there is built-in positive feedback. Oscillation, therefore, begins in a manner similar
to a reflex klystron, also a single cavity tube.

At millimeter wavelengths, a dominant mode cavity resonator becomes very small

and loses its power-handling capability. In a laddertron, however, the cavity resonator is
a multiple higher-mode resonator. Therefore, the Q of the cavity is high and the size of
the cavity resonator is reasonably large. This being the case, the laddertron can handle
tens of watts of power at frequency as high as 50 GHz.

The catcher of a two-cavity klystron amplifier can be replaced by the slow-wave

structure of a traveling wave tube, as shown in

Figure 6.88b

. This type of tube is termed

the twystron [14]. The slow-wave structure provides a broader frequency bandwidth
than a regular two-cavity klystron. A microwave input signal fed into the buncher cavity
results in velocity modulation of the electron beam. Bunching occurs while the beam
drifts, and the bunched electrons induce a microwave voltage in the slow-wave struc-
ture. The electron beam is focused by the use of longitudinally applied magnetic flux
density B.

Figure 6.87 Basic structure of the ubitron. (

From [13]. Used with permission.)

Figure 6.88 Functional schematic diagrams of modified klystrons: (

a) laddertron, (b)

twystron. (

From [13]. Used with permission.)

(

a)

(

b)

© 1999 CRC Press LLC

Commercial twystrons typically operate at 26 GHz and are used in pulsed radar

transmitter power amplifiers. Pulsed peak output powers range from 1 to 7 MW (pulse
width of 1050

µs) [10].

6.9

Microwave Tube Life

Any analysis of microwave tube life must first identify the parameters that define life.
The primary wear-out mechanism in a microwave power tube is the electron gun at
the cathode. In principle, the cathode eventually will evaporate the activating material
and cease to produce the required output power. Tubes, however, rarely fail because of
low emission, but for a variety of other reasons that are usually external to the device.

Power tubes designed for microwave applications provide long life when operated

within their designed parameters. The point at which the device fails to produce the re-
quired output power can be predicted with some accuracy, based on design data and
in-service experience. Most power tubes, however, fail because of mechanisms other
than predictable chemical reactions inside the device itself. External forces, such as
transient overvoltages caused by lightning, cooling system faults, and improper tuning,
more often than not lead to the failure of a microwave tube.

6.9.1 Life-Support System

Transmitter control logic usually is configured for two states of operation:

·

An operational level, which requires all of the "life-support" systems to be pres-
ent before the high-voltage (HV) command is enabled.

·

An overload level, which removes HV when one or more fault conditions occur.

The cooling system is the primary life-support element in most RF generators. The

cooling system should be fully operational before the application of voltages to the
tube. Likewise, a cool-down period usually is recommended between the removal of
beam and filament voltages and shutdown of the cooling system.

Most microwave power tubes require a high-voltage removal time of less than 100

ms after the occurrence of an overload. If the trip time is longer, damage to the device
may result. Arc detectors often are installed in the cavities of high-power tubes to sense
fault conditions and shut down the high-voltage power supply before damage can be
done to the tube. Depending on the circuit parameters, arcs can be sustaining, requiring
removal of high voltage to squelch the arc. A number of factors can cause RF arcing, in-
cluding:

·

Overdrive condition

·

Mistuning of one or more cavities

·

Poor cavity fit (applies to external types only)

·

Undercoupling of the output to the load

·

Lightning strike at the antenna

© 1999 CRC Press LLC

·

High VSWR

Regardless of the cause, arcing can destroy internal elements or the vacuum seal if
drive and/or high voltage are not removed quickly. A lamp usually is included with
each arc detector photocell for test purposes.

6.9.2 Protection Measures

A microwave power tube must be protected by control devices in the amplifier sys-
tem. Such devices offer either visual indications or aural alarm warnings, or they ac-
tuate interlocks within the system.

Figure 6.89

shows a klystron amplifier and the ba-

sic components associated with its operation, including metering for each of the
power supplies. Other types of microwave power devices use similar protection
schemes. Sections of coaxial transmission line, representing essential components,
are shown in the figure attached to the RF input and RF output ports of the tube. A
single magnetic coil is shown to represent any coil configuration that may exist; its
position in the drawing is for convenience only and does not represent its true position
in the system.

Heater Supply

The heater power supply can be either ac or dc. If it is dc, the positive terminal must
be connected to the common heater-cathode terminal and the negative terminal to the
second heater terminal. The amount of power supplied to the heater is important be-
cause it establishes the cathode operating temperature. The temperature must be high
enough to provide ample electron emission, but not so high that emission life is jeop-
ardized.

Because the cathode and heater are connected to the negative side of the beam sup-

ply, they must be insulated to withstand the full beam potential. This dictates the use of
a filament transformer designed specifically for the specific application.

Beam Supply

The high-voltage beam supply furnishes the dc input power to the microwave device.
In a klystron, the positive side of the beam supply is connected to the body and collec-
tor. The negative terminal is connected to the common heater-cathode terminal. Never
connect the negative terminal of the beam supply to the heater-only terminal because
the beam current will then flow through the heater to the cathode and cause premature
heater failure. The voltmeter, E

b

in

Figure 6.89

, measures the beam voltage applied

between the cathode and the body of the klystron.

Current meter I

c

measures collector current, typically 95 percent or more of the total

device current. Current meter I

by

measures body current. An interlock should interrupt

the beam supply if the body current exceeds a specified maximum value.

The sum of the body current (I

by

) and collector current (I

c

) is equal to the beam cur-

rent (I

b

), which should stay constant as long as the beam voltage and modulating-anode

voltage are held constant.

© 1999 CRC Press LLC

Magnet Supply

Electrical connections to the dc magnet supply typically include two meters, one for
measuring current through the circuit (I

m

in

Figure 6.89

), and one for measuring volt-

age (E

m

). When a microwave device is installed in its magnet assembly, both parame-

ters should be measured and recorded for future reference. If excessive body current
or other unusual symptoms occur, this data will be valuable for system analysis.

Undercurrent protection should be provided to remove beam voltage if the magnetic

circuit current falls below a preset value. The interlock also should prevent the beam
voltage from being applied if the magnetic circuit is not energized. This scheme, how-
ever, will not provide protection if windings of the magnet are short-circuited.
Short-circuited conditions can be determined by measuring the normal values of volt-
age and current and recording them for future reference.

The body-current overload protection should actuate if the magnetic field is reduced

for any reason.

Figure 6.89 Protection and metering system for a klystron amplifier.

© 1999 CRC Press LLC

RF Circuits

In

Figure 6.89

, monitoring devices are shown on the RF input and output of the kly-

stron. These monitors protect the device in case of a failure in the RF output circuit.
Two directional couplers and a photodetector are attached to the output of the device.
These components and an RF switching device on the input form a protective network
against output transmission line mismatch. The RF switch is activated by the
photodetector or the reflected power monitor and must be capable of removing RF
drive power from the tube in less than 10 ms (typically).

In the RF output circuit, the forward power coupler is used to measure the relative

power output of the microwave device. The reflected power coupler measures the RF
energy reflected by the output circuit components or antenna. Damaged components or
foreign material in the RF line will increase the RF reflected power. The amount of re-
flected power should be no more than 5 percent of the actual forward RF output power
of the tube in most applications. An interlock monitors the reflected power and removes
RF drive to the device if the reflected energy reaches an unsafe level. To protect against
arcs occurring between the monitor and the output window, a photodetector is placed
between the monitor and the window. Light from an arc will trigger the photodetector,
which actuates the interlock system to remove RF drive before the window is damaged.

6.9.3 Filament Voltage Control

Extending the life of a microwave tube begins with accurate adjustment of filament
voltage, as noted in the previous section. The filament should not be operated at a re-
duced voltage in an effort to extend tube life, as in the case of the thoriated tungsten
grid tube. In a microwave tube, reduced filament voltage may cause uneven emission
from the surface of the cathode with little or no improvement in cathode life.

Voltage should be applied to the filament for a specified warm-up period before the

application of beam current to minimize thermal stress on the cathode/gun structure.
However, voltage should not be applied to the filaments for extended periods (typically
2 hours or more) if no beam voltage is present. The net rate of evaporation of emissive
material from the cathode surface is greater without beam voltage. Subsequent conden-
sation of material on gun components may lead to voltage standoff problems.

6.9.4 Cooling System

The cooling system is vital to any RF generator. In a high-power microwave transmit-
ter, the cooling system may need to dissipate as much as 70 percent of the input ac
power in the form of waste heat. For vapor-phase-cooled devices, pure (distilled or de-
mineralized) water must be used. Because the collector is usually only several volts
above ground potential, it is generally not necessary to use deionized water. (Cooling
considerations for microwave power tubes are discussed in Chapter 8. Cooling system
maintenance is examined in Chapter 9.)

© 1999 CRC Press LLC

6.9.5 Reliability Statistics

Determination of the mean time between failure (MTBF) of a microwave tube pro-
vides valuable information on the operation of a given device in a given installation.
The following formulas can be used to predict tube life in operating years, and to esti-
mate the number of replacement tubes that will be needed during the life of an RF
generator.

Y

MTBF

H

=

(6.4)

N

L

S

Y

= ×

(6.5)

R

N

S

= -

(6.6)

Where:
Y = Tube life in operating years
MTBF = Tube mean time between failure (gathered from manufacturer literature or

on-site experience)

H = Hours of operation per year (= 365 hours per operating day)
N = Number of tubes needed over the life of the transmitter
L = Anticipated life of the transmitter in years
S = Number of tubes per transmitter
R = Number of replacement tubes needed over the life of the transmitter

MTTR

The mean time to repair (MTTR) of a given piece of RF equipment is an important
consideration for any type of facility. MTTR defines the maintainability of a system.
In the case of a microwave generator, the time required to change a tube is an impor-
tant factor, especially if standby equipment is not available. The example time-change
estimates given previously in this chapter (1 hour for an integral-cavity klystron and 4
to 6 hours for a 60 kW UHF external-cavity tube) assume that no preparation work
has been performed on the spare device. Pretuning a tube is one way to shorten the
MTTR for an external-cavity device. The spare tube is installed during a maintenance
period, and the system is tuned for proper operation. After the positions of all tuning
controls are documented, the tube is removed and returned to its storage container,
along with the list of tuning control settings. In this way, the external-cavity tube can
be placed into service much faster during an emergency. This procedure probably will
not result in a tube tuned for optimum performance, but it may provide a level of per-
formance that is acceptable on a temporary basis.

Consideration of MTTR is important for a facility because most microwave tube

failure is caused by mechanisms other than reduced cathode emission, which is a soft

© 1999 CRC Press LLC

failure that can be anticipated with some degree of accuracy. Catastrophic failures, on
the other hand, offer little, if any, warning.

Reliability engineering is discussed in greater detail in Section 9.3.

6.10 References

1. Ostroff, Nat S., "A Unique Solution to the Design of an ATV Transmitter," Proceed-

ings of the 1996 NAB Broadcast Engineering Conference, National Association of
Broadcasters, Washington, DC, pg. 144, 1996.

2. Hulick, Timothy P., "60 kW Diacrode UHF TV Transmitter Design, Performance

and Field Report," Proceedings of the 1996 NAB Broadcast Engineering Confer-
ence, National Association of Broadcasters, Washington, DC, pg. 442, 1996.

3. "Integral Cavity Klystrons for UHF-TV Transmitters," Varian Associates, Palo

Alto, CA.

4. Clayworth, G. T., H. P. Bohlen, and R. Heppinstall, "Some Exciting Adventures in

the IOT Business," NAB 1992 Broadcast Engineering Conference Proceedings,
National Association of Broadcasters, Washington, DC, pp. 200 - 208, 1992.

5. Symons, Robert S., "The Constant Efficiency Amplifier," Proceedings of the NAB

Broadcast Engineering Conference, National Association of Broadcasters, Wash-
ington, D.C., pp. 523530, 1997.

6. Priest, D. H., and M.B. Shrader, "The Klystrode--An Unusual Transmitting Tube

with Potential for UHF-TV," Proc. IEEE, Vol. 70, No. 11, pp. 13181325, Novem-
ber 1982.

7. Gilmore, A. S., Microwave Tubes, Artech House, Dedham, Mass., pp. 196200,

1986.

8. Symons, R., M. Boyle, J. Cipolla, H. Schult, and R. True, "The Constant Efficiency

Amplifier--A Progress Report," Proceedings of the NAB Broadcast Engineering
Conference, National Association of Broadcasters, Washington, D.C., pp. 7784,
1998.

9. Ishii, T. K., "Traveling Wave Tubes," in The Electronics Handbook, Jerry C.

Whitaker (ed.), CRC Press, Boca Raton, Fla., pp. 428443, 1996.

10. Liao, S. Y., Microwave Electron Tube Devices, Prentice-Hall, Englewood Cliffs,

N.J., 1988.

11. Sims, G. D., and I. M Stephenson, Microwave Tubes and Semiconductor Devices,

Interscience, London, 1963.

12. Hutter, R. G. E., Beam and Wave Electronics in Microwave Tubes, Interscience,

London, 1960.

13. Ishiik, T. K., "Other Microwave Vacuum Devices," in The Electronics Handbook,

Jerry C. Whitaker (ed.), CRC Press, Boca Raton, Fla., pp. 444457, 1996.

14. Ishii, T. K., Microwave Engineering, Harcourt-Brace-Jovanovich, San Diego, CA,

1989.

15. Coleman, J. T., Microwave Devices, Reston Publishing, Reston, VA, 1982.
16. McCune, E. W., "Fision Plasma Heating with High-Power Microwave and Milli-

meter Wave Tubes," Journal of Microwave Power, Vol. 20, pp. 131136, April,
1985.

© 1999 CRC Press LLC

6.11 Bibliography

Badger, George, "The Klystrode: A New High-Efficiency UHF-TV Power Ampli-

fier," Proceedings of the NAB Engineering Conference, National Association of
Broadcasters, Washington, DC, 1986.

Collins, G. B., Radar System Engineering, McGraw-Hill, New York, 1947.
Crutchfield, E. B., (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, 1992.

Dick, Bradley, "New Developments in RF Technology," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, May 1986.

Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 3rd Ed.,

McGraw-Hill, New York, 1989.

Fisk, J. B., H. D. Hagstrum, and P. L. Hartman, "The Magnetron As a Generator of

Centimeter Waves," Bell System Tech J., Vol. 25, pg. 167, April 1946.

Ginzton, E. L., and A. E. Harrison, "Reflex Klystron Oscillators," Proc. IRE, Vol. 34,

pg. 97, March 1946.

IEEE Standard Dictionary of Electrical and Electronics Terms, Institute of Electrical

and Electronics Engineers, Inc., New York, 1984.

McCune, Earl, "Final Report: The Multi-Stage Depressed Collector Project," Pro-

ceedings of the NAB Engineering Conference, National Association of Broad-
casters, Washington, DC, 1988.

Ostroff, N., A. Kiesel, A. Whiteside, and A. See, "Klystrode-Equipped UHF-TV

Transmitters: Report on the Initial Full Service Station Installations," Proceedings
of the NAB Engineering Conference, National Association of Broadcasters, Wash-
ington, DC, 1989.

Ostroff, N., A. Whiteside, A. See, and A. Kiesel, "A 120 kW Klystrode Transmitter for

Full Broadcast Service," Proceedings of the NAB Engineering Conference, Na-
tional Association of Broadcasters, Washington, DC, 1988.

Ostroff, N., A. Whiteside, and L. Howard, "An Integrated Exciter/Pulser System for

Ultra High-Efficiency Klystron Operation," Proceedings of the NAB Engineering
Conference, National Association of Broadcasters, Washington, DC, 1985.

Pierce, J. R., "Reflex Oscillators," Proc. IRE, Vol. 33, pg. 112, February 1945.
Pierce, J. R., "Theory of the Beam-Type Traveling Wave Tube," Proc. IRE, Vol. 35,

pg. 111, February 1947.

Pierce, J. R., and L. M. Field, "Traveling-Wave Tubes," Proc. IRE, Vol. 35, pg. 108,

February 1947.

Pond, N. H., and C. G. Lob, "Fifty Years Ago Today or On Choosing a Microwave

Tube," Microwave Journal, pp. 226 - 238, September 1988.

Priest, D., and M. Shrader, "The Klystrode--An Unusual Transmitting Tube with Po-

tential for UHF-TV," Proceedings of the IEEE, Vol. 70, no. 11, IEEE, New York,
November 1982.

Shrader, Merrald B., "Klystrode Technology Update," Proceedings of the NAB Engi-

neering Conference, National Association of Broadcasters, Washington, DC, 1988.

Spangenberg, Karl, Vacuum Tubes, McGraw-Hill, New York, 1947.
Terman, F. E., Radio Engineering, 3rd Ed., McGraw-Hill, New York, 1947.
Varian, R., and S. Varian, "A High-Frequency Oscillator and Amplifier," J. Applied

Phys., Vol. 10, pg. 321, May 1939.

Webster, D. L., "Cathode Bunching," J. Applied Physics, Vol. 10, pg. 501, July 1939.

© 1999 CRC Press LLC

Whitaker, Jerry C., and T. Blankenship, "Comparing Integral and External Cavity Kly-

strons," Broadcast Engineering, Intertec Publishing, Overland Park, KS, Novem-
ber 1988.

Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1991.

Whitaker, Jerry C., (ed.), NAB Engineering Handbook, 9th Ed., National Association

of Broadcasters, Washington, DC, 1998.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

7 RF Interconnection and Switching"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

7

RF Interconnection and Switching

7.1

Introduction

The components that connect, interface, transfer, and filter RF energy within a given
system--or between systems--are critical elements in the operation of vacuum tube
devices. Such hardware, usually passive, determines to a large extent the overall per-
formance of the RF generator. To optimize the performance of power vacuum de-
vices, first it is necessary to understand and optimize the components upon which the
tube depends.

The mechanical and electrical characteristics of the transmission line, waveguide,

and associated hardware that carry power from a power source (usually a transmitter) to
the load (usually an antenna) are critical to proper operation of any RF system. Me-
chanical considerations determine the ability of the components to withstand tempera-
ture extremes, lightning, rain, and wind. In other words, they determine the overall reli-
ability of the system.

7.1.1 Skin Effect

The effective resistance offered by a given conductor to radio frequencies is consider-
ably higher than the ohmic resistance measured with direct current. This is because of
an action known as the skin effect, which causes the currents to be concentrated in cer-
tain parts of the conductor and leaves the remainder of the cross section to contribute
little or nothing toward carrying the applied current.

When a conductor carries an alternating current, a magnetic field is produced that

surrounds the wire. This field continually expands and contracts as the ac wave in-
creases from zero to its maximum positive value and back to zero, then through its neg-
ative half-cycle. The changing magnetic lines of force cutting the conductor induce a
voltage in the conductor in a direction that tends to retard the normal flow of current in
the wire. This effect is more pronounced at the center of the conductor. Thus, current
within the conductor tends to flow more easily toward the surface of the wire. The
higher the frequency, the greater the tendency for current to flow at the surface. The
depth of current flow is a function of frequency, and it is determined from the following
equation:

© 1999 CRC Press LLC

d

f

=

2 6

.

µ

(7.1)

Where:
d = depth of current in mils
µ = permeability (copper = 1, steel = 300)
f = frequency of signal in mHz

It can be calculated that at a frequency of 100 kHz, current flow penetrates a conduc-
tor by 8 mils. At 1 MHz, the skin effect causes current to travel in only the top 2.6 mils
in copper, and even less in almost all other conductors. Therefore, the series imped-
ance of conductors at high frequencies is significantly higher than at low frequencies.

Figure 7.1

shows the distribution of current in a radial conductor.

When a circuit is operating at high frequencies, the skin effect causes the current to

be redistributed over the conductor cross section in such a way as to make most of the
current flow where it is encircled by the smallest number of flux lines. This general
principle controls the distribution of current regardless of the shape of the conductor in-
volved. With a flat-strip conductor, the current flows primarily along the edges, where
it is surrounded by the smallest amount of flux.

7.2

Coaxial Transmission Line

Two types of coaxial transmission line are in common use today: rigid line and corru-
gated (semiflexible) line. Rigid coaxial cable is constructed of heavy-wall copper
tubes with Teflon or ceramic spacers. (Teflon is a registered trademark of DuPont.)

Figure 7.1

Skin effect on an isolated round conductor carrying a moderately high fre-

quency signal.

© 1999 CRC Press LLC

Rigid line provides electrical performance approaching an ideal transmission line, in-
cluding:

·

High power-handling capability

·

Low loss

·

Low VSWR (voltage standing wave ratio)

Rigid transmission line is, however, expensive to purchase and install.

The primary alternative to rigid coax is semiflexible transmission line made of cor-

rugated outer and inner conductor tubes with a spiral polyethylene (or Teflon) insulator.
The internal construction of a semiflexible line is shown in

Figure 7.2

. Semiflexible

line has four primary benefits:

·

It is manufactured in a continuous length, rather than the 20-ft sections typically
used for rigid line.

·

Because of the corrugated construction, the line may be shaped as required for
routing from the transmitter to the antenna.

·

The corrugated construction permits differential expansion of the outer and inner
conductors.

·

Each size of line has a minimum bending radius. For most installations, the flexi-
ble nature of corrugated line permits the use of a single piece of cable from the
transmitter to the antenna, with no elbows or other transition elements. This
speeds installation and provides for a more reliable system.

7.2.1 Electrical Parameters

A signal traveling in free space is unimpeded; it has a free-space velocity equal to the
speed of light. In a transmission line, capacitance and inductance slow the signal as it
propagates along the line. The degree to which the signal is slowed is represented as a
percentage of the free-space velocity. This quantity is called the relative velocity of
propagation and is described by the equation:

V

L

C

p

=

×

1

(7.2)

Where:
L = inductance in henrys per foot
C = capacitance in farads per foot

and

V

V

c

r

p

=

×100%

(7.3)

© 1999 CRC Press LLC

Figure 7.2

Semiflexible coaxial cable: (

a) a section of cable showing the basic con-

struction, (

b) cable with various terminations. (Courtesy of Andrew.)

(

b)

(

a)

© 1999 CRC Press LLC

Where:
V

p

= velocity of propagation

c = 9.842

× 10

8

feet per second (free-space velocity)

V

r

= velocity of propagation as a percentage of free-space velocity

Transverse Electromagnetic Mode

The principal mode of propagation in a coaxial line is the transverse electromagnetic
mode (TEM). This mode will not propagate in a waveguide, and that is why coaxial
lines can propagate a broad band of frequencies efficiently. The cutoff frequency for a
coaxial transmission line is determined by the line dimensions. Above cutoff, modes
other than TEM can exist and the transmission properties are no longer defined. The
cutoff frequency is equivalent to:

F

V

D

D

c

r

i

o

=

×

+

7 50

.

(7.4)

Where:
F

c

= cutoff frequency in gigahertz

V

r

= velocity (percent)

D

i

= inner diameter of outer conductor in inches

D

o

= outer diameter of inner conductor in inches

At dc, current in a conductor flows with uniform density over the cross section of the

conductor. At high frequencies, the current is displaced to the conductor surface. The
effective cross section of the conductor decreases, and the conductor resistance in-
creases because of the skin effect.

Center conductors are made from copper-clad aluminum or high-purity copper and

can be solid, hollow tubular, or corrugated tubular. Solid center conductors are found
on semiflexible cable with 1/2 in or smaller diameter. Tubular conductors are found in
7/8 in or larger-diameter cables. Although the tubular center conductor is used primar-
ily to maintain flexibility, it also can be used to pressurize an antenna through the
feeder.

Dielectric

Coaxial lines use two types of dielectric construction to isolate the inner conductor
from the outer conductor. The first is an air dielectric, with the inner conductor sup-
ported by a dielectric spacer and the remaining volume filled with air or nitrogen gas.
The spacer, which may be constructed of spiral or discrete rings, typically is made of
Teflon or polyethylene. Air-dielectric cable offers lower attenuation and higher aver-
age power ratings than foam-filled cable but requires pressurization to prevent mois-
ture entry.

Foam-dielectric cables are ideal for use as feeders with antennas that do not require

pressurization. The center conductor is surrounded completely by foam-dielectric ma-

© 1999 CRC Press LLC

terial, resulting in a high dielectric breakdown level. The dielectric materials are poly-
ethylene-based formulations, which contain antioxidants to reduce dielectric deteriora-
tion at high temperatures.

Impedance

The expression transmission line impedance applied to a point on a transmission line
signifies the vector ratio of line voltage to line current at that particular point. This is
the impedance that would be obtained if the transmission line were cut at the point in
question, and the impedance looking toward the receiver were measured.

Because the voltage and current distribution on a line are such that the current tends

to be small when the voltage is large (and vice versa), as shown in

Figure 7.3

, the imped-

ance will, in general, be oscillatory in the same manner as the voltage (large when the
voltage is high and small when the voltage is low). Thus, in the case of a short-circuited
receiver, the impedance will be high at distances from the receiving end that are odd
multiples of 1/4 wavelength, and it will be low at distances that are even multiples of 1/4
wavelength.

The extent to which the impedance fluctuates with distance depends on the standing

wave ratio (ratio of reflected to incident waves), being less as the reflected wave is pro-
portionally smaller than the incident wave. In the particular case where the load imped-
ance equals the characteristic impedance, the impedance of the transmission line is
equal to the characteristic impedance at all points along the line.

The power factor of the impedance of a transmission line varies according to the

standing waves present. When the load impedance equals the characteristic impedance,
there is no reflected wave, and the power factor of the impedance is equal to the power
factor of the characteristic impedance. At radio frequencies, the power factor under
these conditions is accordingly resistive. However, when a reflected wave is present, the
power factor is unity (resistive) only at the points on the line where the voltage passes
through a maximum or a minimum. At other points the power factor will be reactive, al-
ternating from leading to lagging at intervals of 1/4 wavelength. When the line is
short-circuited at the receiver, or when it has a resistive load less than the characteristic
impedance so that the voltage distribution is of the short-circuit type, the power factor is
inductive for lengths corresponding to less than the distance to the first voltage maxi-
mum. Thereafter, it alternates between capacitive and inductive at intervals of 1/4
wavelength. Similarly, with an open-circuited receiver or with a resistive load greater
than the characteristic impedance so that the voltage distribution is of the open-circuit
type, the power factor is capacitive for lengths corresponding to less than the distance to
the first voltage minimum. Thereafter, the power factor alternates between capacitive
and inductive at intervals of 1/4 wavelength, as in the short-circuited case.

Resonant Characteristics

A transmission line can be used to perform the functions of a resonant circuit. If the
line, for example, is short-circuited at the receiver, at frequencies in the vicinity of a
frequency at which the line is an odd number of 1/4 wavelengths long, the impedance

© 1999 CRC Press LLC

will be high and will vary with frequency in the vicinity of resonance. This character-
istic is similar in nature to a conventional parallel resonant circuit. The difference is
that with the transmission line, there are a number of resonant frequencies, one for
each of the infinite number of frequencies that make the line an odd number of 1/4
wavelengths long. At VHF, the parallel impedance at resonance and the circuit Q ob-
tainable are far higher than can be realized with lumped circuits. Behavior corre-
sponding to that of a series resonant circuit can be obtained from a transmission line
that is an odd number of 1/4 wavelengths long and open-circuited at the receiver.

Transmission lines also can be used to provide low-loss inductances or capacitances

if the proper combination of length, frequency, and termination is employed. Thus, a
line short-circuited at the receiver will offer an inductive reactance when less than 1/4
wavelength, and a capacitive reactance when between 1/4 and 1/2 wavelength. With an
open-circuited receiver, the conditions for inductive and capacitive reactances are re-
versed.

Figure 7.3

Magnitude and power factor of line impedance with increasing distance from

the load for the case of a short-circuited receiver and a line with moderate attenuation:
(

a) voltage distribution, (b) impedance magnitude, (c) impedance phase.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

7.2.2 Electrical Considerations

VSWR, attenuation, and power-handling capability are key electrical factors in the
application of coaxial cable. High VSWR can cause power loss, voltage breakdown,
and thermal degradation of the line. High attenuation means less power delivered to
the antenna, higher power consumption at the transmitter, and increased heating of the
transmission line itself.

VSWR is a common measure of the quality of a coaxial cable. High VSWR indi-

cates nonuniformities in the cable that can be caused by one or more of the following
conditions:

·

Variations in the dielectric core diameter

·

Variations in the outer conductor

·

Poor concentricity of the inner conductor

·

Nonhomogeneous or periodic dielectric core

Although each of these conditions may contribute only a small reflection, they can
add up to a measurable VSWR at a particular frequency.

Rigid transmission line typically is available in a standard length of 20 ft, and in al-

ternative lengths of 19.5 ft and 19.75 ft. The shorter lines are used to avoid VSWR
buildup caused by discontinuities resulting from the physical spacing between line sec-
tion joints. If the section length selected and the operating frequency have a 1/2-wave
correlation, the connector junction discontinuities will add. This effect is known as
flange buildup. The result can be excessive VSWR. The critical frequency at which a
1/2-wave relationship exists is given by:

F

n

L

cr

=

×

490 4

.

(7.5)

Where:
F

cr

= the critical frequency

n = any integer
L = transmission line length in feet

For most applications, the critical frequency for a chosen line length should not fall
closer than ±2 MHz of the passband at the operating frequency.

Attenuation is related to the construction of the cable itself and varies with fre-

quency, product dimensions, and dielectric constant. Larger-diameter cable exhibits
lower attenuation than smaller-diameter cable of similar construction when operated at
the same frequency. It follows, therefore, that larger-diameter cables should be used for
long runs.

Air-dielectric coax exhibits less attenuation than comparable-size foam-dielectric

cable. The attenuation characteristic of a given cable also is affected by standing waves
present on the line resulting from an impedance mismatch.

Table 7.1

shows a represen-

tative sampling of semiflexible coaxial cable specifications for a variety of line sizes.

© 1999 CRC Press LLC

7.2.3 Coaxial Cable Ratings

Selection of a type and size of transmission line is determined by a number of param-
eters, including power-handling capability, attenuation, and phase stability.

Power Rating

Both peak and average power ratings are required to fully describe the capabilities of
a transmission line. In most applications, the peak power rating limits the low fre-
quency or pulse energy, and the average power rating limits high-frequency applica-
tions, as shown in

Figure 7.4

. Peak power ratings usually are stated for the following

conditions:

·

VSWR = 1.0

·

Zero modulation

·

One atmosphere of absolute dry air pressure at sea level

The peak power rating of a selected cable must be greater than the following expres-

sion, in addition to satisfying the average-power-handling criteria:

E

P

M

VSWR

pk

t

>

× +

×

(

)

1

2

(7.6)

Where:
P

pk

= cable peak power rating in kilowatts

P

t

= transmitter power in kilowatts

M = amplitude modulation percentage expressed decimally (100 percent = 1.0)
VSWR = voltage standing wave ratio

From this equation, it can be seen that 100 percent amplitude modulation will in-
crease the peak power in the transmission line by a factor of 4. Furthermore, the peak
power in the transmission line increases directly with VSWR.

The peak power rating is limited by the voltage breakdown potential between the in-

ner and outer conductors of the line. The breakdown point is independent of frequency.

Table 7.1

Representative Specifications for Various Types of Flexible Air-Dielectric Co-

axial Cable

Cable

Size

(in.)

Maximum

Frequency

(MHz)

Velocity

(percent)

Peak

Power 1

MHz (kW)

Average Power

Attenuation

1

100 MHz

(kW)

1 MHz

(kW)

100 MHz

(dB)

1 Mhz

(dB)

1

5

8

2.7

92.1

145

145

14.4

0.020

0.207

3

1.64

93.3

320

320

37

0.013

0.14

4

1.22

92

490

490

56

0.010

0.113

5

0.96

93.1

765

765

73

0.007

0.079

1

Attenuation specified in dB/100 ft.

© 1999 CRC Press LLC

Figure 7.4

Power rating data for a variety of coaxial transmission lines: (

a) 50

line, (b)

75

line.

(

a)

(

b)

© 1999 CRC Press LLC

It varies, however, with the line pressure (for an air-dielectric cable) and the type of
pressurizing gas.

The average power rating of a transmission line is limited by the safe long-term op-

erating temperature of the inner conductor and the dielectric. Excessive temperatures
on the inner conductor will cause the dielectric material to soften, leading to mechani-
cal instability inside the line.

The primary purpose of pressurization of an air-dielectric cable is to prevent the in-

gress of moisture. Moisture, if allowed to accumulate in the line, can increase attenua-
tion and reduce the breakdown voltage between the inner and outer conductors. Pres-
surization with high-density gases can be used to increase both the average power and
the peak power ratings of a transmission line. For a given line pressure, the increased
power rating is more significant for peak power than for average power. High-density
gases used for such applications include Freon 116 and sulfur hexafluoride.

Figure 7.5

illustrates the effects of pressurization on cable power rating.

An adequate safety factor is necessary for peak and average power ratings. Most

transmission lines are tested at two or more times their rated peak power before ship-
ment to the customer. This safety factor is intended as a provision for transmitter tran-
sients, lightning-induced effects, and high-voltage excursions resulting from unfore-
seen operating conditions.

Connector Effects

Foam-dielectric cables typically have a greater dielectric strength than air-dielectric
cables of similar size. For this reason, foam cables would be expected to exhibit
higher peak power ratings than air lines. Higher values, however, usually cannot be re-
alized in practice because the connectors commonly used for foam cables have air
spaces at the cable/connector interface that limit the allowable RF voltage to "air ca-
ble" values.

The peak-power-handling capability of a transmission line is the smaller of the val-

ues for the cable and the connectors attached to it.

Table 7.2

lists the peak power ratings

of several common RF connectors at standard conditions (defined in the previous sec-
tion).

Attenuation

The attenuation characteristics of a transmission line vary as a function of the operat-
ing frequency and the size of the line itself. The relationships are shown in

Figure 7.6

.

The efficiency of a transmission line dictates how much power output by the trans-

mitter actually reaches the antenna. Efficiency is determined by the length of the line
and the attenuation per unit length.

The attenuation of a coaxial transmission line is defined by the equation:

=

×

10

1

2

log

P

P

(7.7)

© 1999 CRC Press LLC

Figure 7.5

Effects of transmission line pressurization on peak power rating. Note that

P

= the rating of the line at the increased pressure, and P = the rating of the line at atmo-

spheric pressure.

Table 7.2

Electrical Characteristics of Common RF Connectors

Connector Type

DC Test Voltage (kW)

Peak Power (kW)

SMA

1.0

1.2

BNC, TNC

1.5

2.8

N, UHF

2.0

4.9

GR

3.0

11

HN, 7/16

4.0

20

LC

5.0

31

7/8 EIA, F Flange

6.0

44

1-5/8 EIA

11.0

150

3-1/8 EIA

19.0

4

© 1999 CRC Press LLC

Figure 7.6

Attenuation characteristics for a selection of coaxial cables: (

a) 50

line, (b)

75

line.

(

b)

(

a)

© 1999 CRC Press LLC

Where:

= attenuation in decibels per 100 meters

P

1

= input power into a 100-meter line terminated with the nominal value of its

characteristic impedance

P

2

= power measured at the end of the line

Stated in terms of efficiency (E, percent):

E

P

P

o

i

=

×

100

(7.8)

Where:
P

i

= power delivered to the input of the transmission line

P

o

= power delivered to the antenna

The relationship between efficiency and loss in decibels (insertion loss) is illustrated
in

Figure 7.7

.

Manufacturer-supplied attenuation curves typically are guaranteed to within ap-

proximately ±5 percent. The values given usually are rated at 24ºC (75ºF) ambient tem-
perature. Attenuation increases slightly with higher temperature or applied power. The
effects of ambient temperature on attenuation are illustrated in

Figure 7.8

.

Loss in connectors is negligible, except for small (SMA and TNC) connectors at fre-

quencies of several gigahertz and higher. Small connectors used at high frequencies
typically add 0.1 dB of loss per connector.

When a transmission line is attached to a load, such as an antenna, the VSWR of the

load increases the total transmission loss of the system. This effect is small under con-
ditions of low VSWR.

Figure 7.9

illustrates the interdependence of these two elements.

Phase Stability

A coaxial cable expands as the temperature of the cable increases, causing the electri-
cal length of the line to increase as well. This factor results in phase changes that are a
function of operating temperature. The phase relationship can be described by the
equation:

=

×

× × × ×

-

366

10

7

.

P

L

T

F

(7.9)

Where:
= phase change in degrees
P = phase-temperature coefficient of the cable
L = length of coax in feet
T = temperature range (minimum-to-maximum operating temperature)
F = frequency in Mhz

Phase changes that are a function of temperature are important in systems utilizing
multiple transmission lines, such as a directional array fed from a single phasing

© 1999 CRC Press LLC

source. To maintain proper operating parameters, the phase changes of the cables
must be minimized. Specially designed coaxial cables that offer low-phase-tempera-
ture characteristics are available. Two types of coax are commonly used for this pur-
pose:

Figure 7.7

Conversion chart showing the relationship between decibel loss and effi-

ciency of a transmission line: (

a) high-loss line, (b) low-loss line.

(

a)

(

b)

© 1999 CRC Press LLC

Figure 7.8

The variation of coaxial cable attenuation as a function of ambient tempera-

ture.

Figure 7.9

The effect of load VSWR on transmission line loss.

© 1999 CRC Press LLC

·

Phase-stabilized cables, which have undergone extensive temperature cycling
until such time as they exhibit their minimum phase-temperature coefficient

·

Phase-compensated cables, in which changes in the electrical length have been
minimized through adjustment of the mechanical properties of the dielectric and
inner/outer conductors

7.2.4 Mechanical Parameters

Corrugated copper cables are designed to withstand bending with no change in prop-
erties. Low-density foam- and air-dielectric cables generally have a minimum bend-
ing radius of 10 times the cable diameter. Super flexible versions provide a smaller al-
lowable bending radius.

Rigid transmission line will not tolerate bending. Instead, transition elements (el-

bows) of various sizes are used. Individual sections of rigid line are secured by multiple
bolts around the circumference of a coupling flange.

When a large cable must be used to meet attenuation requirements, short lengths of a

smaller cable (jumpers or pigtails) may be used on either end for ease of installation in
low-power systems. The tradeoff is slightly higher attenuation and some additional
cost.

The tensile strength of a cable is defined as the axial load that may be applied to the

line with no more than 0.2 percent permanent deformation after the load is released.
When divided by the weight per foot of cable, this gives an indication of the maximum
length of cable that is self-supporting and therefore can be installed readily on a tower
with a single hoisting grip. This consideration usually applies only to long runs of cor-
rugated line; rigid line is installed one section at a time.

The crush strength of a cable is defined as the maximum force per linear inch that

may be applied by a flat plate without causing more than a 5 percent deformation of the
cable diameter. Crush strength is a good indicator of the ruggedness of a cable and its
ability to withstand rough handling during installation.

Cable jacketing affords mechanical protection during installation and service.

Semiflexible cables typically are supplied with a jacket consisting of low-density poly-
ethylene blended with 3 percent carbon black for protection from the sun's ultraviolet
rays, which can degrade plastics over time. This approach has proved to be effective,
yielding a life expectancy of more than 20 years. Rigid transmission line has no cover-
ing over the outer conductor.

For indoor applications, where fire-retardant properties are required, cables can be

supplied with a fire-retardant jacket, usually listed by Underwriters Laboratories. Note
that under the provisions of the National Electrical Code, outside plant cables such as
standard black polyethylene-jacketed coaxial line may be run as far as 50 ft inside a
building with no additional protection. The line also may be placed in conduit for lon-
ger runs.

Low-density foam cables are designed to prevent water from traveling along their

length, should it enter through damage to the connector or the cable sheath. This is ac-
complished by mechanically locking the outer conductor to the foam dielectric by an-

© 1999 CRC Press LLC

nular corrugations. Annular or ring corrugations, unlike helical or screw-thread-type
corrugations, provide a water block at each corrugation. Closed-cell polyethylene di-
electric foam is bonded to the inner conductor, completing the moisture seal.

A coaxial cable line is only as good as the connectors used to tie it together. The con-

nector interface must provide a weatherproof bond with the cable to prevent water from
penetrating the connection. This is ensured by using O-ring seals. The cable-connector
interface also must provide a good electrical bond that does not introduce a mismatch
and increase VSWR. Good electrical contact between the connector and the cable en-
sures that proper RF shielding is maintained.

7.3

Waveguide

As the operating frequency of a system reaches into the UHF band, waveguide-based
transmission line systems become practical. From the mechanical standpoint, wave-
guide is simplicity itself. There is no inner conductor; RF energy is launched into the
structure and propagates to the load. Several types of waveguide are available, includ-
ing rectangular, square, circular, and elliptical. Waveguide offers several advantages
over coax. First, unlike coax, waveguide can carry more power as the operating fre-
quency increases. Second, efficiency is significantly better with waveguide at higher
frequencies.

Rectangular waveguide commonly is used in high-power transmission systems. Cir-

cular waveguide also may be used, especially for applications requiring a cylindrical
member, such as a rotating joint for an antenna feed. The physical dimensions of the
guide are selected to provide for propagation in the dominant (lowest-order) mode.

Waveguide is not without its drawbacks, however. Rectangular or square guide con-

stitutes a large windload surface, which places significant structural demands on a
tower. Because of the physical configuration of rectangular and square guide, pressur-
ization is limited, depending on the type of waveguide used (0.5 psi is typical). Exces-
sive pressure can deform the guide shape and result in increased VSWR. Wind also
may cause deformation and ensuing VSWR problems. These considerations have led
to the development of circular and elliptical waveguide.

7.3.1 Propagation Modes

Propagation modes for waveguide fall into two broad categories:

·

Transverse-electric (TE) waves

·

Transverse-magnetic (TM) waves

With TE waves, the electric vector (E vector) is perpendicular to the direction of prop-
agation. With TM waves, the magnetic vector (H vector) is perpendicular to the direc-
tion of propagation. These propagation modes take on integers (from 0 or 1 to infin-
ity) that define field configurations. Only a limited number of these modes can be
propagated, depending on the dimensions of the guide and the operating frequency.

© 1999 CRC Press LLC

Energy cannot propagate in waveguide unless the operating frequency is above the

cutoff frequency. The cutoff frequency for rectangular guide is:

F

C

a

c

=

×

2

(7.10)

Where:
F

c

= waveguide cutoff frequency

c = 1.179

× 10

10

inches per second (the velocity of light)

a = the wide dimension of the guide

The cutoff frequency for circular waveguide is defined by the following equation:

F

c

a

c

=

×

3 41

.

(7.11)

Where:
a

= the radius of the guide

There are four common propagation modes in waveguide:

·

TE

0,1

, the principal mode in rectangular waveguide.

·

TE

1,0

, also used in rectangular waveguide.

·

TE

1,1

, the principal mode in circular waveguide. TE

1,1

develops a complex propa-

gation pattern with electric vectors curving inside the guide. This mode exhibits
the lowest cutoff frequency of all modes, which allows a smaller guide diameter
for a specified operating frequency.

·

TM

0,1

, which has a slightly higher cutoff frequency than TE

1,1

for the same size

guide. Developed as a result of discontinuities in the waveguide, such as flanges
and transitions, TM

0,1

energy is not coupled out by either dominant or cross-polar

transitions. The parasitic energy must be filtered out, or the waveguide diameter
chosen carefully to reduce the unwanted mode.

The field configuration for the dominant mode in rectangular waveguide is illus-

trated in

Figure 7.10

. Note that the electric field is vertical, with intensity maximum at

the center of the guide and dropping off sinusoidally to zero intensity at the edges. The
magnetic field is in the form of loops that lie in planes that are at right angles to the elec-
tric field (parallel to the top and bottom of the guide). The magnetic field distribution is
the same for all planes perpendicular to the Y-axis. In the X direction, the intensity of the
component of magnetic field that is transverse to the axis of the waveguide (the compo-
nent in the direction of X) is at any point in the waveguide directly proportional to the in-
tensity of the electric field at that point. This entire configuration of fields travels in the
direction of the waveguide axis (the Z direction in

Figure 7.10

).

© 1999 CRC Press LLC

The field configuration for the TE

1,1

mode in circular waveguide is illustrated in

Fig-

ure 7.11

. The TE

1,1

mode has the longest cutoff wavelength and is, accordingly, the

dominant mode. The next higher mode is the TM

0,1

, followed by TE

2,1

.

Dual-Polarity Waveguide

Waveguide will support dual-polarity transmission within a single run of line. A com-
bining element (dual-polarized transition) is used at the beginning of the run, and a
splitter (polarized transition) is used at the end of the line. Square waveguide has
found numerous applications in such systems. Theoretically, the TE

1,0

and TE

0,1

modes

are capable of propagation without cross coupling, at the same frequency, in lossless
waveguide of square cross section. In practice, surface irregularities, manufacturing
tolerances, and wall losses give rise to TE

1,0

- and TE

0,1

-mode cross conversion. Be-

cause this conversion occurs continuously along the waveguide, long guide runs usu-
ally are avoided in dual-polarity systems.

Efficiency

Waveguide losses result from the following:

Figure 7.10

Field configuration of the dominant or TE

1,0

mode in a rectangular wave-

guide: (

a) side view, (b) end view, (c) top view.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

·

Power dissipation in the waveguide walls and the dielectric material filling the
enclosed space

·

Leakage through the walls and transition connections of the guide

·

Localized power absorption and heating at the connection points

The operating power of waveguide may be increased through pressurization. Sulfur

hexafluoride commonly is used as the pressurizing gas.

7.3.2 Ridged Waveguide

Rectangular waveguide may be ridged to provide a lower cutoff frequency, thereby
permitting use over a wider frequency band. As illustrated in

Figure 7.12

, one- and

two-ridged guides are used. Increased bandwidth comes at the expense of increased
attenuation, relative to an equivalent section of rectangular guide.

7.3.3 Circular Waveguide

Circular waveguide offers several mechanical benefits over rectangular or square
guide. The windload of circular guide is two-thirds that of an equivalent run of rectan-
gular waveguide. It also presents lower and more uniform windloading than rectangu-
lar waveguide, reducing tower structural requirements.

The same physical properties of circular waveguide that give it good power handling

and low attenuation also result in electrical complexities. Circular waveguide has two
potentially unwanted modes of propagation: the cross-polarized TE

1,1

and TM

0,1

modes.

Figure 7.11

Field configuration of the dominant mode in circular waveguide.

© 1999 CRC Press LLC

Circular waveguide, by definition, has no short or long dimension and, conse-

quently, no method to prevent the development of cross-polar or orthogonal energy.
Cross-polar energy is formed by small ellipticities in the waveguide. If the cross-polar
energy is not trapped out, the parasitic energy can recombine with the dominant-mode
energy.

Parasitic Energy

Hollow circular waveguide works as a high-Q resonant cavity for some energy and as
a transmission medium for the rest. The parasitic energy present in the cavity formed
by the guide will appear as increased VSWR if not disposed of. The polarization in
the guide meanders and rotates as it propagates from the source to the load. The end
pieces of the guide, typically circular-to-rectangular transitions, are polarization-sen-
sitive. See

Figure 7.13(a)

. If the polarization of the incidental energy is not matched

to the transition, energy will be reflected.

Several factors can cause this undesirable polarization. One cause is out-of-round

guides that result from nonstandard manufacturing tolerances. In

Figure 7.13

, the solid

lines depict the situation at launching: perfectly circular guide with perpendicular po-
larization. However, certain ellipticities cause polarization rotation into unwanted
states, while others have no effect. A 0.2 percent change in diameter can produce a 40
dB cross-polarization component per wavelength. This is roughly 0.03 in for 18 in of
guide length.

Other sources of cross polarization include twisted and bent guides, out-of-round-

ness, offset flanges, and transitions. Various methods are used to dispose of this energy
trapped in the cavity, including absorbing loads placed at the ground and/or antenna
level.

7.3.4 Doubly Truncated Waveguide

The design of doubly truncated waveguide (DTW) is intended to overcome the prob-
lems that can result from parasitic energy in a circular waveguide. As shown in

Figure

Figure 7.12

Ridged waveguide: (

a) single-ridged, (b) double-ridged.

(

a) (b)

© 1999 CRC Press LLC

7.14

, DTW consists of an almost elliptical guide inside a circular shell. This guide

does not support cross polarization; tuners and absorbing loads are not required. The
low windload of hollow circular guide is maintained, except for the flange area.

Each length of waveguide is actually two separate pieces: a doubly truncated center

section and a circular outer skin, joined at the flanges on each end. A large hole in the
broadwall serves to pressurize the circular outer skin. Equal pressure inside the DTW
and inside the circular skin ensures that the guide will not "breathe" or buckle as a result
of rapid temperature changes.

DTW exhibits about 3 percent higher windloading than an equivalent run of circular

waveguide (because of the transition section at the flange joints), and 32 percent lower
loading than comparable rectangular waveguide.

7.3.5 Impedance Matching

The efficient flow of power from one type of transmission medium to another requires
matching of the field patterns across the boundary to launch the wave into the second
medium with a minimum of reflections. Coaxial line typically is matched into rectan-
gular waveguide by extending the center conductor of the coax through the broadwall

RF and Switching 377

Figure 7.13

The effects of parasitic energy in circular waveguide: (

a) trapped

cross-polarization energy, (

b) delayed transmission of the trapped energy.

(

a)

(

b)

© 1999 CRC Press LLC

of the guide, parallel to the electric field lines across the guide. Alternatively, the cen-
ter conductor can be formed into a loop and oriented to couple the magnetic field to
the guide mode.

Standing waves are generally to be avoided in waveguide for the same reasons that

they are to be avoided in transmission lines. Accordingly, it is usually necessary to pro-
vide impedance-matching systems in waveguides to eliminate standing waves. One ap-
proach involves the introduction of a compensating reflection in the vicinity of a load
that neutralizes the standing waves that would exist in the system because of an imper-
fect match. A probe or tuning screw commonly is used to accomplish this, as illustrated
in

Figure 7.15

. The tuning screw projects into the waveguide in a direction parallel to

the electric field. This is equivalent to shunting a capacitive load across the guide. The
susceptance of the load increases with extension into the guide up to 1/4 wavelength.
When the probe is exactly 1/4 wavelength long, it becomes resonant and causes the
guide to behave as though there were an open circuit at the point of the resonant probe.
Probes longer than 1/4 wavelength but shorter than 3/4 wavelength introduce inductive
loading. The extent to which such a probe projects into the waveguide determines the
compensating reflection, and the position of the probe with respect to the standing wave
pattern to be eliminated determines the phasing of the reflected wave.

Dielectric slugs produce an effect similar to that of a probe. The magnitude of the ef-

fect depends upon the following considerations:

·

The dielectric constant of the slug

·

Thickness of the slug in an axial direction

·

Whether the slug extends entirely across the waveguide

The phase of the reflected wave is controlled by varying the axial position of the slug.

Figure 7.14

Physical construction of doubly truncated waveguide.

© 1999 CRC Press LLC

There are several alternatives to the probe and slug for introducing controllable ir-

regularity for impedance matching, including a metallic barrier or window placed at
right angles to the axis of the guide, as illustrated in

Figure 7.16

. Three configurations

are shown:

·

The arrangement illustrated in

Figure 7.16a

produces an effect equivalent to

shunting the waveguide with an inductive reactance.

·

The arrangement shown in

Figure 7.16b

produces the effect of a shunt capacitive

susceptance.

·

The arrangement shown in

Figure 7.16c

produces an inductive shunt

susceptance.

The waveguide equivalent of the coaxial cable tuning stub is a tee section, illustrated

in

Figure 7.17

. The magnitude of the compensating effect is controlled by the position

of the short-circuiting plug in the branch. The phase of the compensating reflected
wave produced by the branch is determined by the position of the branch in the guide.

Waveguide Filters

A section of waveguide beyond cutoff constitutes a simple high-pass reflective filter.
Loading elements in the form of posts or stubs may be employed to supply the
reactances required for conventional lumped-constant filter designs.

Absorption filters prevent the reflection of unwanted energy by incorporating lossy

material in secondary guides that are coupled through leaky walls (small sections of

Figure 7.15

A probe configured to introduce a reflection in a waveguide that is adjust-

able in magnitude and phase.

© 1999 CRC Press LLC

guide beyond cutoff in the passband). Such filters typically are used to suppress har-
monic energy.

7.3.6 Installation Considerations

Waveguide system installation is both easier and more difficult than traditional trans-
mission line installation. There is no inner conductor to align, but alignment pins
must be set and more bolts are required per flange. Transition hardware to accommo-
date loads and coax-to-waveguide interfacing also is required.

Flange reflections can add up in phase at certain frequencies, resulting in high

VSWR. The length of the guide must be chosen so that flange reflection buildup does
not occur within the operating bandwidth.

Flexible sections of waveguide are used to join rigid sections or components that

cannot be aligned otherwise. Flexible sections also permit controlled physical move-
ment resulting from thermal expansion of the line. Such hardware is available in a vari-
ety of forms. Corrugated guide commonly is produced by shaping thin-wall seamless
rectangular tubing. Flexible waveguide can accommodate only a limited amount of me-
chanical movement. Depending on the type of link, the manufacturer may specify a
maximum number of bends.

Figure 7.16

Waveguide obstructions used to introduce compensating reflections: (

a)

inductive window, (

b) capacitive window, (c) post (inductive) element.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

Tuning

Circular waveguide must be tuned. This requires a two-step procedure. First, the
cross-polar TE

1,1

component is reduced, primarily through axial ratio compensators or

mode optimizers. These devices counteract the net system ellipticity and indirectly
minimize cross-polar energy. The cross-polar filters also may be rotated to achieve
maximum isolation between the dominant and cross-polar modes. Cross-polar energy
manifests itself as a net signal rotation at the end of the waveguide run. A perfect sys-
tem would have a net rotation of zero.

In the second step, tuning slugs at both the top and bottom of the waveguide run are

adjusted to reduce the overall system VSWR. Tuning waveguide can be a complicated
and time-consuming procedure. Once set, however, tuning normally does not drift and
must be repeated only if major component changes are made.

Waveguide Hardware

Increased use of waveguide has led to the development of waveguide-based hardware
for all elements from the output of the RF generator to the load. Waveguide-based fil-
ters, elbows, directional couplers, switches, combiners, and diplexers are currently
available. The RF performance of a waveguide component is usually better than the
same item in coax. This is especially true in the case of diplexers and filters. Wave-
guide-based hardware provides lower attenuation and greater power-handling capa-
bility for a given physical size.

Figure 7.17

Waveguide stub elements used to introduce compensating reflections: (

a)

series

tee element, (b) shunt tee element.

(

a)

(

b)

© 1999 CRC Press LLC

7.3.7 Cavity Resonators

Any space completely enclosed by conducting walls can contain oscillating electro-
magnetic fields. Such a cavity possesses certain frequencies at which it will resonate
when excited by electrical oscillations. These cavity resonators find extensive use as
resonant circuits at VHF and above. (The use of cavities in power grid tube amplifiers
is discussed in Chapter 4.) Advantages of cavity resonators over conventional LC cir-
cuits include:

·

Simplicity in design.

·

Relatively large physical size compared with alternative methods of obtaining
resonance. This attribute is important in high-power, high-frequency applica-
tions.

·

High Q.

·

The capability to configure the cavity to develop an extremely high shunt imped-
ance.

Cavity resonators commonly are used at wavelengths on the order of 10 cm or less.

The simplest cavity resonator is a section of waveguide shorted at each end with a

length l equal to:

l

g

=

2

(7.12)

Where:

g

= the guide wavelength

This configuration results in a resonance similar to that of a 1/2-wavelength transmis-
sion line short-circuited at the receiving end.

A sphere or any other enclosed surface (irrespective of how irregular the outline)

also may be used to form a cavity resonator.

Any given cavity is resonant at a number of frequencies, corresponding to the differ-

ent possible field conditions that can exist within the space. The resonance having the
longest wavelength (lowest frequency) is termed the dominant or fundamental reso-
nance. The resonant wavelength is proportional to the size of the resonator. If all di-
mensions are doubled, the wavelength corresponding to resonance will likewise be
doubled. The resonant frequency of a cavity can be changed by incorporating one or
more of the following mechanisms:

·

Altering the mechanical dimensions of the cavity. Small changes can be achieved
by flexing walls, but large changes require some form of sliding member.

·

Coupling reactance into the resonator through a coupling loop.

·

Introducing a movable copper paddle into the cavity. A paddle placed inside the
resonator will affect the normal distribution of flux and tend to raise the resonant
frequency by an amount determined by the orientation of the paddle.

© 1999 CRC Press LLC

The Q of a cavity resonator has the same significance as for a conventional resonant

circuit. Q can be defined for a cavity by the following relationship:

Q

E

E

s

l

= 2

(7.13)

Where:
E

s

= energy stored

E

l

= energy lost per cycle

The energy stored is proportional to the square of the magnetic flux density integrated
throughout the volume of the resonator. The energy lost in the walls is proportional to
the square of the magnetic flux density integrated over the surface of the cavity. To
obtain high Q, the resonator should have a large ratio of volume to surface area, be-
cause it is the volume that stores energy and the surface area that dissipates energy.

Coupling can be obtained from a resonator by means of a coupling loop or coupling

electrode. Magnetic coupling is accomplished through the use of a loop oriented so as
to enclose magnetic flux lines existing in the desired mode of operation. This technique
is illustrated in

Figure 7.18

. A current passed through the loop will excite oscillations of

this mode. Conversely, oscillations existing in the resonator will induce a voltage in the
coupling loop. The magnitude of the coupling can be controlled by rotating the loop;
the coupling reduces to zero when the plane of the loop is parallel to the magnetic flux.

Coupling of a resonator also may be accomplished through the use of a probe or

opening in one wall of the cavity.

7.4

RF Combiner and Diplexer Systems

The basic purpose of an RF combiner is to add two or more signals to produce an out-
put signal that is a composite of the inputs. The combiner performs this signal addi-
tion while providing isolation between inputs. Combiners perform other functions as
well, and can be found in a wide variety of RF equipment utilizing power vacuum
tubes. Combiners are valuable devices because they permit multiple amplifiers to

Figure 7.18

Cavity resonator coupling: (

a) coupling loop, (b) equivalent circuit.

(

a)

(

b)

© 1999 CRC Press LLC

drive a single load. The isolation provided by the combiner permits tuning adjust-
ments to be made on one amplifier--including turning it on or off--without signifi-
cantly affecting the operation of the other amplifier. In a typical application, two am-
plifiers drive the hybrid and provide two output signals:

·

A combined output representing the sum of the two input signals, typically di-
rected toward the antenna.

·

A difference output representing the difference in amplitude and phase between
the two input signals. The difference output typically is directed toward a dummy
(reject) load.

For systems in which more than two amplifiers must be combined, two or more
combiners are cascaded.

Diplexers are similar in nature to combiners but permit the summing of output sig-

nals from two or more amplifiers operating at different frequencies. This allows, for ex-
ample, the outputs of several transmitters operating on different frequencies to utilize a
single broadband antenna.

7.4.1 Passive Filters

A filter is a multiport-network designed specifically to respond differently to signals
of different frequency [1]. This definition excludes networks, which incidentally be-
have as filters, sometimes to the detriment of their main purpose. Passive filters are
constructed exclusively with passive elements (i.e., resistors, inductors, and capaci-
tors). Filters are generally categorized by the following general parameters:

·

Type

·

Alignment (or class)

·

Order

Filter Type

Filters are categorized by type, according to the magnitude of the frequency response,
as one of the following [1]:

·

Low-pass (LP)

·

High-pass (HP)

·

Band-pass (BP)

·

Band-stop (BS).

The terms band-reject or notch are also used as descriptive of the BS filter. The term
all-pass is sometimes applied to a filter whose purpose is to alter the phase angle
without affecting the magnitude of the frequency response. Ideal and practical inter-
pretations of the types of filters and the associated terminology are illustrated in

Fig-

ure 7.19

.

© 1999 CRC Press LLC

In general, the voltage gain of a filter in the stop band (or attenuation band) is less

than 2 2 (

0.707) times the maximum voltage gain in the pass band. In logarithmic

terms, the gain in the stop band is at least 3.01 dB less than the maximum gain in the
pass band. The cutoff (break or corner) frequency separates the pass band from the stop
band. In BP and BS filters, there are two cutoff frequencies, sometimes referred to as
the lower and upper cutoff frequencies. Another expression for the cutoff frequency is
half-power frequency, because the power delivered to a resistive load at cutoff fre-
quency is one-half the maximum power delivered to the same load in the pass band. For
BP and BS filters, the center frequency is the frequency of maximum or minimum re-
sponse magnitude, respectively, and bandwidth is the difference between the upper and
lower cutoff frequencies. Rolloff is the transition from pass band to stop band and is
specified in gain unit per frequency unit (e.g., gain unit/Hz, dB/decade, dB/octave, etc.)

Filter Alignment

The alignment (or class) of a filter refers to the shape of the frequency response [1].
Fundamentally, filter alignment is determined by the coefficients of the filter network

Figure 7.19

Filter characteristics by type: (

a) low-pass, (b) high-pass, (c) bandpass, (d)

bandstop. (

From [1]. Used with permission.)

(

a)

(

b)

(

c)

(

d)

© 1999 CRC Press LLC

transfer function, so there are an indefinite number of filter alignments, some of
which may not be realizable. The more common alignments are:

·

Butterworth

·

Chebyshev

·

Bessel

·

Inverse Chebyshev

·

Elliptic (or Cauer)

Each filter alignment has a frequency response with a characteristic shape, which pro-
vides some particular advantage. (See

Figure 7.20

.) Filters with Butterworth,

Chebyshev, or Bessel alignment are called all-pole filters because their low-pass
transfer functions have no zeros.

Table 7.3

summarizes the characteristics of the stan-

dard filter alignments.

Filter Order

The order of a filter is equal to the number of poles in the filter network transfer func-
tion [1]. For a lossless LC filter with resistive (nonreactive) termination, the number
of reactive elements (inductors or capacitors) required to realize a LP or HP filter is
equal to the order of the filter. Twice the number of reactive elements are required to
realize a BP or a BS filter of the same order. In general, the order of a filter deter-
mines the slope of the rolloff--the higher the order, the steeper the rolloff. At fre-
quencies greater than approximately one octave above cutoff (i.e., f > 2 f

c

), the rolloff

for all-pole filters is 20n dB/decade (or approximately 6n dB/octave), where n is the

Figure 7.20

Filter characteristics by alignment, third-order, all-pole filters: (

a) magni-

tude, (

b) magnitude in decibels. (From [1]. Used with permission.)

(

a)

(

b)

© 1999 CRC Press LLC

order of the filter (

Figure 7.21

). In the vicinity of f

c

, both filter alignment and filter or-

der determine rolloff.

7.4.2 Four-Port Hybrid Combiner

A hybrid combiner (coupler) is a reciprocal four-port device that can be used for ei-
ther splitting or combining RF energy over a wide range of frequencies. An exploded
view of a typical 3 dB 90° hybrid is illustrated in

Figure 7.22

. The device consists of

two identical parallel transmission lines coupled over a distance of approximately
one-quarter wavelength and enclosed within a single outer conductor. Ports at the

Table 7.3

Summary of Standard Filter Alignments (

After [1].)

Alignment

Pass Band

Description

Stop Band

Description

Comments

Butterworth

Monotonic

Monotonic

All-pole; maximally
flat

Chebyshev

Rippled

Monotonic

All-pole

Bessel

Monotonic

Monotonic

All-pole; constant
phase shift

Inverse Chebyshev Monotonic

Rippled

Elliptic (or Cauer)

Rippled

Rippled

Figure 7.21

The effects of filter order on rolloff (Butterworth alignment). (

From [1]. Used

with permission.)

© 1999 CRC Press LLC

same end of the coupler are in phase, and ports at the opposite end of the coupler are
in quadrature (90° phase shift) with respect to each other.

The phase shift between the two inputs or outputs is always 90° and is virtually inde-

pendent of frequency. If the coupler is being used to combine two signals into one out-
put, these two signals must be fed to the hybrid in phase quadrature. When the coupler is
used as a power splitter, the division is equal (half-power between the two ports). The
hybrid presents a constant impedance to match each source.

Operation of the combiner can best be understood through observation of the device

in a practical application.

Figure 7.23

shows a four-port hybrid combiner used to add

the outputs of two transmitters to feed a single load. The combiner accepts one RF
source and splits it equally into two parts. One part arrives at output port C with 0°
phase (no phase delay; it is the reference phase). The other part is delayed by 90° at port
D. A second RF source connected to input port B, but with a phase delay of 90°, also
will split in two, but the signal arriving at port C now will be in phase with source 1, and
the signal arriving at port D will cancel, as shown in the figure.

Output port C, the summing point of the hybrid, is connected to the load. Output port

D is connected to a resistive load to absorb any residual power resulting from slight dif-
ferences in amplitude and/or phase between the two input sources. If one of the RF in-
puts fails, half of the remaining transmitter output will be absorbed by the resistive load
at port D.

The four-port hybrid works only when the two signals being mixed are identical in

frequency and amplitude, and when their relative phase is 90°.

Operation of the hybrid can best be described by a scattering matrix in which vectors

are used to show how the device operates. Such a matrix is shown in

Table 7.4

. In a 3 dB

hybrid, two signals are fed to the inputs. An input signal at port 1 with 0° phase will ar-
rive in phase at port 3, and at port 4 with a 90° lag (90°) referenced to port 1. If the sig-
nal at port 2 already contains a 90° lag (90° referenced to port 1), both input signals
will combine in phase at port 4. The signal from port 2 also experiences another 90°

Figure 7.22

Physical model of a 90° hybrid combiner.

© 1999 CRC Press LLC

change in the hybrid as it reaches port 3. Therefore, the signals from ports 1 and 2 can-
cel each other at port 3.

If the signal arriving at port 2 leads by 90° (mode 1 in the table), the combined power

from ports 1 and 2 appears at port 4. If the two input signals are matched in phase (mode
4), the output ports (3 and 4) contain one-half of the power from each of the inputs.

If one of the inputs is removed, which would occur in a transmitter failure, only one

hybrid input receives power (mode 5). Each output port then would receive one-half the
input power of the remaining transmitter, as shown.

The input ports present a predictable load to each amplifier with a VSWR that is

lower than the VSWR at the output port of the combiner. This characteristic results
from the action of the difference port, typically connected to a dummy load. Reflected
power coming into the output port will be directed to the reject load, and only a portion
will be fed back to the amplifiers.

Figure 7.24

illustrates the effect of output port

VSWR on input port VSWR, and on the isolation between ports.

As noted previously, if the two inputs from the separate amplifiers are not equal in

amplitude and not exactly in phase quadrature, some power will be dissipated in the dif-
ference port reject load.

Figure 7.25

plots the effect of power imbalance, and

Figure

7.26

plots the effects of phase imbalance. The power lost in the reject load can be re-

duced to a negligible value by trimming the amplitude and/or phase of one (or both) am-
plifiers.

7.4.3 Non-Constant-Impedance Diplexer

Diplexers are used to combine amplifiers operating on different frequencies (and at
different power levels) into a single output. Such systems typically are utilized to sum
different transmitter outputs to feed a single broadband antenna.

Figure 7.23

Operating principles of a hybrid combiner. This circuit is used to add two

identical signals at inputs A and B.

© 1999 CRC Press LLC

Table 7.4

Single 90° Hybrid System Operating Modes

© 1999 CRC Press LLC

The branch diplexer is the typical configuration for a diplexer that does not exhibit

constant-impedance inputs. As shown in

Figure 7.27

, the branch diplexer consists of

two banks of filters each feeding into a coaxial tee. The electrical length between each
filter output and the centerline of the tee is frequency-sensitive, but this fact is more of a
tuning nuisance than a genuine user concern.

For this type of diplexer, all of the electrical parameters are a function of the filter

characteristics. The VSWR, insertion loss, group delay, and rejection/isolation will be
the same for the overall system as they are for the individual banks of cavities. The ma-
jor limitation of this type of combiner is the degree of isolation that can be obtained for
closely spaced channels.

7.4.4 Constant-Impedance Diplexer

The constant-impedance diplexer employs 3 dB hybrids and filters with a terminating
load on the isolated port. The filters in this type of combiner can be either notch-type
or bandpass-type. The performance characteristics are noticeably different for each
design.

Figure 7.24

The effects of load VSWR on input VSWR and isolation: (

a) respective

curves, (

b) coupler schematic.

(

a)

(

b)

© 1999 CRC Press LLC

Figure 7.26

Phase sensitivity of a hybrid coupler.

Figure 7.25

The effects of power imbalance at the inputs of a hybrid coupler.

© 1999 CRC Press LLC

Band-Stop Diplexer

The band-stop (notch) constant-impedance diplexer is configured as shown in

Figure

7.28

. For this design, the notch filters must have a high Q response to keep insertion

loss low in the passband skirts. The high Q characteristic results in a sharp notch. De-
pending on the bandwidth required of the diplexer, two or more cavities may be lo-
cated in each leg of the diplexer. They are typically stagger tuned, one high and one
low for the two-cavity case. With this dual-cavity reject response in each leg of the
band-stop diplexer system, the following analysis explains the key performance spec-
ifications.

If frequency f

1

is fed into the top left port of

Figure 7.28

, it will be split equally into

the upper and lower legs of the diplexer. Both of these signals will reach the filters in
their respective leg and be rejected/reflected back toward the input hybrid, recombine,
and emerge through the lower left port, also known as the wideband output.

The VSWR looking into the f

1

input is near 1:1 at all frequencies in the band. Within

the bandwidth of the reject skirts, the observed VSWR is equal to the termination of the
wideband output. Outside of the passband, the signals will pass by the cavities, enter the
rightmost hybrid, recombine, and emerge into the dummy load. Consequently, the
out-of-band VSWR is, in fact, the VSWR of the load.

The insertion loss from the f

1

input to the wideband output is low, typically on the or-

der of 0.1 dB at carrier. This insertion loss depends on perfect reflection from the cavi-
ties. As the rejection diminishes on the skirts of the filters, the insertion loss from f

1

to

the wideband output increases.

The limitation in reject bandwidth of the cavities causes the insertion loss to rise at

the edges of the passband. The isolation from f

1

to the wideband input consists of a com-

bination of the reject value of the cavities plus the isolation of the rightmost hybrid.

Figure 7.27

Non-constant-impedance branch diplexer. In this configuration, two banks

of filters feed into a coaxial tee.

© 1999 CRC Press LLC

A signal entering at f

1

splits and proceeds in equal halves rightward through both the

upper and lower legs of the diplexer. It is rejected by the filters at carrier and, to a lesser
extent, on both sides of the carrier. Any residual signal that gets by the cavities reaches
the rightmost hybrid. There it recombines and emerges from the load port. The hybrid
provides a specified isolation from the load port to the wideband input port. This hybrid
isolation must be added to the filter rejection to obtain the total isolation from the f

1

in-

put to the wideband input.

If a signal is fed into the wideband input of the combiner shown in

Figure 7.28

, the

energy will split equally and proceed leftward along the upper and lower legs of the di-
plexer. Normally, f

1

is not fed into the wideband input. If it were, f

1

would be rejected by

the notch filters and recombine into the load. All other signals sufficiently removed
from f

1

will pass by the cavities with minimal insertion loss and recombine into the

wideband output.

The VSWR looking into the wideband input is equal to the VSWR at the output for

frequencies other than f

1

. If f

1

were fed into the wideband input, the VSWR would be

equal to the VSWR of the load.

Isolation from the wideband input to the f

1

input is simply the isolation available in

the leftmost 3 dB hybrid. This isolation is usually inadequate for high-power applica-
tions. To increase the isolation from the wideband input to the f

1

input, it is necessary to

use additional cavities between the f

1

transmitter and the f

1

input that will reject the fre-

quencies fed into the wideband input. Unfortunately, adding these cavities to the input

Figure 7.28

Band-stop (notch) constant-impedance diplexer module. This design in-

corporates two 3 dB hybrids and filters, with a terminating load on the isolated port.

© 1999 CRC Press LLC

line also cancels the constant-impedance input. Although the notch diplexer as shown
in the figure is truly a constant-impedance type of diplexer, the constant impedance is
presented to the two inputs by virtue of using the hybrids at the respective inputs.

The hybrids essentially cause the diplexer to act as an absorptive type of filter to all

out-of-band signals. The out-of-band signals generated by the transmitter are absorbed
by the load rather than reflected to the transmitter.

If a filter is added to the input to supplement isolation, the filter will reflect some

out-of-band signals back at the transmitter. The transmitter then will be seeing the
passband impedance of this supplemental filter rather than the constant impedance of
the notch diplexer. This is a serious deficiency for applications that require a true con-
stant-impedance input.

Bandpass Constant-Impedance Diplexer

The bandpass constant-impedance diplexer is shown in

Figure 7.29

. This system

takes all of the best features of diplexers and combines them into one unit. It also pro-
vides a constant-impedance input that need not be supplemented with input cavities
that rob the diplexer of its constant-impedance input.

The bandpass filters exhibit good bandwidth, providing near 1:1 VSWR across the

operating bandpass. Insertion is low at carrier (0.28 dB is typical), rising slightly at
bandpass extremes. Diplexer rejection, when supplemented by isolation of the hybrids,
provides ample transmitter-to-transmitter isolation. Group delay is typically excep-
tional, providing performance specifications similar to those of a branch-style
bandpass system. This configuration has the additional capability of providing high
port-to-port isolation between closely spaced operating channels, as well as a true con-
stant-impedance input.

Figure 7.29

Bandpass constant-impedance diplexer.

© 1999 CRC Press LLC

The hybrids shown in

Figure 7.29

work in a manner identical to those described for a

band-stop diplexer. However, the bandpass filters cause the system to exhibit perfor-
mance specifications that exceed the band-stop system in every way. Consider a signal
entering at the f

1

input.

Within the passbands of the filters, which are tuned to f

1

, the VSWR will be near 1:1

at carrier, rising slightly at the bandpass extremes. Because of the characteristic of the
leftmost hybrid, the VSWR is, in fact, a measure of the similarity of response of the top
and bottom bands of filters. The insertion loss looking from the f

1

input to the wideband

output will be similar to the insertion loss of the top and bottom filters individually. A
value of approximately 0.28 dB at carrier is typical.

Both the insertion loss and group delay can be determined by the design bandwidth

of the filters. Increasing bandwidth causes the insertion loss and group delay deviation
to decrease. Unfortunately, as the bandwidth increases with a given number of cavities,
the isolation suffers for closely spaced channels because the reject skirt of the filter de-
creases with increasing bandwidth.

Isolation of f

1

to the wideband input is determined as follows: A signal enters at the f

1

input, splits equally into the upper and lower banks of filters, passes with minimal loss
through the filters, and recombines into the wideband output of the rightmost hybrid.
Both the load and the wideband input ports are isolated by their respective hybrids to
some given value below the f

1

input level. Isolation of the f

1

input to the wideband input

is supplemented by the reject skirt of the next module.

A signal fed into the wideband input could be any frequency removed from f

1

by

some minimal amount. As the wideband signal enters, it will be split into equal halves
by the hybrid, then proceed to the left until the two components reach the reject skirts of
the filters. The filters will shunt all frequencies removed from f

1

by some minimal

amount. If the shunt energy is in phase for the given frequency when the signal is re-
flected back to the right hybrid, it will recombine into the wideband output. The VSWR
under these conditions will be equal to the termination at the wideband output. If the re-
ject skirts of the filter are sufficient, the insertion loss from wideband input to
wideband output will be minimal (0.03 dB is typical).

The isolation from the wideband input to f

1

can be determined as follows: A signal

enters at the wideband input, splits equally into upper and lower filters, and is rejected
by the filters. Any residual signal that passes through the filters in spite of the rejection
still will be in the proper phase to recombine into the load, producing additional isola-
tion to the f

1

input port. Thus, the isolation of the wideband input to the f

1

input is the

sum of the rejection from the filters and from the left hybrid.

Isolation of

f

1

to

f

2

Extending the use of the diplexer module into a multiplexer application supplements
the deficient isolation described previously (narrowband input performance), while
maintaining the constant-impedance input. In a multiplexer system, the wideband in-
put of one module is connected to the wideband output of the next module, as illus-
trated in

Figure 7.30

.

© 1999 CRC Press LLC

Figure 7.30

Schematic diagram of a six-module bandpass multiplexer. This configuration accommodates a split antenna design

and incorporates patch panels for bypass purposes.

©

1
999
CRC Press LLC

It has already been stated that the isolation from the f

1

input to the wideband input is

deficient, but additional isolation is provided by the isolation of the wideband output to
the f

2

input of the next module. Consider that f

1

has already experienced 30 dB isolation

to the wideband input of the same module. When this signal continues to the next mod-
ule through the wideband output of module 2, it will be split into equal halves and pro-
ceed to the left of module 2 until it reaches the reject skirts of the filters in module 2. As-
sume that these filters are tuned to f

2

and reject f

1

by at least 25 dB. The combined total

isolation of f

1

to f

2

is the sum of the 30 dB of the right hybrid in module 1, plus the 25 dB

of the reject skirts of module 2, for a total of 55 dB.

Intermodulation Products

The isolation just described is equal in magnitude to that for a band-stop module, but
it provides further protection against the generation of intermodulation products. The
most troublesome intermodulation (intermod) products usually occur when an incom-
ing (secondary) signal mixes with the second harmonic of a primary transmitter. (The
properties of intermodulation are discussed in more detail in Chapter 10.)

When the primary transmitter is operating on frequency A, the intermod will occur

at the frequency (

)

2

×

-

A

B. This formula invariably places the intermod from the pri-

mary transmitter symmetrically about the operating frequency. By an interesting coin-
cidence, the bandpass filters in the bandpass module also provide symmetrical reject
response on both sides of the primary operating frequency.

Assume again that the incoming signal is attenuated by 30 dB in the respective hy-

brid and by 25 dB in the filter, for a total of 55 dB. If an intermod still is generated in
spite of this isolation, it will emerge on the other skirt of the filter attenuated by 25 dB.
In the bandpass system, the incoming signal is attenuated by 55 dB and the resulting
outgoing spur by 25 dB, for a total of 80 dB suppression.

Interestingly, the entire 80 dB of attenuation is supplied by the diplexer regardless of

the turnaround loss of the transmitter. The tendency toward wideband final stage am-
plifiers in transmitters requires constant-impedance inputs. The transmitters also re-
quire increased isolation because they offer limited turnaround loss.

Group Delay

Group delay in the bandpass multiplexer module is equal to the sum of the
narrowband input group delay and the wideband input group delay of all modules be-
tween the input and the load (antenna). (Group delay is discussed in detail in Chapter
10.) The narrowband input group delay is a U-shaped response, with minimum at cen-
ter and rising to a maximum on both sides at the frequency where the reject rises to 3
dB. Group delay then decreases rapidly at first, then more slowly.

If the bandwidth of the pass band is made such that the group delay is ±25 ns over

±150 kHz (in an example system operating near 100 MHz), the 3 dB points will be at
±400 kHz and the out-of-band group delay will fall rapidly at ±800 kHz and possibly
±1.0 MHz. If there are no frequencies 800 kHz or 1.0 MHz removed upstream in other
modules, this poses no problem.

© 1999 CRC Press LLC

If modules upstream are tuned to 800 kHz or 1.0 MHz on either side, then the group

delay (when viewed at the upstream module) will consist of its own narrowband input
group delay plus the rapidly falling group delay of the wideband input of the closely
spaced downstream module. Under these circumstances, if good group delay is desired,
it is possible to utilize a group delay compensation module.

A group delay compensation module consists of a hybrid and two cavities used as

notch cavities. It typically provides a group delay response that is inverted, compared
with a narrowband input group delay. Because group delay is additive, the inverted re-
sponse subtracts from the standard response, effectively reducing the group delay devi-
ation.

It should be noted that the improvement in group delay is obtained at a cost of inser-

tion loss. In large systems (8 to 10 modules), the insertion loss can be high because of
the cumulative total of all wideband losses. Under these conditions, it may be more pru-
dent to accept higher group delay and retain minimal insertion loss.

7.4.5 Microwave Combiners

Hybrid combiners typically are used in microwave amplifiers to add the output energy
of individual power modules to provide the necessary output from an RF generator.
Quadrature hybrids effect a VSWR-canceling phenomenon that results in
well-matched power amplifier inputs and outputs that can be broadbanded with
proper selection of hybrid tees. Several hybrid configurations are possible, including
the following:

·

Split-tee

·

Branch-line

·

Magic-tee

·

Backward-wave

Key design parameters include coupling bandwidth, isolation, and ease of fabrication.
The equal-amplitude quadrature-phase reverse-coupled TEM 1/4-wave hybrid is par-
ticularly attractive because of its bandwidth and amenability to various physical im-
plementations. Such a device is illustrated in

Figure 7.31

.

7.4.6 Hot Switching Combiners

Switching RF is nothing new. Typically, the process involves coaxial switches, cou-
pled with the necessary logic to ensure that the "switch" takes place with no RF en-
ergy on the contacts. This process usually takes the system off-line for a few seconds
while the switch is completed. Through the use of hybrid combiners, however, it is
possible to redirect RF signals without turning the carrier off. This process is referred
to as hot switching.

Figure 7.32

illustrates two of the most common switching func-

tions (SPST and DPDT) available from hot switchers.

© 1999 CRC Press LLC

The unique phase-related properties of an RF hybrid make it possible to use the de-

vice as a switch. The input signals to the hybrid in

Figure 7.33a

are equally powered but

differ in phase by 90°. This phase difference results in the combined signals being
routed to the output terminal at port 4. If the relative phase between the two input sig-
nals is changed by 180°, the summed output then appears on port 3, as shown in

Figure

7.33b

. The 3 dB hybrid combiner, thus, functions as a switch.

This configuration permits the switching of two RF generators to either of two

loads. Remember, however, that the switch takes place when the phase difference be-
tween the two inputs is 90°. To perform the switch in a useful way requires adding a
high-power phase shifter to one input leg of the hybrid. The addition of the phase shifter
permits the full power to be combined and switched to either output. This configuration
of hybrid and phase shifter, however, will not permit switching a main or standby gener-
ator to a main or auxiliary load (DPDT function). To accomplish this additional switch,
a second hybrid and phase shifter must be added, as shown in

Figure 7.34

. This configu-

ration then can perform the following switching functions:

·

RF source 1 routed to output B

·

RF source 2 routed to output A

·

RF source 1 routed to output A

·

RF source 2 routed to output B

The key element in developing such a switch is a high-power phase shifter that does

not exhibit reflection characteristics. In this application, the phase shifter allows the
line between the hybrids to be electrically lengthened or shortened. The ability to adjust
the relative phase between the two input signals to the second hybrid provides the
needed control to switch the input signal between the two output ports.

If a continuous analog phase shifter is used, the transfer switch shown in

Figure 7.34

also can act as a hot switchless combiner where RF generators 1 and 2 can be combined

Figure 7.31

Reverse-coupled 1/4-wave hybrid combiner.

© 1999 CRC Press LLC

and fed to either output A or B. The switching or combining functions are accomplished
by changing the physical position of the phase shifter.

Note that it does not matter whether the phase shifter is in one or both legs of the sys-

tem. It is the phase difference (

1

2

- ) between the two input legs of the second hybrid

that is important. With 2-phase shifters, dual drives are required. However, the phase
shifter needs only two positions. In a 1-phase shifter design, only a single drive is re-
quired, but the phase shifter must have four fixed operating positions.

Figure 7.33

Hybrid switching configurations: (

a) phase set so that the combined energy

is delivered to port 4, (

b) phase set so that the combined energy is delivered to port 3.

(

a)

(

b)

Figure 7.32

Common RF switching configurations.

© 1999 CRC Press LLC

Phase Relationships

To better understand the dual-hybrid switching and combining process, it is necessary
to examine the primary switching combinations.

Table 7.5

lists the various combina-

tions of inputs, relative phase, and output configurations that are possible with the
single-phase shifter design.

Using vector analysis, note that when two input signals arrive in phase (mode 1) at

ports 1 and 2 with the phase shifter set to 0°, the circuit acts like a crossover network
with the power from input port 1 routed to output port 4. Power from input port 2 is
routed to output port 3. If the phase shifter is set to 180°, the routing changes, with port
1 being routed to port 3 and port 2 being routed to port 4.

Mode 2 represents the case where one of the dual-input RF generators has failed.

The output signal from the first hybrid arrives at the input to the second hybrid with a
90° phase difference. Because the second hybrid introduces a 90° phase shift, the vec-
tors add at port 4 and cancel at port 3. This effectively switches the working transmitter
connected to port 1 to output port 4, the load.

By introducing a 180° phase shift between the hybrids, as shown in modes 4 and 5, it

is possible to reverse the circuit. This allows the outputs to be on the same side of the cir-
cuit as the inputs. This configuration might be useful if generator 1 fails, and all power
from generator 2 is directed to a diplexer connected to output 4.

Normal operating configurations are shown in modes 6 and 7. When both genera-

tors are running, it is possible to have the combined power routed to either output port.
The switching is accomplished by introducing a ±90º phase shift between the hybrids.

As shown in the table, it is possible to operate in all the listed modes through the use

of a single-phase shifter. The phase shifter must provide four different phase positions.
A similar analysis would show that a 2-phase shifter design, with two positions for each
shifter, is capable of providing the same operational modes.

Figure 7.34

Additional switching and combining functions enabled by adding a second

hybrid and another phase shifter to a hot switching combiner.

© 1999 CRC Press LLC

Table 7.5

Operating Modes of the Dual 90° Hybrid/Single-Phase Shifter Combiner Sys-

tem

© 1999 CRC Press LLC

The key to making hybrid switches work in the real world lies in the phase shifter.

The dual 90° hybrid combiner just discussed requires a phase shifter capable of intro-
ducing a fixed phase offset of 90°, 0°, +90°, and +180°. This can be accomplished eas-
ily at low power levels through the use of a sliding short-circuit (trombone-type) line
stretcher. However, when high-frequency and high-power signals are being used, the
sliding short circuit is not an appropriate design choice. In a typical case, the phase
shifter must be able to handle 100 kW or more at UHF. Under these conditions, sliding
short-circuit designs are often unreliable. Therefore, three other methods have been de-
veloped:

·

Variable-dielectric vane

·

Dielectric post

·

Variable-phase hybrid

Variable-Dielectric Vane

The variable-dielectric vane consists of a long dielectric sheet mounted in a section of
rectangular waveguide, as illustrated in

Figure 7.35

. The dielectric sheet is long

enough to introduce a 270° phase shift when located in the center of the waveguide.
As the dielectric sheet is moved toward the wall, into the lower field, the phase shift
decreases. A single-sided phase shifter easily can provide the needed four positions.
A 2-stage 1/4-wave transformer is used on each end of the sheet to maintain a proper
match for any position over the desired operating band. The performance of a typical
switchless combiner, using the dielectric vane, is given in

Table 7.6

.

Dielectric Posts

Dielectric posts, shown in

Figure 7.36

, operate on the same principle as the dielectric

vane. The dielectric posts are positioned 1/4-wavelength apart from each other to can-
cel any mismatch, and to maintain minimal VSWR.

Figure 7.35

The dielectric vane switcher, which consists of a long dielectric sheet

mounted within a section of rectangular waveguide.

© 1999 CRC Press LLC

Variable-Phase Hybrid

The variable-phase hybrid, shown in

Figure 7.37

, relies on a 90° hybrid, similar to

those used in a combiner. With a unit vector incident on port 1, the power is split by
the 90° hybrid. The signal at ports 3 and 4 is reflected by the short circuit. These re-
flected signals are out of phase at port 1 and in phase at port 2. The relative phase of
the hybrid can be changed by moving the short circuit.

The variable-phase hybrid is linear with respect to position. Noncontacting

choke-type short-circuits, with high front-to-back ratios, are typically used in the de-
vice. The performance available from a typical high-power variable-phase switchless
combiner is given in

Table 7.7

.

Type

Input

Phase

Change

(deg)

VSWR

Input

Attenuation

Output

Attenuation

1 (dB)

2 (dB)

3 (dB)

4 (dB)

Single
input

T1

180

1.06

-

39

0.1

39

T1

0

1.05

-

39

39

0.1

T2

180

1.05

39

-

39

0.1

T2

0

1.06

39

-

0.1

39

Dual in-
put

T1+T2

270

1.06

-

-

0.1

36

T1+T2

90

1.06

-

-

36

0.1

Table 7.6

Typical Performance of a Dielectric Vane Phase Shifter

Figure 7.36

Dielectric post waveguide phase shifter.

© 1999 CRC Press LLC

7.4.7 Phased-Array Antenna Systems

The phased-array antenna, commonly used in radar applications, is a variation on the
hot switching combiner systems discussed in the previous section. The hot switching
combiner is used to control the flow of RF energy from one or more generators to two
or more loads, while the phased-array antenna is used to modify--in a controlled
fashion--the radiating characteristics of an antenna.

Phased-array antennas are steered by tilting the phase front independently in two or-

thogonal directions called the array coordinates. Scanning in either array coordinate
causes the beam to move along a cone whose center is at the center of the array. As the
beam is steered away from the array normal, the projected aperture in the beam's direc-
tion varies, causing the beamwidth to vary proportionately.

Arrays can be classified as either active or passive. Active arrays contain duplexers

and amplifiers behind every element or group of elements of the array. Passive arrays
are driven from a single feed point. Active arrays are capable of high-power operation.
Both passive and active arrays must divide the signal from a single transmission line
among all the elements of the system. This can be accomplished through one of the fol-
lowing methods:

·

Optical feed: a single feed, usually a monopulse horn, is used to illuminate the ar-
ray with a spherical phase front, illustrated in

Figure 7.38

. Power collected by the

rear elements of the array is transmitted through the phase shifters to produce a
planar front and steer the array. The energy then may be radiated from the other

Figure 7.37

Variable-phase hybrid phase shifter.

© 1999 CRC Press LLC

side of the array or reflected and reradiated through the collecting elements. In the
latter case, the array acts as a steerable reflector.

·

Corporate feed: a system utilizing a series-feed network (

Figure 7.39

) or paral-

lel-feed network (

Figure 7.40

). Both designs use transmission-line components

to divide the signal among the elements. Phase shifters can be located at the ele-
ments or within the dividing network. Both the series- and parallel-feed systems
have several variations, as shown in the figures.

·

Multiple-beam network: a system capable of forming simultaneous beams with a
given array. The Butler matrix, shown in

Figure 7.41

, is one such technique. It

connects the N elements of a linear array to N feed points corresponding to N
beam outputs. The phase shifter is one of the most critical components of the sys-
tem. It produces controllable phase shift over the operating band of the array.
Digital and analog phase shifters have been developed using both ferrites and pin
diodes.

Frequency scan is another type of multiple-beam network, but one that does not re-
quire phase shifters, dividers, or beam-steering computers. Element signals are cou-
pled from points along a transmission line. The electrical path length between ele-
ments is longer than the physical separation, so a small frequency change will cause a
phase change between elements that is large enough to steer the beam. This technique
can be applied only to one array coordinate. If a two-dimensional array is required,
phase shifters normally are used to scan the other coordinate.

Phase-Shift Devices

The design of a phase shifter must meet two primary criteria:

·

Low transmission loss

·

High power-handling capability

Table 7.7

Typical Performance of a Variable-Phase Hybrid Phase Shifter

Type

Input

Phase Change

(deg)

VSWR

Input

Attenuation

Output

Attenuation

1 (dB)

2 (dB)

3 (dB)

4 (dB)

Single
input

T1

180

1.06

-

36

0.1

52

T1

0

1.04

-

36

50

0.1

T2

180

1.06

36

-

52

0.1

T2

0

1.07

36

-

0.1

50

Dual
input

T1+T2

270

1.06

-

-

0.1

36

T1+T2

90

1.06

-

-

36

0.1

© 1999 CRC Press LLC

The Reggia-Spencer phase shifter meets both requirements. The device, illustrated in

Figure 7.42

, consists of a ferrite rod mounted inside a waveguide that delays the RF

signal passing through the waveguide, permitting the array to be steered. The amount
of phase shift can be controlled by the current in the solenoid, because of the effect a

Figure 7.38

Optical antenna feed systems: (

a) lens, (b) reflector.

(

b)

(

a)

© 1999 CRC Press LLC

magnetic field has on the permeability of the ferrite. This design is a reciprocal phase
shifter, meaning that the device exhibits the same phase shift for signals passing in ei-
ther direction (forward or reverse). Nonreciprocal phase shifters also are available,
where phase-shift polarity reverses with the direction of propagation.

Phase shifters also may be developed using pin diodes in transmission line net-

works. One configuration, shown in

Figure 7.43

, uses diodes as switches to change the

signal path length of the network. A second type uses pin diodes as switches to connect
reactive loads across a transmission line. When equal loads are connected with
1/4-wave separation, a pure phase shift results.

Figure 7.39

Series-feed networks: (

a) end feed, (b) center feed, (c) separate optimiza-

tion, (

d) equal path length, (e) series phase shifters.

(

a)

(

b)

(

c)

(

d)

(

e)

© 1999 CRC Press LLC

Figure 7.40

Types of parallel-feed networks: (

a) matched corporate feed, (b) reactive

corporate feed, (

c) reactive stripline, (d) multiple reactive divider.

(

a)

(

b)

(

c)

(

d)

Figure 7.41

The Butler beam-forming network.

© 1999 CRC Press LLC

Radar System Duplexer

The duplexer is an essential component of any radar system. The switching elements
used in a duplexer include gas tubes, ferrite circulators, and pin diodes. Gas tubes are
the simplest. A typical gas-filled TR tube is shown in

Figure 7.44

. Low-power RF sig-

nals pass through the tube with little attenuation. Higher-power signals, however,
cause the gas to ionize and present a short circuit to the RF energy.

Figure 7.45

illustrates a balanced duplexer using hybrid junctions and TR tubes.

When the transmitter is on, the TR tubes fire and reflect the RF power to the antenna
port of the input hybrid. During the receive portion of the radar function, signals picked
up by the antenna are passed through the TR tubes and on to the receiver port of the out-
put hybrid.

Newer radar systems often use a ferrite circulator as the duplexer. A TR tube is re-

quired in the receiver line to protect input circuits from transmitter power reflected by
the antenna because of an imperfect match. A four-port circulator generally is used
with a load between the transmitter and receiver ports so that power reflected by the TR
tube is properly terminated.

Pin diode switches also have been used in duplexers to perform the protective

switching function of TR tubes. Pin diodes are more easily applied in coaxial circuitry,

Figure 7.42

Basic concept of a Reggia-Spencer phase shifter.

© 1999 CRC Press LLC

and at lower microwave frequencies. Multiple diodes are used when a single diode can-
not withstand the expected power.

Microwave filters sometimes are used in the transmit path of a radar system to sup-

press spurious radiation, or in the receive signal path to suppress spurious interference.

Figure 7.44

Typical construction of a TR tube.

Figure 7.43

Switched-line phase shifter using

pin diodes.

© 1999 CRC Press LLC

Harmonic filters commonly are used in the transmission chain to absorb harmonic

energy output by the system, preventing it from being radiated or reflected back from
the antenna.

Figure 7.46

shows a filter in which harmonic energy is coupled out

through holes in the walls of the waveguide to matched loads.

Narrowband filters in the receive path, often called preselectors, are built using me-

chanically tuned cavity resonators or electrically tuned TIG resonators. Preselectors
can provide up to 80 dB suppression of signals from other radar transmitters in the same
RF band, but at a different operating frequency.

7.5

High-Power Isolators

The high-power ferrite isolator offers the ability to stabilize impedance, isolate the RF
generator from load discontinuities, eliminate reflections from the load, and absorb
harmonic and intermodulation products. The isolator also can be used to switch be-
tween an antenna or load under full power, or to combine two or more generators into
a common load.

Isolators commonly are used in microwave transmitters at low power to protect the

output stage from reflections. Until recently, however, the insertion loss of the ferrite
made use of isolators impractical at high-power levels (25 kW and above). Ferrite isola-

Figure 7.45

Balanced duplexer circuit using dual TR tubes and two short-slot hybrid

junctions: (

a) transmit mode, (b) receive mode.

(

a)

(

b)

© 1999 CRC Press LLC

tors are now available that can handle 500 kW or more of forward power with less than
0.1 dB of forward power loss.

7.5.1 Theory of Operation

High-power isolators are three-port versions of a family of devices known as
circulators. The circulator derives its name from the fact that a signal applied to one
of the input ports can travel in only one direction, as shown in

Figure 7.47

. The input

port is isolated from the output port. A signal entering port 1 appears only at port 2; it
does not appear at port 3 unless reflected from port 2. An important benefit of this
one-way power transfer is that the input VSWR at port 1 is dependent only on the
VSWR of the load placed at port 3. In most applications, this load is a resistive
(dummy) load that presents a perfect load to the RF generator.

The unidirectional property of the isolator results from magnetization of a ferrite al-

loy inside the device. Through correct polarization of the magnetic field of the ferrite,
RF energy will travel through the element in only one direction (port 1 to 2, port 2 to 3,
and port 3 to 1). Reversing the polarity of the magnetic field makes it possible for RF
flow in the opposite direction. Recent developments in ferrite technology have resulted
in high isolation with low insertion loss.

In the basic design, the ferrite is placed in the center of a Y-junction of three trans-

mission lines, either waveguide or coax. Sections of the material are bonded together to
form a thin cylinder perpendicular to the electric field. Even though the insertion loss is
low, the resulting power dissipated in the cylinder can be as high as 2 percent of the for-

Figure 7.46

Construction of a dissipative waveguide filter.

© 1999 CRC Press LLC

ward power. Special provisions must be made for heat removal. It is efficient heat-re-
moval capability that makes high-power operation possible.

The insertion loss of the ferrite must be kept low so that minimal heat is dissipated.

Values of ferrite loss on the order of 0.05 dB have been produced. This equates to an ef-
ficiency of 98.9 percent. Additional losses from the transmission line and matching
structure contribute slightly to loss. The overall loss is typically less than 0.1 dB, or 98

Figure 7.47

Basic characteristics of a circulator: (

a) operational schematic, (b) distrib-

uted constant circulator, (

c) lump constant circulator. (From [2]. Used with permission.)

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

percent efficiency. The ferrite element in a high-power system is usually water-cooled
in a closed-loop path that uses an external radiator.

The two basic circulator implementations are shown in

Figures 7.47a

and

7.47b

.

These designs consist of Y-shaped conductors sandwiched between magnetized ferrite
discs [2]. The final shape, dimensions, and type of material varies according to fre-
quency of operation, power handling requirements, and the method of coupling. The
distributed constant circulator is the older design; it is a broad-band device, not quite as
efficient in terms of insertion loss and leg-to-leg isolation, and considerably more ex-
pensive to produce. It is useful, however, in applications where broad-band isolation is
required. More commonly today is the lump constant circulator, a less expensive and
more efficient, but narrow-band, design.

At least one filter is always installed directly after an isolator, because the ferrite ma-

terial of the isolator generates harmonic signals. If an ordinary band-pass or band-re-
ject filter is not to be used, a harmonic filter will be needed.

7.5.2 Applications

The high-power isolator permits an RF generator to operate with high performance
and reliability despite a load that is less than optimum. The problems presented by ice
formations on a transmitting antenna provide a convenient example. Ice buildup will
detune an antenna, resulting in reflections back to the transmitter and high VSWR. If
the VSWR is severe enough, transmitter power will have to be reduced to keep the
system on the air. An isolator, however, permits continued operation with no degrada-
tion in signal quality. Power output is affected only to the extent of the reflected en-
ergy, which is dissipated in the resistive load.

A high-power isolator also can be used to provide a stable impedance for devices that

are sensitive to load variations, such as klystrons. This allows the device to be tuned for
optimum performance regardless of the stability of the RF components located after the
isolator.

Figure 7.48

shows the output of a wideband (6 MHz) klystron operating into a

resistive load, and into an antenna system. The power loss is the result of an impedance
difference. The periodicity of the ripple shown in the trace is a function of the distance
of the reflections from the source.

Hot Switch

The circulator can be made to perform a switching function if a short circuit is placed
at the output port. Under this condition, all input power will be reflected back into the
third port. The use of a high-power stub on port 2, therefore, permits redirecting the
output of an RF generator to port 3.

At odd 1/4-wave positions, the stub appears as a high impedance and has no effect on

the output port. At even 1/4-wave positions, the stub appears as a short circuit.
Switching between the antenna and a test load, for example, can be accomplished by
moving the shorting element 1/4 wavelength.

© 1999 CRC Press LLC

Diplexer

An isolator may be configured to combine the aural and visual outputs of a TV trans-
mitter into a single output for the antenna. The approach is shown in

Figure 7.49

. A

single notch cavity at the aural frequency is placed on the visual transmitter output

Figure 7.48

Output of a klystron operating into different loads through a high-power iso-

lator: (

a) resistive load, (b) an antenna system.

(

a)

(

b)

© 1999 CRC Press LLC

(circulator input), and the aural signal is added (as shown). The aural signal will be
routed to the antenna in the same manner as it is reflected (because of the hybrid ac-
tion) in a conventional diplexer.

Multiplexer

A multiplexer can be formed by cascading multiple circulators, as illustrated in

Fig-

ure 7.50

. Filters must be added, as shown. The primary drawback of this approach is

the increased power dissipation that occurs in circulators nearest the antenna.

Figure 7.49

Use of a circulator as a diplexer in TV applications.

Figure 7.50

Using multiple circulators to form a multiplexer.

© 1999 CRC Press LLC

7.6

References

1. Harrison, Cecil, "Passive Filters," in The Electronics Handbook, Jerry C. Whitaker

(ed.), CRC Press, Boca Raton, Fla., pp. 279290, 1996.

2. Surette, Robert A., "Combiners and Combining Networks," in The Electronics

Handbook, Jerry C. Whitaker (ed.), CRC Press, Boca Raton, Fla., pp. 13681381,
1996.

7.7

Bibliography

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1063H, Orland Park, IL, 1982.

Andrew Corporation, "Circular Waveguide: System Planning, Installation and

Tuning," Technical Bulletin 1061H, Orland Park, IL, 1980.

Ben-Dov, O., and C. Plummer, "Doubly Truncated Waveguide," Broadcast Engi-

neering, Intertec Publishing, Overland Park, KS, January 1989.

Benson, K. B., and J. C. Whitaker, Television and Audio Handbook for Technicians

and Engineers, McGraw-Hill, New York, 1989.

Cablewave Systems, "The Broadcaster's Guide to Transmission Line Systems," Tech-

nical Bulletin 21A, North Haven, CT, 1976.

Cablewave Systems, "Rigid Coaxial Transmission Lines," Cablewave Systems Cata-

log 700, North Haven, CT, 1989.

Crutchfield, E. B. (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, 1992.

DeComier, Bill, "Inside FM Multiplexer Systems," Broadcast Engineering, Intertec

Publishing, Overland Park, KS, May 1988.

Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 3rd Ed.,

McGraw-Hill, New York, 1989.

Fink, D., and D. Christiansen (eds.), Electronics Engineers' Handbook, 2nd Ed.,

McGraw-Hill, New York, 1982.

Heymans, Dennis, "Hot Switches and Combiners," Broadcast Engineering, Overland

Park, KS, December 1987.

Jordan, Edward C., Reference Data for Engineers: Radio, Electronics, Computer and

Communications, 7th Ed., Howard W. Sams, Indianapolis, IN, 1985.

Krohe, Gary L., "Using Circular Waveguide," Broadcast Engineering, Intertec Pub-

lishing, Overland Park, KS, May 1986.

Perelman, R., and T. Sullivan, "Selecting Flexible Coaxial Cable," Broadcast Engi-

neering, Intertec Publishing, Overland Park, KS, May 1988.

Stenberg, James T., "Using Super Power Isolators in the Broadcast Plant," Proceed-

ings of the Broadcast Engineering Conference, Society of Broadcast Engineers, In-
dianapolis, IN, 1988.

Terman, F. E., Radio Engineering, 3rd Ed., McGraw-Hill, New York, 1947.
Vaughan, T., and E. Pivit, "High Power Isolator for UHF Television," Proceedings of

the NAB Engineering Conference, National Association of Broadcasters, Washing-
ton, DC, 1989.

Whitaker, Jerry C., G. DeSantis, and C. Paulson, Interconnecting Electronic Systems,

CRC Press, Boca Raton, FL, 1993.

© 1999 CRC Press LLC

Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1990.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

8 Cooling Considerations"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

8

Cooling Considerations

8.1

Introduction

Adequate cooling of the tube envelope and seals is one of the principal factors affect-
ing vacuum tube life [1]. Deteriorating effects increase directly with the temperature
of the tube envelope and ceramic/metal seals. Inadequate cooling is almost certain to
invite premature failure of the device.

Tubes operated at VHF and above are inherently subjected to greater heating action

than tubes operated at lower frequencies. This results directly from the following:

·

The flow of larger RF charging currents into the tube capacitances at higher fre-
quencies

·

Increased dielectric losses at higher frequencies

·

The tendency of electrons to bombard parts of the tube structure other than the
normal elements as the operating frequency is increased

Greater cooling, therefore, is required at higher frequencies. The technical data sheet
for a given tube type specifies the maximum allowable operating temperature. For
forced-air- and water-cooled tubes, the recommended amount of air or water is also
specified in the data sheet. Both the temperature and quantity of coolant should be
measured to be certain that cooling is adequate.

8.1.1 Thermal Properties

In the commonly used model for materials, heat is a form of energy associated with
the position and motion of the molecules, atoms, and ions of the material [2]. The po-
sition is analogous with the state of the material and is potential energy, while the mo-
tion of the molecules, atoms and ions is kinetic energy. Heat added to a material
makes it hotter, and heat withdrawn from a material makes it cooler. Heat energy is
measured in calories (cal), British Thermal Units (Btu), or joules. A calorie is the
amount of energy required to raise the temperature of one gram of water one degree
Celsius (14.5 to 15.5 ºC). A Btu is the unit of energy necessary to raise the tempera-
ture of one pound of water by one degree Fahrenheit. A joule is an equivalent amount

© 1999 CRC Press LLC

of energy equal to the work done when a force of one newton acts through a distance
of one meter.

Temperature is a measure of the average kinetic energy of a substance. It can also be

considered a relative measure of the difference of the heat content between bodies.

Heat capacity is defined as the amount of heat energy required to raise the tempera-

ture of one mole or atom of a material by one ºC without changing the state of the mate-
rial. Thus, it is the ratio of the change in heat energy of a unit mass of a substance to its
change in temperature. The heat capacity, often referred to as thermal capacity, is a
characteristic of a material and is measured in cal/gram per ºC or Btu/lb per ºF.

Specific heat is the ratio of the heat capacity of a material to the heat capacity of a

reference material, usually water. Because the heat capacity of water is one Btu/lb and
one cal/gram, the specific heat is numerically equal to the heat capacity.

Heat transfers through a material by conduction resulting when the energy of atomic

and molecular vibrations is passed to atoms and molecules with lower energy. As heat is
added to a substance, the kinetic energy of the lattice atoms and molecules increases.
This, in turn, causes an expansion of the material that is proportional to the temperature
change, over normal temperature ranges. If a material is restrained from expanding or
contracting during heating and cooling, internal stress is established in the material.

8.1.2 Heal Transfer Mechanisms

The process of heat transfer from one point or medium to another is a result of tem-
perature differences between the two. Thermal energy may be transferred by any of
three basic modes:

·

Conduction

·

Convection

·

Radiation

A related mode is the convection process associated with the change of phase of a
fluid, such as condensation or boiling. A vacuum tube amplifier or oscillator may use
several of these mechanisms to maintain the operating temperature of the device
within specified limits.

Conduction

Heat transfer by conduction in solid materials occurs whenever a hotter region with
more rapidly vibrating molecules transfers a portion of its energy to a cooler region
with less rapidly vibrating molecules. Conductive heat transfer is the most common
form of thermal exchange in electronic equipment. Thermal conductivity for solid
materials used in electronic equipment spans a wide range of values, from excellent
(high conductivity) to poor (low conductivity). Generally speaking, metals are the
best conductors of heat, whereas insulators are the poorest.

Table 8.1

lists the thermal

conductivity of materials commonly used in the construction (and environment) of

© 1999 CRC Press LLC

power vacuum tubes.

Table 8.2

compares the thermal conductivity of various sub-

stances as a percentage of the thermal conductivity of copper.

With regard to vacuum tube devices, conduction is just one of the cooling mecha-

nisms involved in maintaining element temperatures within specified limits. At high
power levels, there is always a secondary mechanism that works to cool the device. For
example, heat generated within the tube elements is carried to the cooling surfaces of
the device by conduction. Convection or radiation then is used to remove heat from the
tube itself.

Thermal conduction has been used successfully to cool vacuum tubes through direct

connection of the anode to an external heat sink, but only at low power levels [1].

Table 8.2 Relative Thermal Conductivity of Various Materials As a Percentage of the
Thermal Conductivity of Copper

Material

Relative Conductivity

Silver

105

Copper

100

Berlox high-purity BeO

62

Aluminum

55

Beryllium

39

Molybdenum

39

Steel

9.1

High-purity alumina

7.7

Steatite

0.9

Mica

0.18

Phenolics, epoxies

0.13

Fluorocarbons

0.05

Material

Btu/(h·ft·°F

W/(m·°C

Silver

242

419

Copper

228

395

Gold

172

298

Beryllia

140

242

Phosphor bronze

30

52

Glass (borosilicate)

0.67

1.67

Mylar

0.11

0.19

Air

0.015

0.026

Table 8.1 Thermal Conductivity of Common Materials

© 1999 CRC Press LLC

Convection

Heat transfer by natural convection occurs as a result of a change in the density of a
fluid (including air), which causes fluid motion. Convective heat transfer between a
heated surface and the surrounding fluid is always accompanied by a mixing of fluid
adjacent to the surface. Power vacuum tube devices relying on convective cooling in-
variably utilize forced air or water passing through a heat-transfer element, typically
the anode in a grid tube or the collector in a microwave tube [1]. This forced convec-
tion provides for a convenient and relatively simple cooling system. In such an ar-
rangement, the temperature gradient is confined to a thin layer of fluid adjacent to the
surface so that the heat flows through a relatively thin boundary layer. In the main
stream outside this layer, isothermal conditions exist.

Radiation

Cooling by radiation is a function of the transfer of energy by electromagnetic wave
propagation. The wavelengths between 0.1 and 100 m are referred to as thermal radi-
ation wavelengths. The ability of a body to radiate thermal energy at any particular
wavelength is a function of the body temperature and the characteristics of the surface
of the radiating material.

Figure 8.1

charts the ability to radiate energy for an ideal ra-

diator, a blackbody, which, by definition, radiates the maximum amount of energy at
any wavelength. Materials that act as perfect radiators are rare. Most materials radiate
energy at a fraction of the maximum possible value. The ratio of the energy radiated
by a given material to that emitted by a blackbody at the same temperature is termed
emissivity.

Table 8.3

lists the emissivity of various common materials.

8.1.3 The Physics of Boiling Water

The Nukiyama curve shown in

Figure 8.2

charts the heat-transfer capability (mea-

sured in watts per square centimeter) of a heated surface, submerged in water at vari-
ous temperatures [1]. The first portion of the curve--zone A--indicates that from 100
to about 108°C, heat transfer is a linear function of the temperature differential be-
tween the hot surface and the water, reaching a maximum of about 5 W/cm

2

at 108°C.

This linear area is known as the convection cooling zone. Boiling takes place in the
heated water at some point away from the surface.

From 108 to 125°C--zone B--heat transfer increases as the fourth power of

T un-

til, at 125°C, it reaches 135 W/cm

2

. This zone is characterized by nucleate boiling; that

is, individual bubbles of vapor are formed at the hot surface, break away, and travel up-
ward through the water to the atmosphere.

Above 125°C, an unstable portion of the Nukiyama curve is observed, where in-

creasing the temperature of the heated surface actually reduces the unit thermal con-
ductivity. At this area--zone C--the vapor partially insulates the heated surface from
the water until a temperature of approximately 225°C is reached on the hot surface. At
this point--called the Leidenfrost point--the surface becomes completely covered
with a sheath of vapor, and all heat transfer is accomplished through this vapor cover.
Thermal conductivity of only 30 W/cm

2

is realized at this region.

© 1999 CRC Press LLC

Table 8.3 Emissivity Characteristics of Common Materials at 80°F

Surface Type

Finish

Emissivity

Metal

Copper (polished)

0.018

Metal

Nickel

0.21

Metal

Silver

0.10

Metal

Gold

0.040.23

Glass

Smooth

0.90.95

Ceramic

Cermet

1

0.58

1

Ceramic containing sintered metal

Figure 8.1 Blackbody energy distribution characteristics.

© 1999 CRC Press LLC

From the Leidenfrost point on through zone D--the film vaporization zone--heat

transfer increases with temperature until at about 1000°C the value of 135 W/cm

2

again

is reached.

The linear plot of the Nukiyama curve indicates that zones A and B are relatively

narrow areas and that a heated surface with unlimited heat capacity will tend to pass
from zone A to zone D in a short time. This irreversible superheating is known as
calefaction. For a cylindrical vacuum tube anode, the passing into total calefaction

Figure 8.2 Nukiyama heat-transfer curves: (

a) logarithmic, (b) linear.

© 1999 CRC Press LLC

would not be tolerated, because any unit heat-transfer density above 135 W/cm

2

would

result in temperatures above 1000°C, well above the safe limits of the tube.

These properties dictate many of the physical parameters of water- and va-

por-cooled power vacuum tubes. For a given dissipation, the anode structure that comes
in contact with the coolant must be sufficient to keep the tube operating within the de-
sired cooling zone of the Nukiyama curve and, in any event, at less than 135 W/cm

2

dis-

sipation on the anode structure.

8.2

Application of Cooling Principles

Excessive dissipation is perhaps the single greatest cause of catastrophic failure in a
power tube [1]. PA tubes used in communications, industrial, and research applica-
tions can be cooled using one of three methods: forced-air, liquid, and vapor-phase
cooling (discussed in Chapter 3). Forced-air cooling is the simplest and most common
method used.

The critical points of almost every PA tube type are the metal-to-ceramic junctions

or seals. At temperatures below 250°C these seals remain secure, but above that tem-
perature, the bonding in the seal may begin to disintegrate. Warping of grid structures
also may occur at temperatures above the maximum operating level of the device. The
result of prolonged overheating is shortened tube life or catastrophic failure. Several
precautions usually are taken to prevent damage to tube seals under normal operating
conditions. Air directors or sections of tubing may be used to provide spot cooling at
critical surface areas of the device. Airflow sensors typically prevent operation of the
RF system in the event of a cooling system failure.

Temperature control is important for vacuum tube operation because the properties

of many of the materials used to build a tube change with increasing temperature. In
some applications, these changes are insignificant. In others, however, such changes
can result in detrimental effects, leading to--in the worst case--catastrophic failure.

Table 8.4

details the variation of electrical and thermal properties with temperature for

various substances.

8.2.1 Forced-Air Cooling Systems

Air cooling is the simplest and most common method of removing waste heat from a
vacuum tube device [1]. The basic configuration is illustrated in

Figure 8.3

. The nor-

mal flow of cooling air is upward, making it consistent with the normal flow of con-
vection currents. In all cases, the socket is an open structure or has adequate vent
holes to allow cooling of the base end of the tube. Cooling air enters at the grid circuit
compartment below the socket through a screened opening, passes through the socket
to cool the base end of the tube, sweeps upward to cool the envelope, and enters the
output circuit compartment.

The output compartment is provided with a mesh-covered opening that permits the

air to vent out readily. High-power tubes typically include a chimney to direct airflow,
as shown in

Figure 8.4

. For a forced-air design, a suitable fan or blower is used to pres-

surize the compartment below the tube. No holes should be provided for the passage of

© 1999 CRC Press LLC

air from the lower to the upper compartment other than the passages through the socket
and tube base. A certain amount of pressure must be built up to force the proper amount
of air through the socket to cool the anode.

Attention must be given to airflow efficiency and turbulence in the design of a cool-

ing system. Consider the case shown in

Figure 8.5

. Improper layout has resulted in inef-

ficient movement of air because of circulating thermal currents. Anode cooling will be
insufficient in this case.

The cooling arrangements illustrated in

Figures 8.3

and

8.4

provide for the uniform

passage of cooling air over the tube base and anode. An arrangement that forces cooling
air transversely across the tube base and/or anode will not provide the same cooling ef-
fectiveness and may result in hot spots on the device.

In many cases, packaged air system sockets and chimneys are designed specifically

for a tube or family of tube types. The technical data sheet specifies the recommended
socketing for adequate cooling. The tube data sheet also will specify the back pressure,
in inches of water, and the cubic feet per minute required for adequate cooling. In an ac-
tual application, the back pressure may be measured by means of a simple manometer.
This device consists of a U-shaped glass tube partially filled with water, as shown in

Figure 8.6

.

Table 8.4 Variation of Electrical and Thermal Properties of Common Insulators As a
Function of Temperature

Parameters

20°C

120°C

260°C

400°C

538°C

Thermal
conductivity

1

99.5%
BeO

140

120

65

50

40

99.5%
Al

2

O

3

20

17

12

7.5

6

95.0%
Al

2

O

3

13.5

Glass

0.3

Power
dissipation

2

BeO

2.4

2.1

1.1

0.9

0.7

Electrical
resistivity

3

BeO

10

16

10

14

5

×10

12

10

12

10

11

Al

2

O

3

10

14

10

14

10

12

10

12

10

11

Glass

10

12

10

10

10

8

10

6

Dielectric
constant

4

BeO

6.57

6.64

6.75

6.90

7.05

Al

2

O

3

9.4

9.5

9.6

9.7

9.8

Loss tangent

4

BeO

0.00044

0.00040 0.00040

0.00049

0.00080

1

Heat transfer in Btu/ft

2

/hr/°F

2

Dissipation in W/cm/°C

3

Resistivity in

-cm

4

At 8.5 GHz

© 1999 CRC Press LLC

Figure 8.3 Cooling system design for a power grid tube.

Figure 8.4 The use of a chimney to improve cooling of a power grid tube.

© 1999 CRC Press LLC

Figure 8.5 A poorly designed cooling system in which circulating air in the output com-
partment reduces the effectiveness of the heat-removal system.

Figure 8.6 A manometer, used to measure air pressure.

© 1999 CRC Press LLC

Cooling Airflow Data

Power tube manufacturers typically outline minimum cooling airflow requirements
for large external-anode tubes in the form of one or more charts [1].

Figure 8.7

plots

P

t

/

T as a function of A

m

, where P

t

= total power dissipated in watts,

T = tube tem-

perature rise in degrees Celsius, and A

m

= mass airflow rate in pounds of air per min-

ute. This type of graph is used in calculating the cooling requirements of a vacuum
tube device. The graph applies to a specified tube and socket-chimney combination;
furthermore, the direction of airflow is specified. When reverse airflow (from anode
to base) is used, the cooling requirements are sharply increased because the air ap-
plied to the base seals already will have been heated by its passage through the anode
cooler, losing much of its cooling effectiveness.

To use the type of graph shown in

Figure 8.7

, first determine the minimum cooling

requirements using the following steps:

1. Calculate the total power dissipated (P

t

) by adding all of the power dissipated by

the tube during operation in a given installation. This value includes plate and fila-

Figure 8.7 Cooling airflow requirements for a power grid tube. (

Data courtesy of

Varian.)

© 1999 CRC Press LLC

ment dissipations, plus maximum anticipated grid and screen dissipations (as ap-
plicable).

2. Calculate the tube temperature rise (

T) by taking the difference between the

maximum rated tube temperature specified in the appropriate data sheet and the
maximum expected air-inlet temperature.

3. To convert the mass airflow rate M (pounds per minute) to volumetric airflow
rate Q (cubic feet per minute) at 25°C and at sea level, divide the mass airflow rate
by the density of air at 25°C and 29.9 in mercury.

4. The curve on the right side of the graph is the pressure drop (

P) in inches of

water across the tube and its specified socket-chimney combination. This value is
valid at 25°C at sea level only.

To adjust the 25°C sea-level laboratory test conditions to any other atmospheric

(socket-inlet) condition, multiply both the Q and

P values by the ratio of the labora-

tory standard density (0.074 lb/ft

3

; 25°C at sea level) to the density at the new socket-in-

let condition.

A shorter method may be used to approximately correct the 25°C sea level require-

ments to both a different temperature and/or barometric socket-inlet condition. These
corrections are made by multiplying the Q and

P values by the appropriate correction

factors listed in

Table 8.5

.

Figure 8.8

is a graph of the combined correction factors that can be applied to the

25°C sea-level information for land-based installations located at elevations up to
10,000 ft, and for socket-inlet air temperatures between 10 and 50°C.

Figure 8.9

pro-

vides a method to convert the mass airflow rate in pounds per minute into volumetric
airflow rate (cfm) at 25°C and sea level.

Good engineering practice must be used when applying altitude and temperature

corrections to the 25°C sea-level cooling requirement for airborne applications. Al-
though the air outside the aircraft may be very cold at high altitudes, the air actually en-
tering the tube socket may be many degrees warmer. This inlet temperature (and pres-
sure) is affected by the installation design (compressed, ram, static, or recirculating air
in a pressurized heat exchanger).

Blower Selection

In the previous section, a method of determining minimum air cooling requirements
for external-anode tubes was described, pertaining to any altitude and air temperature
[1]. Because most blower manufacturers furnish catalog data on their products in the
form of volumetric airflow (Q, cfm) vs. operating back pressure (

P, inches of water)

for sea level conditions only, the information gained from the foregoing procedure
cannot be compared directly with data furnished by manufacturers for the purpose of
selecting the proper device. The following method is recommended for use in select-
ing a blower for applications above sea level from existing blower catalog data:

© 1999 CRC Press LLC

1. Determine the Q and

P requirements for the tube socket-chimney combination

for an ambient air temperature of 25°C at sea level. Include the estimated

P of

ducting and filters in the system.

2. Determine the corrected Q and

P system requirements for the actual inlet tem-

perature and altitude conditions by multiplying by the correction factor shown in

Figure 8.7

.

3. Multiply the

P--but not the Q--requirement by the correction factor cited in

step 2.

4. Use the corrected Q factor and doubly-corrected

P value to select a blower

from the manufacturer's published sea-level curves. Although this blower will
overcool the tube at sea level when operated in an ambient temperature of 25°C, it
will provide adequate cooling at the actual inlet temperature and at high-altitude
conditions.

Table 8.5 Correction Factors for

Q and

P

Socket-Inlet Air Temperature (°C)

Q and

P Correction Factor

0

0.917

5

0.933

10

0.950

15

0.967

20

0.983

25

1.000

30

1.017

35

1.034

40

1.051

45

1.067

50

1.084

Socket-Inlet Air Pressure

1

Altitude (ft) Q and

P Correction Factor

29.92

0

1.00

24.90

5000

1.20

20.58

10,000

1.46

16.89

15,000

1.77

13.75

20,000

2.17

11.10

25,000

2.69

8.89

30,000

3.37

7.04

35,000

4.25

1

Pressure in inches of mercury

© 1999 CRC Press LLC

8.2.2 Water Cooling

A water-cooled tube depends upon an adequate flow of fluid to remove heat from the
device and transport it to an external heat sink [1]. The recommended flow as speci-
fied in the technical data sheet should be maintained at all times when the tube is in
operation. Inadequate water flow at high temperature may cause the formation of
steam bubbles at the anode surface where the water is in direct contact with the anode.
This condition can contribute to premature tube failure.

Circulating water can remove about 1.0 kW/cm

2

of effective internal-anode area. In

practice, the temperature of water leaving the tube is limited to 70°C to preclude the
possibility of spot boiling. The water then is passed through a heat exchanger where it is
cooled to 30 to 40°C before being pumped over the tube anode again.

Cooling System Design

A liquid cooling system consists of the following principal components:

·

A source of coolant

·

Circulation pump

Figure 8.8 Combined correction factors for land-based tube applications.

© 1999 CRC Press LLC

·

Heat exchanger

·

Coolant purification loop

·

Various connection pipes, valves, and gauges

·

Flow interlocking devices (required to ensure coolant flow anytime the equip-
ment is energized)

Such a system is shown schematically in

Figure 8.10

. In most cases the liquid coolant

will be water, but if there is a danger of freezing, it will be necessary to use an anti-
freeze solution such as ethylene glycol. In these cases, coolant flow must be increased
or plate dissipation reduced to compensate for the poorer heat capacity of the ethylene
glycol solution. A mixture of 60 percent ethylene glycol to 40 percent water by weight
will be about 75 percent as efficient as pure water at 25°C. Regardless of the choice of
liquid, the system volume must be maintained above the minimum required to ensure
proper cooling of the vacuum tube(s).

The main circulation pump must be of sufficient size to ensure necessary flow and

pressure as specified on the tube data sheet. Care must be taken when connecting the
coolant lines to the tube to be certain that flow is in the direction specified. Improper di-
rection of flow may result in inadequate cooling as well as excessive pressure and possi-

Figure 8.9 Conversion of mass airflow rate to volumetric airflow rate.

© 1999 CRC Press LLC

ble mechanical deformation of the anode. Precautions should be taken during system
design to protect against water hammering of the anode. A filter screen of at least 60
mesh usually is installed in the pump outlet line to trap any circulating debris that might
clog coolant passages within the tube.

The heat exchanger system is sized to maintain the outlet temperature such that the

outlet water from the tube at full plate dissipation does not exceed 70°C. Filament or
grid coolant courses may be connected in parallel or series with the main supply as long
as the maximum outlet temperature is not exceeded.

Valves and pressure meters are installed on the inlet lines to the tube to permit ad-

justment of flow and measurement of pressure drop, respectively. A pressure meter and
check valve are employed in the outlet line. In addition, a flow meter--sized in accor-
dance with the tube data sheet--and a thermometer are included in the outlet line of
each coolant course. These flow meters are equipped with automatic interlock
switches, wired into the system electrical controls, so that the tube will be completely
deenergized in the event of a loss of coolant flow in any one of the coolant passages. In
some tubes, filament power alone is sufficient to damage the tube structure in the ab-
sence of proper water flow.

The lines connecting the plumbing system to the inlet and outlet ports of the tube are

made of a flexible insulating material configured so as to avoid excessive strain on the
tube flanges. Polypropylene tubing is the best choice for this service, but chlorinated
polyvinyl chloride (CPVC) pipe, which is stronger, also is acceptable. Reinforced poly-

Figure 8.10 Functional schematic of a water cooling system for a high-power amplifier
or oscillator.

© 1999 CRC Press LLC

propylene, such as Nylobraid, is expensive, but excellent in this application. The cool-
ant lines must be of sufficient length to keep electrical leakage below 4 mA at full oper-
ating power.

The hoses are coiled or otherwise supported so that they do not contact each other or

any conducting surface between the high-voltage end and ground. Conducting hose
barbs, connected to ground, are provided at the low-potential end so that the insulating
column of water is broken and grounded at the point where it exits the equipment cabi-
net.

All metallic components within the water system, including the pump, must be of

copper, stainless steel, or unleaded brass or bronze; copper or stainless steel is pre-
ferred. If brass or bronze is used, some zinc may be leached out of the metal into the sys-
tem. Any other material, such as iron or cold rolled steel, will grossly contaminate the
water.

Even if the cooling system is constructed using the recommended materials, and is

filled with distilled or deionized water, the solubility of the metals, carbon dioxide, and
free oxygen in the water make the use of a coolant purification (regeneration) loop es-
sential. The integration of the purification loop into the overall cooling system is shown
in

Figure 8.10

. The regeneration loop typically taps 5 to 10 percent of the total cooling

system capacity, circulating it through oxygen scavenging and deionization beds, and
submicron filters before returning it into the main system. Theoretically, such a purifi-
cation loop can process water to 18 M

-cm resistivity at 25ºC. In practice, resistivity

will be somewhat below this value.

Figure 8.11

shows a typical purification loop configuration. Packaged systems such

as the one illustrated are available from a number of manufacturers. In general, such

Figure 8.11 Typical configuration of a water purification loop.

© 1999 CRC Press LLC

systems consist of replaceable cartridges that perform the filtering, ion exchange, and
organic-solid-removal functions. The system usually will include flow and pressure
gauges and valves, and conductivity cells for continuous evaluation of the condition of
the water and filters.

Water Purity and Resistivity

The purity of the cooling water is an important operating parameter for any wa-
ter-cooled amplifier or oscillator. The specific resistivity must be maintained at 1
M

-cm minimum at 25ºC. Distilled or deionized water should be used and the purity

and flow protection periodically checked to ensure against degradation. Oxygen and
carbon dioxide in the coolant will form copper oxide, reducing cooling efficiency,
and electrolysis may destroy the coolant passages. In addition, a filter screen should
be installed in the tube inlet line to trap any circulating debris that might clog coolant
passages within the tube.

After normal operation for an extended period, the cooling system should be capable

of holding 3 to 4 M

-cm until the filter beds become contaminated and must be re-

placed. The need to replace the filter bed resins is indicated when the purification loop
output water falls below 5 M

-cm. Although the resistivity measurement is not a test

Figure 8.12 The effect of temperature on the resistivity of ultrapure water.

© 1999 CRC Press LLC

for free oxygen in the coolant, the oxygen filter bed always should be replaced when the
deionizing bed is replaced.

The resistivity of the coolant also is affected by the temperature of the water. The

temperature-dependence of the resistivity of pure water is charted in

Figure 8.12

.

It is recommended that the coolant water be circulated at all times. This procedure

provides the following benefits:

·

Maintains high resistivity

·

Reduces bacteria growth

·

Minimizes oxidation resulting from coolant stagnation

If it is undesirable to circulate the coolant at the regular rate when the tube is
deenergized, a secondary circulating pump can be used to move the coolant at a lower
rate to purge any air that might enter the system and to prevent stagnation. Recom-
mended minimum circulation rates within the coolant lines are as follows:

·

2 m/s during normal operation

·

30 cm/s during standby mode

It is recommended that a circulation rate of 10 m/s be established within the tube it-
self. When it is necessary to turn the coolant flow completely off, the cooling system
should be restarted, and allowed to return to a minimum of 1 M

-cm resistivity before

energizing the tube.

The regeneration loop is typically capable of maintaining the cooling system such

that the following maximum contaminant levels are not exceeded:

·

Copper: 0.05 ppm by weight

·

Oxygen: 0.5 ppm by weight

·

CO

2

: 0.5 ppm by weight

·

Total solids: 3 ppm by weight

These parameters represent maximum levels; if the precautions outlined in this sec-
tion are taken, actual levels will be considerably lower.

If the cooling system water temperature is allowed to reach 50°C, it will be neces-

sary to use cartridges in the coolant regeneration loop that are designed to operate at el-
evated temperatures. Most ordinary cartridges will decompose in high-temperature
service.

Condensation

The temperature of the input cooling water is an important consideration for proper
operation of the cooling system [1]. If the air is humid and the cooling water is cold,
condensation will accumulate on the surfaces of pipes, tube jackets, and other parts
carrying water. This condensation may decrease the surface leakage resistance, or
drops of water may fall on electric components, causing erratic operation or system

© 1999 CRC Press LLC

failure. Some means is necessary, therefore, to control the temperature of the incom-
ing water to keep it above the dew point. Such control is rather easy in a closed cool-
ing system, but in a system that employs tap water and drains the exhaust water into a
sewer, control is difficult.

Preventive Maintenance

Through electrolysis and scale formation, hard water may cause a gradual constric-
tion of some parts of the water system [1]. Therefore, water flow and plumbing fit-
tings must be inspected regularly. The fittings on the positive-potential end of an insu-
lating section of hose or ceramic water coil or column are particularly subject to cor-
rosion or electrolysis unless they have protective targets. The target elements should
be checked periodically and replaced when they have disintegrated.

To extend the life of the resin beds in the purification loop, all the coolant lines

should be flushed with a nonsudsing detergent and a citric acid solution,

1

then rinsed re-

peatedly with tap water before initially connecting the resin beds and filling with dis-
tilled or filtered deionized water. It is also good practice to sterilize the coolant lines
with a chlorine solution

2

before filling to prevent algae and/or bacteria growth.

8.2.3 Vapor-Phase Cooling

Vapor-phase cooling offers several advantages over conventional water cooling sys-
tems by exploiting the latent heat of the evaporation of water [1]. Raising the tempera-
ture of 1 g of water from 40 to 70°C (as in a water system) requires 30 calories of en-
ergy. Transforming 1 g of water at 100°C to steam vapor requires 540 calories. In a
vapor cooling system, then, a given quantity of water will remove nearly 20 times as
much energy as in a water cooling system. Power densities as high as 135 W/cm

2

of

effective internal-anode surface can be attained through vapor cooling.

A typical vapor-phase installation consists of a tube with a specially designed anode

immersed in a boiler filled with distilled water. When power is applied to the tube, an-
ode dissipation heats the water to 100°C; further applied energy causes the water to boil
and be converted into steam vapor. The vapor is passed through a condenser where it
gives up its energy and is converted back into the liquid state. This condensate is then
returned to the boiler, completing the cycle.

To achieve the most efficient heat transfer, the anode must be structured to provide

for optimum contact with the water in the boiler.

Figure 8.13

shows several examples.

Because the boiler usually is at a high potential relative to ground, it must be insu-

lated from the rest of the system. The boiler typically is mounted on insulators, and the
steam and water connections are made through insulated tubing. High-voltage standoff

1

Cirtic acid solution mixtures vary; consult tube manufacturer.

2

Chlorine solution: sodium hypochlorite bleach added in an amount sufficient to give the odor
of chlorine in the circulating water.

© 1999 CRC Press LLC

is more easily accomplished in a vapor-phase system than in a water-cooled system be-
cause of the following factors:

·

There is a minimum of contamination because the water is constantly being
redistilled.

·

There is inherently higher resistance in the system because of the lower wa-
ter-flow rate. The water inlet line is of a smaller diameter, resulting in greater re-
sistance.

In a practical system, a 2-ft section of insulating tubing on the inlet and outlet ports is
capable of 20 kV standoff.

Because of the effects of thermosiphoning, natural circulation of the water elimi-

nates the need for a pump in most systems.

The dramatic increase in heat absorption that results from converting hot water to

steam is repeated in reverse in the condenser. As a result, a condenser of much smaller
thermal capacity is required for a vapor-phase system as opposed to a water-cooled sys-
tem. For example, in a practical water-cooled system, water enters the heat exchanger at
70°C and exits at 40°C; the mean temperature is 55°C. For an ambient external temper-
ature of 25°C, the mean differential between the water and the heat sink (air) is 30°C.
The greater the differential, the more heat transferred. In a practical va-
por-phase-cooled system, water enters the heat exchanger as steam at 100°C and exits
as water at 100°C; the mean temperature is 100°C. For an ambient external temperature
of 25°C, the mean differential between the vapor/water and the heat sink is 75°C. In this
example, the vapor-phase condenser is nearly three times more efficient than its wa-
ter-cooled counterpart.

Note that the condenser in a vapor-phase system may use either air or water as a heat

sink. The water-cooled condenser provides for isolation of the PA tube cooling system

Figure 8.13 Vapor-phase-cooled tubes removed from their companion boilers. (

Cour-

tesy of Varian/Eimac.)

© 1999 CRC Press LLC

from the outside world. This approach may be necessary because of high voltage or
safety reasons. In such a system, the water is considered a secondary coolant.

Where air-cooled condensers are preferred, the higher thermal gradient can be ex-

ploited in reducing the size of condenser equipment and in lowering the blower horse-
power requirement. In some instances where sufficient area is available, natural con-
vection alone is used to cool the steam condensers, resulting in complete elimination of
the condenser blower.

A typical vapor-phase cooling system is shown in

Figure 8.14

. It consists of the fol-

lowing elements:

·

Power tube

·

Boiler

·

Condenser

·

Insulated tubing

·

Control box

·

Reservoir

·

Associated plumbing

Anode Design

The success of vapor-phase cooling is dependent on anode and boiler designs that al-
low a tube to operate at a temperature that results in maximum heat dissipation [1].

Figure 8.14 Vapor-phase cooling system for a power tube.

© 1999 CRC Press LLC

The most common approach has been to incorporate thick vertical fins on the exterior
of the anode to achieve a radial temperature gradient on the surfaces submerged in
water. In this way, a hot spot does not cause instantaneous runaway of the tube; only
the average fin temperature increases, and this merely shifts the temperature range to
a somewhat higher level. The temperatures of the fins typically vary from approxi-
mately 110°C at the tip to about 180°C at the root. With an average overall fin temper-
ature of approximately 115°C, for example, the average transferred heat flux is on the
order of 60 W/cm

2

.

When operating at low dissipation levels, boiling takes place at the root of the fin.

Increasing power density causes this boiling area to move out toward the end of the fin
until, at rated dissipation levels, boiling takes place on the outer half of the fins. Good
design dictates that the anode fin outer edge always remain at less than 125°C.

Anode shape is also important in assuring good cooling by breaking up the sheath of

vapor that might tend to form at the surfaces. To a point, the more complicated the
shape, the more efficient the anode cooler. Horizontal slots often are milled into the
vertical fins to provide more area and to break up bubbles. The more vigorous the boil-
ing action, the lower the possibility that an insulating vapor sheath will form. One de-
sign--known as the "pineapple"--carries this idea to the extreme, and incorporates
square "knobs" all around the anode to provide dozens of heat-radiating surfaces. The
more conventional finned anode, however, is generally adequate and less costly to fab-
ricate.

The holed anode is another popular design for lower power densities (up to 100

W/cm

2

). In this design, a heavy cylindrical anode is made with vertical holes in the out-

side wall. Vapor, formed by the boiling of water within the holes, is siphoned upward
through the holes to the top of the boiler. This type of cooler always operates below the
calefaction point and is used in smaller tube types. The concept is illustrated in

Figure

8.15

.

Boiler

The boiler supports the power tube and contains the water used for cooling [1]. In ad-
dition, it acts as the high-voltage anode connector. The boiler should be mounted so
that the axis of the tube is vertical. For effective cooling, tilt should be limited to less
than 2° to ensure that the anode is covered with water and that the steam outlet is clear.

Figure 8.16

shows a 4CV35,000A tetrode mounted in a boiler.

The anode flange of the tube must seal securely against the O-ring provided on the

boiler. A locking flange presses the anode flange against the O-ring for a vapor-tight
seal. The steam outlet at the top of the steam separation chamber on the boiler and the
water inlet at the bottom of the boiler are equipped with fittings for attaching the Py-
rex-insulated tubing. A target to inhibit electrolytic action is provided in the inlet water
fitting.

In most cases the boiler is at a high potential relative to ground. Therefore, it must be

electrically insulated from the rest of the system. The boiler is mounted on insulators,
and the steam and water connections are made through insulating tubing. Boilers can be

© 1999 CRC Press LLC

constructed with provisions for mounting multiple tubes in parallel. In such an arrange-
ment, a single water inlet and steam outlet fitting typically would be used.

Common anode/boiler hardware is well suited to RF amplifiers operating up into the

HF band, which typically use conventional tuned circuits. Operation in systems at
higher frequencies, however, becomes more complicated than water cooling, and cer-

Figure 8.15 Cross section of a

holed anode design for vapor-phase cooling.

Figure 8.16 A 4CV35,000A tetrode mounted
in its boiler assembly. (

Courtesy of Varian.)

© 1999 CRC Press LLC

tainly more complicated than air cooling. The cooling system in an air-cooled RF gen-
erator does not interfere with the proper operation of the cavity; it is invisible to the cav-
ity. Water cooling introduces some compromises and/or adjustments in the cavity,
which can be readily overcome. Vapor-cooled tubes, however, require special design at-
tention for successful use.

Insulating Tubing

The length of the steam and water insulating lines varies with individual installation
requirements, but will always be shorter than would be needed in a circulating water
system [1]. The length of the insulating tubing is determined by the following consid-
erations:

·

The voltage to be applied to the tube anode

·

Purity of the water

·

Volume of returned cooling water

Control Box

The function of the control box is to monitor and adjust the water level in the boiler
[1]. A typical box is shown in

Figure 8.17

. The control box, an airtight vessel contain-

ing an overflow siphon and two float switches, also serves as a partial reservoir. When
the water level drops approximately 1/4 in below the recommended level, the first
switch is closed. This signal may be used, at the same time, to activate a solenoid-con-
trolled water valve to admit more distilled water from an external reservoir; it also
may actuate a warning alarm.

Figure 8.17

Control box for a va-

por-phase cooling system. (

Courtesy

of Varian.)

© 1999 CRC Press LLC

The second float switch is operated if the water level drops approximately 1/2 in be-

low the optimum level. This would be tantamount to a water system failure; the switch
would be used to open the control circuit interlocks and remove power from the tube.

For the control box to perform its protective function properly, the water-level mark

must be precisely in line with the water-level mark on the boiler. For electrical reasons,
the control box generally is mounted some distance from the boiler, and therefore level-
ing of the two components must be carefully checked during installation.

Figure 8.18

shows a cutaway drawing of a classic boiler and tube combination.

Figure 8.19

is a

cutaway drawing of a control box, showing the position of the float switches and the
overflow pipe.

During extended operation, some quantity of water and steam being circulated

through the condenser will be lost. The amount is dependent on the size of the system.
The water level in the boiler, therefore, will gradually drop. The use of the control box
as a reservoir minimizes this effect. In large or multiple-tube installations, an auxiliary
reservoir is connected to the control box to increase the ratio of stored water to circulat-
ing water and steam. Where it may be necessary to operate multiple tubes at different
physical elevations, individual control boxes are required. A multiple-tube system is
shown in

Figure 8.20

.

Equalizer Line

For the control box to duplicate the same pressure conditions that exist in the boiler,
the vapor-phase system must be fitted with an equalizer line [1]. This length of tubing
connects the steam side of the system with the top of the control box. As partial steam

Figure 8.18 Cutaway view of a boiler and tube combination.

© 1999 CRC Press LLC

pressure begins to build up in the boiler, the equalizer line allows this same pressure
to appear in the control box. Although the steam pressure is low, less than 0.5 psi
above atmosphere, an error in the operation of the control box would be introduced
unless the vessel were equalized.

The fitting used to connect the equalizer line to the steam outlet tube must be con-

structed to prevent a venturi effect from developing because of the velocity of the vapor.
This is best accomplished by directing an elbow within the adapter fitting toward the
boiler, as shown in

Figure 8.21

.

Condenser

Both air-cooled and water-cooled condensers are available for vapor cooling systems
[1]. Choose condensers with good reserve capabilities and low pressure drop. Air-
and water-cooled condensers may be mounted in any position, provided they allow
the condensed water to flow freely by gravity to the boiler return line. Water must not
be allowed to stand in the condenser where it might cause back pressure to the enter-
ing steam.

The condenser should be mounted above the level of the boiler(s) so that water will

drain from it to the boiler return line. Where it is necessary to mount the condenser at a
lower physical level than the system water level, an auxiliary pump must be used to re-

Figure 8.19 Cutaway view of a control box.

© 1999 CRC Press LLC

Figure 8.20 Typical four-tube vapor cooling system with a common water supply.

©

1
999
CRC Press LLC

turn water to the boiler. This arrangement is recommended for the "steam-out-the-bot-
tom" boiler system, as illustrated in

Figure 8.22

.

Pressure Interlock

Most tube manufacturers recommend the use of a steam pressure interlock switch on
the steam or inlet side of the condenser [1]. This switch, set to about 0.5 lbs/in

2

, is

used as a power interlock that senses abnormal steam pressure resulting from con-
strictions in the condenser or piping.

Piping

Piping should be of copper or glass throughout the system [1]. The steam piping
should be the same diameter as the Pyrex tube from the boiler. The size is dependent
on the power level and the volume of generated steam, and it will range from 1-1/4 in
at the 8 kW level to 6 in for the 250 kW level of dissipation. The steam path should be
as direct as practical and must be sloped to prevent condensate from collecting at a
low point where it might cause back pressure. All low spots should be drained back to
the inlet water line.

The diameter of water return piping from the condenser to the control box will vary

from 3/4 to 1-3/4 in, depending again on the power level. This tubing should be the
same diameter as the boiler inlet water fitting. It also should be sloped so that water or
vapor pockets are not allowed to form, and it must allow the condensate to return by
gravity to the control box and the boiler. A vent to air on the outlet side of the condenser
should be incorporated to maintain the water side of the system at atmospheric pres-
sure. Provisions for draining the distilled water should be made at the lowest physical
level of the system.

Figure 8.21 Cutaway view of a pressure
equalizer fitting.

© 1999 CRC Press LLC

Figure 8.22 Typical four-tube vapor cooling system using "steam-out-the-bottom" boilers.

©

1
999
CRC Press LLC

The equalizer line also should be sloped from the adapter fitting on the steam line to

the top of the control box. This will allow the condensate to return to the control box.

Automatic Refilling System

Figure 8.23

shows a typical vapor cooling system with provisions to allow additional

water into the control box [1]. An auxiliary reservoir is connected through a sole-
noid-operated water valve to the controller. When water loss resulting from evapora-
tion causes the water level in the boiler and the control box to drop about

1

4

in below

normal, the first float switch in the control box closes and actuates the solenoid-con-
trolled valve to permit makeup water to enter the system. When the proper level is re-
stored, the switch opens, the valve closes, and the flow of makeup water is stopped.

Alternative Vapor Cooling Systems

The systems described thus far are "classic" designs in which a separate tube, boiler,
condenser, and level control box are used [1]. Variations on this basic scheme are nu-
merous. One such alternative system, offered for use with large power tubes, utilizes a
"steam-out-the-bottom" boiler. This configuration makes it possible to keep the
steam and water systems, plus the plumbing, below the tubes. This configuration of-
ten simplifies the electrical design of the amplifier or oscillator.

Figure 8.24

shows a

typical boiler used in this cooling technique.

A small water pump circulates a continuous flow of water over a weir--or baffle--in

the boiler, maintaining a constant water level. Generated steam is forced under slight

Figure 8.23 Vapor cooling system incorporating a reservoir of distilled water.

© 1999 CRC Press LLC

pressure out the bottom of the boiler, through an insulator tube in the condenser. Water
from the condenser flows into the control box before being pumped back into the boiler.
Protective devices include a water-flow interlock and water-level switch in the control
box to ensure an adequate supply of coolant.

Maintenance

Maintenance problems associated with circulating water systems are practically elim-
inated in a vapor cooling system [1]. As mentioned previously, systems can be de-
signed to eliminate all rotating machinery or moving parts. Good engineering practice
does, however, dictate periodic maintenance. The glass insulator tubes should be in-
spected occasionally to be sure they contain no deposits that might cause high-voltage
flashover. Water conductivity can be checked by measuring the dc resistance, as in a
typical circulating water system. The water should be replaced if the dc resistance
drops below 1.0 M

-cm.

In practice, a vapor-cooled system will remain cleaner longer than a water-cooled

system. In the vapor-cooled boiler, the water is continuously being redistilled, and only
pure water is introduced at the bottom of the boiler. Any contaminants will tend to re-
main in the boiler itself, where they can be removed easily. The periods between equip-
ment shutdown for draining and cleaning will be at least twice as long for a vapor cool-
ing system because of this inherent self-cleaning action.

Each time a tube is removed or replaced, the rubber O-ring between the boiler and

the tube should be inspected and, if necessary, replaced. At the same time, the inside of
the boiler and the control box can be inspected and cleaned as needed. The electrolytic

Figure 8.24 Cutaway view of a "steam-out-the-bottom" boiler.

© 1999 CRC Press LLC

target should be replaced whenever its metallic end is no longer visible in the inlet wa-
ter line.

8.2.4 Temperature Measurements

Before completing any design employing power tubes, check temperatures in critical
areas, such as metal/ceramic seals and (except for water- and vapor-cooled types) the
anode just above and below the cooling fins [1]. Another critical area on many me-
dium and large tubes of coaxial design is the central part of the base, which is often re-
cessed and may need special cooling provisions. Electrical contact to the tube termi-
nals is usually by copper-beryllium collets, which should not exceed a temperature of
150°C for any extended period of time. A number of manufacturers make tempera-
ture-sensitive paints that are useful for such measurements.

3

"Crayon-type" indicators

are also available, in a wide range of temperatures. Conventional temperature measur-
ing-techniques may be affected by the radio frequency energy concentrated at the tube
and nearby components. Furthermore, because all power tubes operate at high volt-
age, there is danger of electric shock with directly connected devices.

When a temperature-sensitive paint or crayon is used for measurement, it must be

remembered that many of the areas of concern are vacuum seals involving relatively
thin metal. It is important, therefore, that the indicator material be noncorrosive.

Considering the importance of tube element temperatures, every design must be

evaluated carefully. All power tubes carry an absolute maximum temperature rating for
the seals and envelope, and in the case of an external-anode forced-air-cooled tube, for
the anode core itself. Operation above the rated maximum can cause early tube failure.
Where long life and consistent performance are important factors, it is normally desir-
able to maintain tube temperatures comfortably below the rated maximum.

The equipment operator has the ultimate responsibility of making sure that the

power tubes used in the system are operating within specified ratings. Most current
tube designs use a ceramic envelope and an externally mounted anode or collector. The
only visible sign of excessive temperature may be the cosmetic plating beginning to
darken or turn black (oxidize). By then, significant damage may have occurred.

"Crayon" Temperature Measurement

When a crystalline solid is heated, a temperature will be reached at which the solid
changes sharply to a liquid [1]. This melting point has a definite, reproducible value
that is virtually unaffected by ambient conditions that may cause errors in other tem-
perature-sensing methods. For example, fusible temperature indicators are practical
in induction heating and in the presence of static electricity or ionized air around elec-
tric equipment, where electronic means of measuring temperatures often function er-
ratically.

3

Tempilaq is one such product (Tempil Division, Big Three Industrial Gas and Equipment
Co., South Plainfield, NJ).

© 1999 CRC Press LLC

The most popular type of fusible indicator is a temperature-sensitive stick closely

resembling a crayon, with a calibrated melting point. These crayon-type indicators are
made in 100 or so different specified temperature ratings in the range of 100 to 2500°F.
Typically, each crayon has a temperature-indicating accuracy within 1 percent of its rat-
ing. The workpiece to be tested is marked with the crayon. When the workpiece attains
the predetermined melting point of the crayon, the mark instantly changes from a solid
to a liquid phase (liquefies), indicating that the workpiece has reached that tempera-
ture.

Under certain circumstances, premarking with crayon is not practical. This is the

case if one or more of the following is true:

·

A prolonged heating period is experienced.

·

The surface is highly polished and does not readily accept a crayon mark.

·

The material being marked is one that gradually absorbs the liquid phase of the
crayon.

Note that a melted mark, upon cooling, will not solidify at the exact temperature at

which it melted. Solidification of a melted crayon mark, therefore, cannot be relied on
for exact temperature indication.

Phase-Change Fluid

A phase-changing fluid, or fusible temperature-indicating lacquer, offers the greatest
flexibility in temperature measurement of a power tube [1]. The lacquer-type fluid
contains a solid material of calibrated melting point suspended in an inert, volatile,
nonflammable vehicle. As with crayon-type indicators, there are approximately 100
different temperature ratings, covering a range from 100 to 2500°F. Accuracy is typi-
cally within 1 percent of the specified value.

The lacquer is supplied in the proper consistency for brushing. If spraying or dip-

ping is desirable as the mode of application, a thinner is available to alter the viscosity
without impairing the temperature-indicating accuracy.

Phase-changing lacquers often are used when a smooth or soft surface must be

tested, or in situations where the surface is not readily accessible for application of a
crayon mark during the heating process. Within seconds after application, the lacquer
dries to a dull matte finish, and it responds rapidly when the temperature to be indicated
is reached. The response delay of a lacquer mark is only a fraction of a second. This
time can be reduced to milliseconds if a mark of minimal thickness is applied.

Upon reaching its rated temperature, the lacquer mark will liquefy. On subsequent

cooling, however, the fluid will not revert to its original unmelted appearance but,
rather, to a glossy or crystalline coating, which is evidence of its having reached the re-
quired temperature. Temperature-indicating lacquers, upon cooling, will not resolidify
at the same temperature at which they melted.

If spraying of the phase-change lacquer is to be the sole means of application, the op-

erator may find it more convenient to use an aerosol-packaged form of this material. An

© 1999 CRC Press LLC

aerosol phase-change temperature indicator is identical to the brush-on material in per-
formance and interpretation.

The first commercial form of the fusible indicator was the pellet, which continues to

be useful in certain applications. Pellets are most frequently employed when extended
heating periods are involved, or where a greater bulk of indicator material is necessary.
They are also useful when observations must be made from a distance and when
air-space temperatures are to be monitored.

Phase-change temperature-indicator pellets are available in flat, 7/16-in-diameter

×

1/8-in-thick tablets. For special applications, miniature pellets, 1/8 in

× 1/8 in, are also

available. The range of 100 to 2500°F covered by the crayon and lacquer-type is, like-
wise, covered by the pellets. Pellets with coverage extending to 3200°F also may be ob-
tained.

Another variation of the phase-change indicator is the temperature-sensitive label.

4

These self-adhesive-backed monitors consist of one or more heat-sensitive indicators
sealed under transparent heat-resistant windows. The centers of the indicator circles
turn from white to black at the temperature ratings shown on the label. The change to
black is irreversible, representing an absorption of the temperature-sensitive substance
into its backing material. After registering the temperature history of the workpiece, the
exposed monitor label can be removed and affixed to a service report to remain part of a
permanent record.

Selection Process

Fusible temperature indicators and lacquers have at least three major advantages over
other methods of determining surface temperature [1]:

·

The temperature indications obtained are unquestionably those of the surface be-
ing tested. It is not necessary to equilibrate a relatively massive probe with the
surface (a probe may conduct heat away from the region being tested), which re-
quires the use of a correction factor to obtain the actual surface temperature.

·

There is no delay in obtaining an indication. Because a mark left by a crayon or a
lacquer is of extremely small mass, it attains rapid equilibrium with the surface.
There is no conduction of heat away from the surface, which would prolong re-
sponse time and result in erroneously low temperature readings.

·

The technique is simple and economical. Determination of surface temperature
by most other means requires technical competence and skill and, in some cases,
sophisticated instrumentation. Accurate surface temperature readings can be ob-
tained with fusible indicators and lacquers with little effort, training, and ex-
pense.

4

Temp-Plate is one such product (Williams-Wahl Corp., Santa Monica, CA).

© 1999 CRC Press LLC

There are numerous instances, particularly in determining heat distribution in com-
plex systems, in which the relative simplicity of fusible indicators and lacquers makes
surface temperature investigations feasible.

8.2.5 Air-Handling System

All modern air-cooled PA tubes use an air-system socket and matching chimney for
cooling [1]. The chimney is designed to be an integral part of the tube cooling system.
Operation without the chimney may significantly reduce airflow through the tube and
result in overdissipation of the device. It also is possible that operation without the
proper chimney could damage other components in the circuit because of excessive
radiated heat. Normally, the tube socket is mounted in a pressurized compartment so
that cooling air passes through the socket and is guided to the anode cooling fins, as
illustrated in

Figure 8.25

.

Cooling of the socket assembly is important for proper cooling of the tube base and

for cooling the contact rings of the tube itself. The contact fingers used in the collet as-
sembly of a socket typically are made of beryllium copper. If subjected to temperatures
above 150°C for an extended period of time, the beryllium copper will lose its temper
(springy characteristic) and will no longer make good contact with the base rings of the
device. In extreme cases, this type of socket problem may lead to arcing, which can
burn through the metal portion of the tube base ring. Such an occurrence ultimately
may lead to catastrophic failure of the device because of a loss of the vacuum envelope.
Other failure modes for a tube socket include arcing between the collet and tube ring,
which can weld a part of the socket and tube together. The result is failure of both the
tube and the socket.

8.3

Operating Environment

Long-term reliability of a power vacuum tube requires regular attention to the operat-
ing environment. Periodic tests and preventive maintenance are important compo-
nents of this effort. Optimum performance of the cooling system can be achieved only
when all elements of the system are functioning properly.

8.3.1 Air-Handling System

The temperature of the intake air supply is a parameter that is usually under the con-
trol of the end user. The preferred cooling air temperature is typically no higher than
75°F, and no lower than the room dew point. The air temperature should not vary be-
cause of an oversized air-conditioning system or because of the operation of other
pieces of equipment at the facility. Monitoring the PA tube exhaust stack temperature
is an effective method of evaluating overall RF system performance. This can be eas-
ily accomplished. It also provides valuable data on the cooling system and final stage
tuning.

Another convenient method for checking the efficiency of the transmitter cooling

system over a period of time involves documenting the back pressure that exists within

© 1999 CRC Press LLC

the PA cavity. This measurement is made with a manometer, a simple device that is
available from most heating, ventilation, and air-conditioning (HVAC) suppliers. The
connection of a simplified manometer to a transmitter PA output compartment is illus-
trated in

Figure 8.26

.

By charting the manometer readings, it is possible to accurately measure the perfor-

mance of the transmitter cooling system over time. Changes resulting from the buildup
of small dust particles (microdust) may be too gradual to be detected except through
back-pressure charting. Deviations from the typical back-pressure value, either higher
or lower, could signal a problem with the air-handling system. Decreased PA input or
output compartment back pressure could indicate a problem with the blower motor or
an accumulation of dust and dirt on the blades of the blower assembly. Increased back
pressure, on the other hand, could indicate dirty PA tube anode cooling fins (for the in-
put compartment case) or a buildup of dirt on the PA exhaust ducting (for the output
compartment case). Either condition is cause for concern. A system suffering from re-

Figure 8.25 Airflow system for an air-cooled power tube.

© 1999 CRC Press LLC

duced air pressure into the PA compartment must be serviced as soon as possible. Fail-
ure to restore the cooling system to proper operation may lead to premature failure of
the PA tube or other components in the input or output compartments. Cooling prob-
lems do not improve with time; they always get worse.

Failure of the PA compartment air-interlock switch to close reliably may be an early

indication of impending trouble in the cooling system. This situation could be caused
by normal mechanical wear or vibration of the switch assembly, or it may signal that the
PA compartment air pressure has dropped. In such a case, documentation of manome-
ter readings will show whether the trouble is caused by a failure of the air pressure
switch or a decrease in the output of the air-handling system.

Figure 8.26 A manometer device used for measuring back pressure in the PA compart-
ment of an RF generator.

© 1999 CRC Press LLC

8.3.2 Air Cooling System Design

Cooling system performance in an RF generator is not necessarily related to airflow
volume. The cooling capability of air is a function of its mass, not its volume. The de-
signer must determine an appropriate airflow rate within the equipment and establish
the resulting resistance to air movement. A specified static pressure that should be
present within the ducting of the transmitter can be a measure of airflow. For any
given combination of ducting, filters, heat sinks, RFI honeycomb shielding, tubes,
tube sockets, and other elements in the transmitter, a specified system resistance to
airflow can be determined. It is important to realize that any changes in the position or
number of restricting elements within the system will change the system resistance
and, therefore, the effectiveness of the cooling. The altitude of operation is also a con-
sideration in cooling system design. As altitude increases, the density (and cooling
capability) of air decreases. A calculated increase in airflow is required to maintain
the cooling effectiveness that the system was designed to achieve.

Figure 8.27

shows a typical high-power transmitter plant. The building is oriented so

that the cooling activity of the blowers is aided by normal wind currents during the sum-
mer months. Air brought in from the outside for cooling is filtered in a hooded air-in-
take assembly. The building includes a heater and air conditioner.

Figure 8.27 A typical heating and cooling arrangement for a high-power transmitter in-
stallation. Ducting of PA exhaust air should be arranged so that it offers minimum resis-
tance to airflow.

© 1999 CRC Press LLC

The layout of a transmitter room HVAC system can have a significant impact on the

life of the PA tube(s) and the ultimate reliability of the RF generator. Air intake and out-
put ports must be designed with care to avoid airflow restrictions and back-pressure
problems. This process, however, is not as easy as it may seem. The science of airflow is
complex and generally requires the advice of a qualified HVAC consultant.

To help illustrate the importance of proper cooling system design and the real-world

problems that some facilities have experienced, consider the following examples taken
from actual case histories:

Case 1

A fully automatic building ventilation system (

Figure 8.28

) was installed to maintain

room temperature at 20°C during the fall, winter, and spring. During the summer,
however, ambient room temperature would increase to as much as 60°C. A field sur-
vey showed that the only building exhaust route was through the transmitter. There-
fore, air entering the room was heated by test equipment, people, solar radiation on
the building, and radiation from the transmitter itself before entering the transmitter.
The problem was solved through the addition of an exhaust fan (3000 cfm). The 1 hp
fan lowered room temperature by 20°C.

Case 2

A simple remote installation was constructed with a heat-recirculating feature for the
winter (

Figure 8.29

). Outside supply air was drawn by the transmitter cooling system

blowers through a bank of air filters, and hot air was exhausted through the roof. A

Figure 8.28 Case study in which excessive summertime heating was eliminated
through the addition of a 1 hp exhaust blower to the building.

© 1999 CRC Press LLC

small blower and damper were installed near the roof exit point. The damper allowed
hot exhaust air to blow back into the room through a tee duct during the winter
months. For summer operation, the roof damper was switched open and the room
damper closed. For winter operation, the arrangement was reversed. The facility, how-
ever, experienced short tube life during winter operation, even though the ambient
room temperature during winter was not excessive.

The solution involved moving the roof damper 12 ft down to just above the tee. This

eliminated the stagnant "air cushion" above the bottom heating duct damper and signif-
icantly improved airflow in the region. Cavity back pressure was, therefore, reduced.
With this relatively simple modification, the problem of short tube life disappeared.

Case 3

An inconsistency regarding test data was discovered within a transmitter manufac-
turer's plant. Units tested in the engineering lab typically ran cooler than those at the
manufacturing test facility.

Figure 8.30

shows the test station difference, a 4-ft ex-

haust stack that was used in the engineering lab. The addition of the stack increased

Figure 8.29 Case study in which excessive back pressure to the PA cavity was experi-
enced during winter periods, when the rooftop damper was closed. The problem was
eliminated by repositioning the damper as shown.

© 1999 CRC Press LLC

airflow by up to 20 percent because of reduced air turbulence at the output port, re-
sulting in a 20°C decrease in tube temperature.

These examples point out how easily a cooling problem can be caused during HVAC

system design. All power delivered to the transmitter either is converted to RF energy
and sent to the antenna or becomes heated air (or water). Proper design of a cooling sys-
tem, therefore, is a part of transmitter installation that should not be taken lightly.

8.3.3 Site Design Guidelines

There are any number of physical plant designs that will provide for reliable operation
of high-power RF systems [3]. One constant, however, is the requirement for tight
temperature control. Cooling designs can be divided into three broad classifications:

·

Closed site design

·

Open site design

·

Hybrid design

If the equipment user is to provide adequately for hot air exhaust and fresh air intake,

the maximum and minimum environmental conditions in which the equipment will op-

Figure 8.30 Case study in which air turbulence at the exhaust duct resulted in reduced
airflow through the PA compartment. The problem was eliminated by adding a 4-ft ex-
tension to the output duct.

© 1999 CRC Press LLC

erate must be known. In addition, the minimum cooling requirements of the equipment
must be provided by the manufacturer. The following parameters should be considered:

·

Site altitude

·

Maximum expected outside air temperature

·

Minimum expected outside air temperature

·

Total airflow through the transmitter

·

Air temperature rise through the transmitter

·

Air exhaust area

Keep in mind that the recommended cooling capacity for a given RF system applies

only to cooling the transmitter; any additional cooling load in the building must be con-
sidered separately when selecting the air system components. The transmitter exhaust
should not be the only exhaust port in the room because heat from the peripheral equip-
ment would then be forced to go out through the transmitter.

The sensible-heat load, then, is the sum of all additional heat loads, including:

·

Solar radiation

·

Heat gains from equipment and lights

·

Heat gains from personnel in the area that is to be cooled

Closed Site Design

Figure 8.31

illustrates a site layout that works well in most climates, as long as the

transmitter is small and the building is sealed and well insulated [3]. In fact, this
closed configuration will ensure the longest possible transmitter life and lowest main-
tenance cost. No outside air laden with moisture and contaminants circulates through
the transmitter.

At sites using this arrangement, it has been observed that periodic transmitter clean-

ing is seldom required. In a typical closed system, the air conditioner is set to cool when
the room temperature reaches 75 to 80°F. The closed system also uses a louvered emer-
gency intake blower, which is set by its own thermostat to pull in outside air if the room
temperature reaches excessive levels (above 90°F). This blower is required in a closed
configuration to prevent the possibility of thermal runaway if the air conditioner fails.
Without such an emergency ventilation system, the transmitter would recirculate its
own heated air, further heating the room. System failure probably would result.

During winter months, the closed system is self-heating (unless the climate is harsh,

or the transmitter power output is low), because the transmitter exhaust is not ducted
outside but simply empties into the room. Also during these months, the emergency in-
take blower can be used to draw cold outside air into the room instead of using the air
conditioner, although this negates some of the cleanliness advantages inherent to the
closed system.

© 1999 CRC Press LLC

An exhaust blower should not be substituted for an intake blower, because positive

room pressure is desired for venting the room. This ensures that all air in the building
has passed through the intake air filter. Furthermore, the louvered emergency exhaust
vent(s) should be mounted high in the room, so that hot air is pushed out of the building
first.

The closed system usually makes economic sense only if the transmitter exhaust

heat load is relatively small.

Periodic maintenance of a closed system involves the following activities:

·

Checking and changing the air conditioner filter periodically

·

Cleaning the transmitter air filter as needed

·

Checking that the emergency vent system works properly

·

Keeping the building sealed from insects and rodents

Open Site Design

Figure 8.32

depicts a site layout that is the most economical to construct and operate

[3]. The main attribute of this approach is that the transmitter air supply is not heated
or cooled, resulting in cost savings. This is not a closed system; outdoor air is pumped
into the transmitter room through air filters. The transmitter then exhausts the hot air.
If the duct work is kept simple and the transmitter has a dedicated exhaust port, such a
direct exhaust system works adequately. Many transmitters do not lend themselves to

Figure 8.31 Closed site ventilation design, with a backup inlet/outlet system.

© 1999 CRC Press LLC

a direct exhaust connection, however, and a hood mounted over the transmitter may be
required to collect the hot air, as illustrated in

Figure 8.33

. With a hooded arrange-

ment, it may be necessary to install a booster fan in the system, typically at the wall or
roof exit, to avoid excessive back pressure.

When building an open air-circulation system, there are some important additional

considerations. The room must be positively pressurized so that only filtered air, which
has come through the intake blower and filter, is available to the transmitter. With a neg-
atively pressurized room, air will enter through every hole and crack in the building,
and will not necessarily be filtered. Under negative pressure, the transmitter blower
also will have to work harder to exhaust air.

The intake blower should be a "squirrel-cage" type rather than a fan type. A fan is

meant to move air within a pressurized environment; it cannot compress the air. A
squirrel-cage blower will not only move air, but also will pressurize the area into which
the air is directed. Furthermore, the intake air blower must be rated for more cubic feet
per minute (cfm) airflow than the transmitter will exhaust outside. A typical 20 kW FM
broadcast transmitter, for example, will exhaust 500 cfm to 1000 cfm. A blower of 1200
cfm, therefore, would be an appropriate size to replenish the transmitter exhaust and
positively pressurize the room.

In moderate and warm climates, the intake blower should be located on a north-fac-

ing outside wall. If the air intake is on the roof, it should be elevated so that it does not
pick up air heated by the roof surface.

Figure 8.32 Open site ventilation design, using no air conditioner.

© 1999 CRC Press LLC

High-quality pleated air filters are recommended. Home-style fiberglass filters are

not sufficient. Local conditions may warrant using a double filtration system, with
coarse and fine filters in series.

Secure the advice of a knowledgeable HVAC shop when designing filter boxes. For

a given cfm requirement, the larger the filtration area, the lower the required air velocity
through the filters. This lower velocity results in better filtration than forcing more air
through a small filter. In addition, the filters will last longer. Good commercial filtra-
tion blowers designed for outdoor installation are available from industrial supply
houses and are readily adapted to RF facility use.

The transmitter exhaust may be ducted through a nearby wall or through the roof.

Avoid ducting straight up, however. Many facilities have suffered water damage to
transmitters in such cases because of the inevitable deterioration of roofing materials.
Normally, ducting the exhaust through an outside wall is acceptable. The duct work
typically is bent downward a foot or two outside the building to keep direct wind from
creating back pressure in the exhaust duct. Minimize all bends in the duct work. If a 90°
bend must be made, it should be a large-radius bend with curved helper vanes inside the
duct to minimize turbulence and back pressure. A 90° L- or T-bend is not recom-
mended, unless oversized and equipped with internal vanes to assist the turning airflow.

For moderate and cool climates, an automatic damper can be employed in the ex-

haust duct to direct a certain amount of hot exhaust air back into the transmitter build-

Figure 8.33 Transmitter exhaust-collection hood.

© 1999 CRC Press LLC

ing as needed for heating. This will reduce outside air requirements, providing clean,
dry, heated air to the transmitter during cold weather.

Hybrid Design

Figure 8.34

depicts a hybrid site layout that is often used for high-power transmitters

or sites supporting multiple transmitters [3]. As with the layout in

Figure 8.32

, the

room is positively pressurized with clean, filtered air from the outside. A portion of
this outside air is then drawn through the air conditioner for cooling before delivery to
the transmitter area. Although not all of the air in the room goes through the air condi-
tioner, enough does to make a difference in the room temperature. In humid areas,
much of the moisture is removed by the air conditioner. The transmitter exhaust is di-
rected outside.

Such a hybrid system is often the choice for larger transmitter sites where a closed

system would prove too costly, but an unconditioned system would run excessively hot
during the summer months.

Some closed or hybrid systems use two parallel air conditioners. Most of the time,

only one is in use. The thermostats of the units are staggered so that if one cannot keep
the air below the ideal operating point, the other will turn on to assist.

Figure 8.34 Hybrid site ventilation design, using an air conditioner in addition to filtered
outside air.

© 1999 CRC Press LLC

8.3.4 Water/Vapor Cooling System Maintenance

The cooling system is vital to any RF generator. In a klystron-based unit, for example,
the cooling system may dissipate a large percentage of the input ac power in the form
of waste heat in the collector. Pure, distilled water should be used. Deionized water is
required unless the isolation of high voltages is unnecessary, as in the case of the col-
lector of a klystron (the collector typically operates at or near ground potential).

Any impurities in the cooling system water eventually will find their way into the

water jacket and cause corrosion of the plate or collector. It is essential to use high-pu-
rity water with low conductivity, and to replace the water in the cooling system as
needed. Efficient energy transfer from the heated surface into the water is necessary for
long vacuum tube life. Oil, grease, soldering flux residue, and pipe sealant containing
silicone compounds must be excluded from the cooling system. This applies to both va-
por- and liquid-conduction cooling systems, although it is usually more critical in the
vapor-phase type. The sight glass in a vapor-phase water jacket provides a convenient
checkpoint for coolant condition.

In general, greater flows and greater pressures are inherent in liquid-cooled vs. va-

por-phase systems, and, when a leak occurs, large quantities of coolant can be lost be-
fore the problem is discovered. The condition of gaskets, seals, and fittings, therefore,
should be checked regularly. Many liquid-cooled tubes use a distilled water and ethyl-
ene glycol mixture. (Do not exceed a 50:50 mix by volume.) The heat transfer of the
mixture is lower than that of pure water, requiring the flow to be increased, typically by
25 percent. Greater coolant flow suggests closer inspection of the cooling system after
adding the glycol.

The action of heat and air on ethylene glycol causes the formation of acidic products.

The acidity of the coolant can be checked with litmus paper. Buffers can be added to the
glycol mixture. Buffers are alkaline salts that neutralize acid forms and prevent corro-
sion. Because they are ionizable chemical salts, the buffers cause the conductivity of
the coolant to increase. Consult the power tube manufacturer before adding buffers.

The following general guidelines are recommended for power tube cooling systems:

·

The resistivity of the water must be maintained above the minimum level estab-
lished by the power tube manufacturer. Although recommended values vary,
generally speaking, resistivity should be maintained at or above 1 M

-cm (at

30ºC) for power tubes operating at a high-voltage, referenced to ground. A resis-
tivity value of 30 k

-cm or greater (at 30ºC) is usually sufficient for tubes whose

collector operates at or near ground potential (such as a klystron).

·

The pH factor must be within the range of 6.0 to 8.0.

·

The particulate matter size must not be greater than 50 microns (325 mesh).

·

The inlet water temperature must not exceed 70°C, and this temperature should
be regulated to ±5ºC.

When the water in the cooling system fails to satisfy any one of these requirements,
prompt action is necessary to correct the deficiency. If the water is contaminated, the
system must be flushed and replaced with clean water.

© 1999 CRC Press LLC

Tests for Purity

The recommended method for measuring resistivity, pH, and particulate matter is by
use of laboratory instruments. However, the following simple tests should provide
sufficient information to determine the general condition of the coolant.

·

The appearance of the water is a good general indicator of coolant condition. If
the water looks turbid or tastes or smells brackish, it is good practice to change the
water and flush the system.

·

The pH factor is easily checked by using pH paper or litmus paper. Sold under a
variety of trade names, pH paper is available at laboratory supply stores. Direc-
tions for use usually are printed on the package.

·

Particulate matter and impurities can be checked by using the "foaming test" (de-
scribed in the next section).

·

Resistivity can be measured accurately with a resistance bridge. Generally speak-
ing, if the water does not pass the foaming test, the resistivity will be low as well.

Foaming Test for Water Purity

One of the many factors affecting the life and operating efficiency of vapor-cooled
tubes is the purity of the water in the cooling system. If impurities are present, foam-
ing may occur that will inhibit heat transfer, thereby lowering the cooling efficiency
of the system. The impurities that most frequently produce foaming are:

·

Cleaning-compound residue

·

Detergents

·

Joint-sealing compounds

·

Oily rust preventives in pumps and other components

·

Valve-stem packing

·

Impurities in tap water

Tests should be conducted periodically to confirm the quality of the coolant in a va-
por-phase system. Conduct such tests after each water change, system cleaning, or
modification. The following items are required:

·

A 1/2-in

× 4-in glass test tube with rubber stopper

·

A 1-pint glass or polypropylene bottle with cap

The procedure is as follows:

·

Fill the cooling system with water, and circulate it until thoroughly mixed (about
30 minutes).

·

Drain a sample of water into the bottle, and cool to room temperature.

© 1999 CRC Press LLC

·

If the water sample stands for more than 1 hour, slowly invert the capped bottle
about 10 times. Avoid shaking the bottle because this will create air bubbles in the
water. (When the water is static, foaming impurities tend to collect at the surface.
This step mixes the sample without generating foam.)

·

Using the sample water, rinse the test tube and stopper three times.

·

Fill the test tube to the halfway mark with the sample water.

·

Shake the test tube vigorously for 15 s.

·

Let the sample stand for 15 s.

·

Observe the amount of foam remaining on top of the water and compare it with
the drawings shown in

Figure 8.35

.

A completely foam-free water surface and test tube, as illustrated in (a), indicates no

foam-producing impurities. If the water surface and test tube wall are partially covered
with foam, but a circle of clear water appears in the center, the impurity level is tempo-
rarily acceptable (b). A second test should be made in about one week. If the foam layer
completely bridges the inside of the test tube (c), the system should be flushed and
cleaned.

8.4

References

1. Laboratory Staff, The Care and Feeding of Power Grid Tubes, Varian Associates,

San Carlos, CA, 1984.

2. Besch, David F., "Thermal Properties," in The Electronics Handbook, Jerry C.

Whitaker (ed.), CRC Press, Boca Raton, FL, pp. 127134, 1996.

Figure 8.35 Results of the "foaming test" for water purity: (

a) pure sample, (b) sample

that is acceptable on a short-term basis, (

c) unacceptable sample.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

3. Harnack, Kirk, "Airflow and Cooling in RF Facilities," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, pp. 3338, November 1992.

8.5

Bibliography

"Cleaning and Flushing Klystron Water and Vapor Cooling Systems," Application En-

gineering Bulletin no. AEB-32, Varian Associates, Palo Alto, CA, February 1982.

Crutchfield, E. B. (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, pp. 547550, 1992.

"Foaming Test for Water Purity," Application Engineering Bulletin no. AEB-26,

Varian Associates, Palo Alto, CA, September 1971.

Harnack, Kirk, "Airflow and Cooling in RF Facilities," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, pp. 3338, November 1992.

Harper, Charles A., Electronic Packaging and Interconnection Handbook,

McGraw-Hill, New York, NY, 1991.

"High Power Transmitting Tubes for Broadcasting and Research," Philips Technical

Publication, Eindhoven, the Netherlands, 1988.

Integral Cavity Klystrons for UHF-TV Transmitters, Varian Associates, Palo Alto,

CA.

Kimmel, E., "Temperature Sensitive Indicators: How They Work, When to Use

Them," Heat Treating, March 1983.

"Power Grid Tubes for Radio Broadcasting," Thomson-CSF Publication #DTE-115,

Thomson-CSF, Dover, NJ, 1986.

"Protecting Integral-Cavity Klystrons Against Water Leakage," Technical Bulletin no.

3834, Varian Associates, Palo Alto, CA, July 1978.

"Temperature Measurements with Eimac Power Tubes," Application Bulletin no. 20,

Varian Associates, San Carlos, CA, January 1984.

"Water Purity Requirements in Liquid Cooling Systems," Application Bulletin no. 16,

Varian Associates, San Carlos, CA, July 1977.

"Water Purity Requirements in Water and Vapor Cooling Systems," Application Engi-

neering Bulletin no. AEB-31, Varian Associates, Palo Alto, CA, September 1991.

Whitaker, Jerry C., Maintaining Electronic Systems, CRC Press, Boca Raton, FL,

1992.

Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, NY, 1990.

Wozniak, Joe, "Using Tetrode Power Amplifiers in High Power UHF-TV Transmit-

ters," NAB 1992 Broadcast Engineering Conference Proceedings, National Associ-
ation of Broadcasters, Washington, DC, pp. 209211, 1992.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

9 Reliability

Considerations"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

9

Reliability Considerations

9.1

Introduction

The ultimate goal of any design engineer or maintenance department is zero down-
time. This is an elusive goal, but one that can be approximated by examining the vul-
nerable areas of plant operation and taking steps to prevent a sequence of events that
could result in system failure. In cases where failure prevention is not practical, a reli-
ability assessment should encompass the stocking of spare parts, circuit boards, or
even entire systems. A large facility may be able to cost-justify the purchase of
backup gear that can be used as spares for the entire complex. Backup hardware is ex-
pensive, but so is downtime. Because of the finite lifetime of power vacuum tubes,
these considerations are of special significance.

Failures can, and do, occur in electronic systems. The goal of product quality assur-

ance at every step in the manufacturing and operating chain is to ensure that failures do
not produce a systematic or repeatable pattern. The ideal is to eliminate failures alto-
gether. Short of that, the secondary goal is to end up with a random distribution of fail-
ure modes. This indicates that the design of the system is fundamentally optimized and
that failures are caused by random events that cannot be predicted. In an imperfect
world, this is often the best that end users can hope for. Reliability and maintainability
must be built into products or systems at every step in the design, construction, and
maintenance process. They cannot be treated as an afterthought.

9.1.1 Terminology

To understand the principles of reliability engineering, the following basic terms must
be defined:

·

Availability. The probability that a system subject to repair will operate satisfac-
torily on demand.

·

Average life. The mean value for a normal distribution of product or component
lives. This term is generally applied to mechanical failures resulting from
"wear-out."

© 1999 CRC Press LLC

·

Burn-in. The initially high failure rate encountered when a component is placed
on test. Burn-in failures usually are associated with manufacturing defects and
the debugging phase of early service.

·

Defect. Any deviation of a unit or product from specified requirements. A unit or
product may contain more than one defect.

·

Degradation failure. A failure that results from a gradual change, over time, in
the performance characteristics of a system or part.

·

Downtime. Time during which equipment is not capable of doing useful work
because of malfunction. This does not include preventive maintenance time.
Downtime is measured from the occurrence of a malfunction to its correction.

·

Failure. A detected cessation of ability to perform a specified function or func

-

tions within previously established limits. A failure is beyond adjustment by the
operator by means of controls normally accessible during routine operation of the
system.

·

Failure mode and effects analysis (FMEA). An iterative documented process
performed to identify basic faults at the component level and determine their ef-
fects at higher levels of assembly.

·

Failure rate. The rate at which failure occurs during an interval of time as a func-
tion of the total interval length.

·

Fault tree analysis (FTA). An iterative documented process of a systematic na-
ture performed to identify basic faults, determine their causes and effects, and es-
tablish their probabilities of occurrence.

·

Lot size. A specific quantity of similar material or a collection of similar units
from a common source; in inspection work, the quantity offered for inspection
and acceptance at any one time. This may be a collection of raw material, parts,
subassemblies inspected during production, or a consignment of finished prod-
ucts to be sent out for service.

·

Maintainability. The probability that a failure will be repaired within a specified
time after it occurs.

·

Mean time between failure (MTBF). The measured operating time of a single
piece of equipment divided by the total number of failures during the measured
period of time. This measurement normally is made during that period between
early life and wear-out failures.

·

Mean time to repair (MTTR). The measured repair time divided by the total
number of failures of the equipment.

·

Mode of failure. The physical description of the manner in which a failure occurs
and the operating condition of the equipment or part at the time of the failure.

·

Part failure rate. The rate at which a part fails to perform its intended function.

© 1999 CRC Press LLC

·

Quality assurance (QA). All those activities, including surveillance, inspection,
control, and documentation, aimed at ensuring that a product will meet its perfor-
mance specifications.

·

Reliability. The probability that an item will perform satisfactorily for a speci

-

fied period of time under a stated set of use conditions.

·

Reliability growth. Actions taken to move a hardware item toward its reliability
potential, during development, subsequent manufacturing, or operation.

·

Reliability predictions. Compiled failure rates for parts, components, subas

-

semblies, assemblies, and systems. These generic failure rates are used as basic
data to predict a value for reliability.

·

Sample. One or more units selected at random from a quantity of product to rep

-

resent that product for inspection purposes.

·

Sequential sampling. Sampling inspection in which, after each unit is inspected,
the decision is made to accept, reject, or inspect another unit. (Note: Sequential
sampling as defined here is sometimes called unit sequential sampling or multiple
sampling.)

·

System. A combination of parts, assemblies, and sets joined together to perform a
specific operational function or functions.

·

Test to failure. Testing conducted on one or more items until a predetermined
number of failures have been observed. Failures are induced by increasing elec-
trical, mechanical, and/or environmental stress levels, usually in contrast to life
tests, in which failures occur after extended exposure to predetermined stress lev-
els. A life test can be considered a test to failure using age as the stress.

9.2

Quality Assurance

Electronic component and system manufacturers design and implement quality assur-
ance procedures for one fundamental reason: Nothing is perfect. The goal of a QA
program is to ensure, for both the manufacturer and the customer, that all but some
small, mutually acceptable percentage of devices or systems shipped will be as close
to perfection as economics and the state of the art allow. There are tradeoffs in this
process. It would be unrealistic, for example, to perform extensive testing to identify
potential failures if the cost of that testing exceeded the cost savings that would be re-
alized by not having to replace the devices later in the field.

The focal points of any QA effort are quality and reliability. These terms are not syn-

onymous. They are related, but they do not provide the same measure of a product:

·

Quality is the measure of a product's performance relative to some established
criteria.

·

Reliability is the measure of a product's life expectancy.

© 1999 CRC Press LLC

Stated from a different perspective, quality answers the question of whether the prod-
uct meets applicable specifications now; reliability answers the question of how long
the product will continue to meet its specifications.

9.2.1 Inspection Process

Quality assurance for components normally is performed through sampling rather
than through 100 percent inspection. The primary means used by QA departments for
controlling product quality at the various processing steps include:

·

Gates. A mandatory sampling of every lot passing through a critical production
stage. Material cannot move on to the next operation until QA has inspected and
accepted the lot.

·

Monitor points. A periodic sampling of some attribute of the component. QA per-
sonnel sample devices at a predetermined frequency to verify that machines and
operators are producing material that meets preestablished criteria.

·

Quality audit. An audit carried out by a separate group within the QA department.
This group is charged with ensuring that all production steps throughout the man-
ufacturer's facility are in accordance with current specifications.

·

Statistical quality control. A technique, based on computer modeling, that incor-
porates data accumulated at each gate and monitor point to construct statistical
profiles for each product, operation, and piece of equipment within the plant.
Analysis of this data over time allows QA engineers to assess trends in product
performance and failure rates.

Quality assurance for a finished subassembly or system may range from a simple

go/no-go test to a thorough operational checkout that may take days to complete.

9.2.2 Reliability Evaluation

Reliability prediction is the process of quantitatively assessing the reliability of a
component or system during development, before large-scale fabrication and field
operation. During product development, predictions serve as a guide by which design
alternatives can be judged for reliability. To be effective, the prediction technique
must relate engineering variables to reliability variables.

A prediction of reliability is obtained by determining the reliability of each critical

item at the lowest system level and proceeding through intermediate levels until an esti-
mate of overall reliability is obtained. This prediction method depends on the availabil-
ity of accurate evaluation models that reflect the reliability of lower-level components.
Various formal prediction procedures are used, based on theoretical and statistical con-
cepts.

© 1999 CRC Press LLC

Parts-Count Method

Although the parts-count method does not relate directly to vacuum tube design, it is
worthwhile to discuss the technique briefly in the context of overall system reliability.
It is fair to point out that this technique has considerable validity when comparisons
of vacuum tube vs. solid-state amplifier designs are being considered. To achieve
high power levels, far more individual components are necessary for a solid-state de-
sign than for a vacuum tube-based system

The parts-count approach to reliability prediction provides an estimate of reliability

based on a count by part type (ICs, transistors, vacuum tube devices, resistors, capaci-
tors, and other components). This method is useful during the early design stage of a
product, when the amount of available detail is limited. The technique involves count-
ing the number of components of each type, multiplying that number by a generic fail-
ure rate for each part type, and summing the products to obtain the failure rate of each
functional circuit, subassembly, assembly, and/or block depicted in the system block di-
agram. The parts-count method is useful in the design phase because it provides rapid
estimates of reliability, permitting assessment of the feasibility of a given concept.

Stress-Analysis Method

The stress-analysis technique is similar to the parts-count method, but utilizes a de-
tailed parts model plus calculation of circuit stress values for each part before deter-
mining the failure rate. Each part is evaluated in its electric circuit and mechanical as-
sembly application based on an electrical and thermal stress analysis. After part fail-
ure rates have been established, a combined failure rate for each functional block is
determined.

9.2.3 Failure Analysis

Failure mode and effects analysis can be performed with data taken from actual fail-
ure modes observed in the field, or from hypothetical failure modes derived from one
of the following:

·

Design analysis

·

Reliability prediction activities

·

Experience with how specific parts fail

In the most complete form of FMEA, failure modes are identified at the component
level. Failures are induced analytically into each component, and failure effects are
evaluated and noted, including the severity and frequency (or probability) of occur-
rence. Using this approach, the probability of various system failures can be calcu-
lated, based on the probability of lower-level failure modes.

Fault tree analysis is a tool commonly used to analyze failure modes found during

design, factory test, or field operations. The approach involves several steps, including
the development of a detailed logic diagram that depicts basic faults and events that can
lead to system failure and/or safety hazards. These data are used to formulate corrective

© 1999 CRC Press LLC

suggestions that, when implemented, will eliminate or minimize faults considered crit-
ical. An example FTA chart is shown in

Figure 9.1

.

9.2.4 Standardization

Standardization and reliability go hand in hand. Standardization of electronic compo-
nents began with military applications in mind; the first recorded work was per-
formed by the U.S. Navy with vacuum tubes. The navy recognized that some control
at the component level was essential to the successful incorporation of electronics
into naval systems.

Standardization and reliability are closely related, although there are many aspects

of standardization whose reliability implications are subtle. The primary advantages of
standardization include:

·

Total product interchangeability. Standardization ensures that products of the
same part number provide the same physical and electrical characteristics. There
have been innumerable instances of a replacement device bearing the same part
number as a failed device, but not functioning identically to it. In many cases, the
differences in performance were so great that the system would not function at all
with the new device.

·

Consistency and configuration control. Component manufacturers constantly re-
define their products to improve yields and performance. Consistency and con-
figuration control assure the user that product changes will not affect the
interchangeability of the part.

·

Efficiencies of volume production. Standardization programs usually result in
production efficiencies that reduce the costs of parts, relative to components with
the same level of reliability screening and control.

·

Effective spares management. The use of standardized components makes the
stocking of spare parts a much easier task. This aspect of standardization is not a
minor consideration. For example, the costs of placing, expediting, and receiving
material against one Department of Defense purchase order may range from $300
to $1100. Accepting the lowest estimate, the conversion of 10 separate part num-
bers to one standardized component could effect immediate savings of $3000 just
in purchasing and receiving costs.

·

Multiple product sources. Standardization encourages second-sourcing. Multi-
ple sources help hold down product costs and encourage manufacturers to strive
for better product performance.

9.3

Reliability Analysis

The science of reliability and maintainability matured during the 1960s with the de-
velopment of sophisticated computer systems and complex military and spacecraft
electronics. Components and systems never fail without a reason. That reason may be

478 Chapter Nine

© 1999 CRC Press LLC

Figure 9.1 Example fault tree analysis diagram: (

a) process steps, (b) fault tree sym-

bols, (

c, next page) example diagram.

(

a)

(

b)

© 1999 CRC Press LLC

difficult to find, but determination of failure modes and weak areas in system design
or installation is fundamental to increasing the reliability of any component or sys-
tem, whether it is a power vacuum tube, integrated circuit, aircraft autopilot, or broad-
cast transmitter.

All equipment failures are logical; some are predictable. A system failure usually is

related to poor-quality components or to abuse of the system or a part within, either be-
cause of underrating or environmental stress. Even the best-designed components can
be badly manufactured. A process can go awry, or a step involving operator interven-
tion may result in an occasional device that is substandard or likely to fail under normal
stress. Hence, the process of screening and/or burn-in to weed out problem parts is a
universally accepted quality control tool for achieving high reliability.

9.3.1 Statistical Reliability

Figure 9.2

illustrates what is commonly known as the bathtub curve. It divides the ex-

pected lifetime of a class of parts into three segments: infant mortality, useful life, and
wear-out. A typical burn-in procedure consists of the following steps:

Figure 9.1

c

© 1999 CRC Press LLC

·

The parts are electrically biased and loaded; that is, they are connected in a circuit
representing a typical application.

·

The parts are cycled on and off (power applied, then removed) for a predeter-
mined number of times. The number of cycles may range from 10 to several thou-
sand during the burn-in period, depending on the component under test.

·

The components under load are exposed to high operating temperatures for a se-
lected time (typically 72 to 168 hours). This constitutes an accelerated life test for
the part.

An alternative approach involves temperature shock testing, in which the compo-

nent product is subjected to temperature extremes, with rapid changes between the
hot-soak and cold-soak conditions. After the stress period, the components are tested
for adherence to specifications. Parts meeting the established specifications are ac-
cepted for shipment to customers. Parts that fail to meet them are discarded.

Figure 9.3

illustrates the benefits of temperature cycling to product reliability. The

charts compare the distribution of component failures identified through steady-state
high-temperature burn-in vs. temperature cycling. Note that cycling screened out a sig-
nificant number of failures. The distribution of failures under temperature cycling usu-
ally resembles the distribution of field failures. Temperature cycling simulates

Figure 9.2 The statistical distribution of equipment or component failures vs. time for
electronic systems and devices.

© 1999 CRC Press LLC

real-world conditions more closely than steady-state burn-in. The goal of burn-in test-
ing is to ensure that the component lot is advanced beyond the infant mortality stage (T1
on the bathtub curve). This process is used not only for individual components, but for
entire systems as well.

Such a systems approach to reliability is effective, but not foolproof. The burn-in pe-

riod is a function of statistical analysis; there are no absolute guarantees. The natural
enemies of electronic parts are heat, vibration, and excessive voltage.

Figure 9.4

docu-

ments failures vs. hours in the field for a piece of avionics equipment. The conclusion is
made that a burn-in period of 200 hours or more will eliminate 60 percent of the ex-
pected failures. However, the burn-in period for another system using different compo-
nents may well be a different number of hours.

The goal of burn-in testing is to catch system problems and potential faults before

the device or unit leaves the manufacturer. The longer the burn-in period, the greater the
likelihood of catching additional failures. The problems with extended burn-in, how-
ever, are time and money. Longer burn-in translates to longer delivery delays and addi-
tional costs for the equipment manufacturer, which are likely to be passed on to the end
user. The point at which a product is shipped is based largely on experience with similar
components or systems and the financial requirement to move products to customers.

Roller-Coaster Hazard Rate

The bathtub curve has been used for decades to represent the failure rate of an elec-
tronic system. More recent data, however, has raised questions regarding the accuracy
of the curve shape. A growing number of reliability scientists now believe that the
probability of failure, known in the trade as the hazard rate, is more accurately repre-
sented as a roller-coaster track, as illustrated in

Figure 9.5

. Hazard rate calculations

Figure 9.3 Distribution of component failures identified through burn-in testing: (

a)

steady-state high-temperature burn-in, (

b) temperature cycling.

(

a)

(

b)

© 1999 CRC Press LLC

require analysis of the number of failures of the system under test, as well as the num-
ber of survivors. Advocates of this approach point out that previous estimating pro-
cesses and averaging tended to smooth the roller-coaster curve so that the humps were
less pronounced, leading to an incorrect conclusion insofar as the hazard rate was
concerned. The testing environment also has a significant effect on the shape of the
hazard curve, as illustrated in

Figure 9.6

. Note that at the higher operating tempera-

ture (greater environmental stress), the roller-coaster hump has moved to an earlier
age.

9.3.2 Environmental Stress Screening

The science of reliability analysis is rooted in the understanding that there is no such
thing as a random failure; every failure has a cause. For reasonably designed and con-
structed electronic equipment, failures not caused by outside forces result from
built-in flaws or latent defects. Because different flaws are sensitive to different
stresses, a variety of environmental forces must be applied to a unit under test to iden-
tify any latent defects. This is the underlying concept behind environmental stress
screening (ESS).

Figure 9.4 The failure history of a piece of avionics equipment vs. time. Note that 60
percent of the failures occurred within the first 200 hours of service. (

After [1].)

© 1999 CRC Press LLC

ESS, which has come into widespread use for aeronautics and military products,

takes the burn-in process a step further by combining two of the major environmental
factors that cause parts or units to fail: heat and vibration. Qualification testing for

Figure 9.5 The roller-coaster hazard rate curve for electronic systems. (

After [2].)

Figure 9.6 The effects of environmental conditions on the roller-coaster hazard rate
curve. (

After [2].)

© 1999 CRC Press LLC

products at a factory practicing ESS involves a carefully planned series of checks for
each unit off the assembly line. Units are subjected to random vibration and tempera-
ture cycling during production (for subassemblies and discrete components) and upon
completion (for systems). The goal is to catch product defects at the earliest possible
stage of production. ESS also can lead to product improvements in design and manu-
facture if feedback from the qualification stage to the design and manufacturing stages
is implemented.

Figure 9.7

illustrates the improvement in reliability that typically can

be achieved through ESS over simple burn-in screening, and through ESS with feed-
back to earlier production stages. Significant reductions in equipment failures in the
field can be gained.

Table 9.1

compares the merits of conventional reliability testing

and ESS.

Designing an ESS procedure for a given product is no easy task. The environmental

stresses imposed on the product must be great enough to cause fallout of marginal com-
ponents during qualification testing. The stresses must not be so great, however, as to
cause failures in good products. Any unit that is stressed beyond its design limits even-
tually will fail. The proper selection of stress parameters--generally, random vibration
on a vibration generator and temperature cycling in an environmental chamber--can,
in minutes, uncover product flaws that might take weeks or months to manifest them-
selves in the field. The result is greater product reliability for the user.

The ESS concept requires that every product undergo qualification testing before

integration into a larger system for shipment to an end user. The flaws uncovered by
ESS vary from one unit to the next, but types of failures tend to respond to particular en-

Figure 9.7 The effects of environmental stress screening on the reliability

bathtub

curve. (

After [3].)

© 1999 CRC Press LLC

vironmental stresses. Available data clearly demonstrate that the burn-in screens must
match the flaws sought; otherwise, the flaws will probably not be found.

The concept of flaw-stimulus relationships can be presented in Venn diagram form.

Figure 9.8

shows a Venn diagram for a hypothetical, but specific, product. The required

screen would be different for a different product. For clarity, not all stimuli are shown.
Note that there are many latent defects that will not be uncovered by any one stimulus.
For example, a solder splash that is just barely clinging to a circuit element probably
would not be broken loose by high-temperature burn-in or voltage cycling, but vibra-
tion or thermal cycling probably would break the particle loose. Remember also that
the defect may be observable only during stimulation and not during a static bench test.

The levels of stress imposed on a product during ESS should be greater than the

stress to which the product will be subjected during its operational lifetime, but still be
below the maximum design parameters. This rule of thumb is pushed to the limits under
an enhanced screening process. Enhanced screening places the component or system at
well above the expected field environmental levels. This process has been found to be
useful and cost-effective for many programs and products. Enhanced screening, how-
ever, requires the development of screens that are carefully identified during product
design and development so that the product can survive the qualification tests. En-
hanced screening techniques often are required for cost-effective products on a cra-
dle-to-grave basis; that is, early design changes for screenability save tremendous costs
over the lifetime of the product.

The types of products that can be checked economically through ESS break down

into two categories: high-dollar items and mass-produced items. Units that are physi-
cally large in size, such as RF generators, are difficult to test in the finished state. Still,
qualification tests using more primitive methods, such as cam-driven truck-bed shak-
ers, are practical. Because most large systems generate a large amount of heat, subject-
ing the equipment to temperature extremes also may be accomplished. Sophisticated
ESS for large systems, however, must rely on qualification testing at the subassembly
stage.

Table 9.1 Comparison of Conventional Reliability Testing and Environmental Stress
Screening (

After [2].)

Parameter

Conventional Testing

Environmental Stress

Screening

Test condition

Simulates operational equip-
ment profile

Accelerated stress condition

Test sample size

Small

100 percent of production

Total test time

Limited

High

Number of failures

Small

Large

Reliability growth

Potential for gathering useful
data small

Potential for gathering useful
data good

© 1999 CRC Press LLC

The basic hardware complement for an ESS test station includes a thermal chamber

shaker and controller/monitor. A typical test sequence includes 10 minutes of exposure
to random vibration, followed by 10 cycles between temperature minimum and maxi-
mum. To save time, the two tests may be performed simultaneously.

9.3.3 Latent Defects

The cumulative failure rate observed during the early life of an electronic system is
dominated by the latent defect content of the product, not its inherent failure rate.
Product design is the major determinant of inherent failure rate. A product design will
show a higher-than-expected inherent rate if the system contains components that are
marginally overstressed, have inadequate functional margin, or contain a
subpopulation of components that exhibit a wear-out life shorter than the useful life of
the product. Industry has grown to expect the high instantaneous failure rate observed
when a new product is placed into service. The burn-in process, whether ESS or con-
ventional, is aimed at shielding customers from the detrimental effects of infant mor-
tality. The key to reducing early-product-life failures lies in reducing the number of
latent defects.

A latent defect is some abnormal characteristic of the product or its parts that is

likely to result in failure at some point, depending on:

·

The degree of abnormality

Figure 9.8 Venn diagram representation of the relationship between flaw precipitation
and applied environmental stress. (

After [4].)

© 1999 CRC Press LLC

·

The magnitude of applied stress

·

The duration of applied stress

For example, consider a solder joint on the connecting pin of a vacuum tube device.
One characteristic of the joint is the degree to which the pin hole is filled with solder,
characterized as "percent fill." All other characteristics being acceptable, a joint that
is 100 percent filled offers the maximum mechanical strength, minimum resistance,
and greatest current carrying capacity. Conversely, a joint that is zero percent filled
has no mechanical strength, and only if the lead is touching the barrel does it have any
significant electrical properties. Between these two extreme cases are degrees of ab-
normality. For a fixed magnitude of applied stress:

·

A grossly abnormal solder joint probably will fail in a short time.

·

A moderately abnormal solder joint probably will fail, but after a longer period of
time than a grossly abnormal joint.

·

A mildly abnormal solder joint probably will fail, but after a much longer period
of time than in either of the preceding conditions.

Figure 9.9

illustrates this concept. A similar time-stress relationship holds for a fixed

degree of abnormality and variable applied stress.

A latent defect eventually will advance to a patent defect when exposed to environ-

mental, or other, stimuli. A patent defect is a flaw that has advanced to the point at
which an abnormality actually exists. To carry on the solder example, a cold solder joint
represents a flaw (latent defect). After vibration and/or thermal cycling, the joint (it is
assumed) will crack. The joint will now have become a detectable (patent) defect. Some

Figure 9.9 Estimation of the probable time to failure from an abnormal solder joint. (

Af-

ter [5].)

© 1999 CRC Press LLC

latent defects can be stimulated into patent defects by thermal cycling, some by vibra-
tion, and some by voltage cycling. Not all flaws respond to all stimuli.

There is strong correlation between the total number of physical and functional de-

fects found per unit of product during the manufacturing process, and the average latent
defect content of shipping product. Product- and process-design changes aimed at re-
ducing latent defects not only improve the reliability of shipping product, but also result
in substantial manufacturing cost savings.

9.3.4 Operating Environment

The operating environment of an electronic system, either because of external envi-
ronmental conditions or unintentional component underrating, may be significantly
more stressful than the system manufacturer or the component supplier anticipated.
Unintentional component underrating represents a design fault, but unexpected envi-
ronmental conditions are possible for many applications, particularly in remote loca-
tions.

Conditions of extreme low or high temperatures, high humidity, and vibration dur-

ing transportation may have a significant impact on long-term reliability of the system.
For example, it is possible--and more likely, probable--that the vibration stress of the
truck ride to a remote transmitting site will represent the worst-case vibration exposure
of the transmitter and all components within it during the lifetime of the product.

Manufacturers report that most of the significant vibration and shock problems for

land-based products arise from the shipping and handling environment. Shipping tends
to be an order of magnitude more severe than the operating environment with respect to
vibration and shock. Early testing for these problems involved simulation of actual
shipping and handling events, such as end-drops, truck trips, side impacts, and rolls
over curbs and cobblestones. Although unsophisticated by today's standards, these
tests are capable of improving product resistance to shipping-induced damage.

9.3.5 Failure Modes

Latent failures aside, the circuit elements most vulnerable to failure in any piece of
electronic hardware are those exposed to the outside world. In most systems, the
greatest threat typically involves one or more of the following components or subsys-
tems:

·

The ac-to-dc power supply

·

Sensitive signal-input circuitry

·

High-power output stages and devices

·

Circuitry operating into an unpredictable load, or into a load that may be exposed
to lightning and other transient effects (such as an antenna)

Derating of individual components is a key factor in improving the overall reliability

of a given system. The goal of derating is the reduction of electrical, mechanical, ther-

© 1999 CRC Press LLC

mal, and other environmental stresses on a component to decrease the degradation rate
and prolong expected life. Through derating, the margin of safety between the operat-
ing stress level and the permissible stress level for a given part is increased. This adjust-
ment provides added protection from system overstress, unforeseen during design.

9.3.6 Maintenance Considerations

The reliability and operating costs over the lifetime of an RF system can be affected
significantly by the effectiveness of the preventive maintenance program designed
and implemented by the engineering staff. In the case of a critical-system unit that
must be operational continuously or during certain periods, maintenance can have a
major impact--either positive or negative--on downtime.

The reliability of any electronic system may be compromised by an enabling event

phenomenon. This is an event that does not cause a failure by itself, but sets up (or en-
ables) a second event that can lead to failure of the system. Such a phenomenon is insid-
ious because the enabling event may not be self-revealing. Examples include the fol-
lowing:

·

A warning system that has failed or has been disabled for maintenance

·

One or more controls that are set incorrectly, providing false readouts for opera-
tions personnel

·

Redundant hardware that is out of service for maintenance

·

Remote metering that is out of calibration

Common-Mode Failure

A common-mode failure is one that can lead to the failure of all paths in a redundant
configuration. In the design of redundant systems, therefore, it is important to iden-
tify and eliminate sources of common-mode failures, or to increase their reliability to
at least an order of magnitude above the reliability of the redundant system. Com-
mon-mode failure points in a high-power RF system include the following:

·

Switching circuits that activate standby or redundant hardware

·

Sensors that detect a hardware failure

·

Indicators that alert personnel to a hardware failure

·

Software that is common to all paths in a redundant system

The concept of software reliability in control and monitoring has limited meaning in

that a good program will always run, and copies of a good program will always run. On
the other hand, a program with one or more errors will always fail, and so will the cop-
ies, given the same input data. The reliability of software, unlike hardware, cannot be
improved through redundancy if the software in the parallel path is identical to that in
the primary path.

© 1999 CRC Press LLC

Spare Parts

The spare parts inventory is a key aspect of any successful equipment maintenance
program. Having adequate replacement components on hand is important not only to
correct equipment failures, but to identify those failures as well. Many parts--partic-
ularly in the high-voltage power supply and RF chain--are difficult to test under
static conditions. The only reliable way to test the component may be to substitute one
of known quality. If the system returns to normal operation, then the original compo-
nent is defective. Substitution is also a valuable tool in troubleshooting intermittent
failures caused by component breakdown under peak power conditions.

9.4

Vacuum Tube Reliability

The end user should keep an accurate record of performance for each tube at the facil-
ity. Shorter-than-normal tube life could point to a problem in the RF amplifier or re-
lated hardware. The average life that may be expected from a power tube is a function
of many operational parameters, including:

·

Filament voltage

·

Ambient operating temperature

·

RF power output

·

Operating frequency

·

Operating efficiency

The best estimate of life expectancy for a given system at a particular location comes
from on-site experience. As a general rule of thumb, however, at least 4000 hours of
service can be expected from most power grid tubes. Klystrons and other microwave
devices typically provide in excess of 15,000 hours. Possible causes of short tube life
include the following:

·

Improper power stage tuning

·

Inaccurate panel meters or external wattmeter, resulting in more demand from the
tube than is actually required

·

Poor filament voltage regulation

·

Insufficient cooling system airflow

·

Improper stage neutralization

An examination of the data sheet for a given vacuum tube will show that a number of

operating conditions are possible, depending upon the class of service required by the
application. As long as the maximum ratings of the device are not exceeded, a wide
choice of operating parameters is possible. When studying the characteristic curves of
each tube, remember that they represent the performance of a typical device. All elec-
tronic products have some tolerance among devices of a single type. Operation of a

© 1999 CRC Press LLC

given device in a particular system may be different than that specified on the data
sheet. This effect is more pronounced at VHF and above.

9.4.1 Thermal Cycling

Most power tube manufacturers recommend a warm-up period between the applica-
tion of filament-on and plate-on commands; about 5 minutes is typical. The minimum
warm-up time is 2 minutes. Some RF generators include a time-delay relay to prevent
the application of a plate-on command until a predetermined warm-up cycle is com-
pleted.

Most manufacturers also specify a recommended cool-down period between the ap-

plication of plate-off and filament-off commands. This cool-down, generally about 10
minutes, is designed to prevent excessive temperatures on the PA tube surfaces when
the cooling air is shut off. Large vacuum tubes contain a significant mass of metal,
which stores heat effectively. Unless cooling air is maintained at the base of the tube
and through the anode cooling fins, excessive temperature rise can occur. Again, the re-
sult can be shortened tube life, or even catastrophic failure because of seal cracks
caused by thermal stress.

Most tube manufacturers suggest that cooling air continue to be directed toward the

tube base and anode cooling fins after filament voltage has been removed to further
cool the device. Unfortunately, however, not all control circuits are configured to per-
mit this mode of operation.

9.4.2 Tube-Changing Procedure

Plug-in power tubes must be seated firmly in their sockets, and the connections to the
anodes of the tubes must be tight. Once in place, the device typically requires no
maintenance during its normal operating lifetime.

Insertion of a replacement tube must be performed properly. Gently rock and

slightly rotate the tube as it is being inserted into the socket. This will help prevent
bending and breaking the fingerstock. Be certain to apply sufficient force to seat the
tube completely into the socket. Never use a lever or other tool on the tube to set it in
place. Manual pressure should be adequate.

An intermediate point is reached when the grid contact fingerstock slides up the

tube sides and encounters the contact area. It is important that the tube is fully inserted
in the socket beyond this initial point of resistance.

Some sockets have adjustable stops that are set so that the tube grid contacts rest in

the middle of the contact area when the tube is fully inserted. This positioning can be
checked by inserting, then removing, a new tube. The scratch marks on the grid con-
tacts will indicate the position of the tube relative to the socket elements.

Figure 9.10

shows the minute scoring in the base contact rings of a power grid tube. A well-main-
tained socket will score the tube contacts when the device is inserted. If all fingers do
not make contact, more current will flow through fewer contact fingers, causing over-
heating and burning, as shown in

Figure 9.11

.

© 1999 CRC Press LLC

Poor contact between a tube and its socket will always lead to problems. If the socket

connectors fail to provide the proper electrical contact with the cylindrical tube ele-
ments, concentration of RF charging currents will result, leading to local overheating
that may be destructive. After the connector loses its spring action, the heating is aggra-
vated, and damage to the tube is likely. Some socket designs include a wire-wound
spring encircling the outer circumference of the fingerstock to increase the contact
pressure of the individual fingers. These springs should be replaced if they break or
lose tension.

When a tube is removed from its socket, the socket should be blown or wiped clean

and carefully inspected. Any discoloration caused by overheating of the socket
fingerstock could signal impending tube/socket failure.

Figure 9.10 Proper fingerstock scor-
ing of filament and control grid con-
tacts on a power tube. (

Courtesy of

Varian.)

Figure 9.11 Damage to the grid con-
tact of a power tube caused by insuffi-
cient fingerstock pressure. (

Courtesy

of Varian.)

© 1999 CRC Press LLC

Many industrial and low-frequency tubes are not socketed, but installed by bolted or

clamped connections. Stainless steel has a much lower coefficient of thermal expan-
sion than copper. Therefore, clamped stainless steel anode connections should have
some method of strain relief to prevent excess pressure from collapsing the tube anode
as it heats.

All bolted or screwed connections should be tight. Verify that the clamps are snug

and provide a good electrical contact around the entire circumference of the device. Be-
cause of the RF fields present, all clamps and bolts should be made from nonmagnetic
materials. Copper, brass, and nonmagnetic series 300 stainless steel fasteners are pre-
ferred. Stainless steel is not a good conductor of electricity, so, although it may be used
for clamping, it should not be part of the current path.

Before a new tube is installed in a system, it should be inspected for cracks or loose

connections (in the case of tubes that do not socket-mount). Manufacturers typically
recommend that after a new tube--or one that has been on the shelf for some time--is
installed in the transmitter, it should be run with filaments only for at least 30 minutes,
after which plate voltage may be applied. Next, the drive (modulation) is slowly
brought up, in the case of an amplitude-modulated system. Residual gas inside the tube
may cause an interelectrode arc (usually indicated as a plate overload) unless it is
burned off in such a warm-up procedure.

9.4.3 Power Tube Conditioning

Large power tubes are subjected to rigorous processing during exhaust pumping at the
time of manufacture [6]. Active elements are processed at temperatures several hun-
dred degrees Celsius higher than those expected during normal operation. This proce-
dure is designed to drive off surface and subsurface gas from the metals in the device
to lower the possibility of these gases being released during the service life of the
tube. Free gas molecules, however, always will be present to some degree in a fully
processed tube. Gas, particularly oxygen-containing compounds, can combine chem-
ically with cathode material to either permanently or temporarily destroy the electron
emission capability of the cathode. Free gas molecules, when struck by electrons
moving from the cathode to the anode, also may be ionized by having one or more
electrons knocked from their systems. If enough of these ions, plus freed electrons,
are present, a conduction path will be established that is not subject to control by the
grid. This can result in runaway arcing that may involve all elements of the device.
Current then will be limited primarily by the source voltage and impedance. (The
space charge, to some degree, is neutralized by the presence of both electrons and
positive ions.)

Electrons from sources other than the heated cathode provide low-current paths be-

tween elements when the voltage gradient is sufficiently high at the negative element
for pure field emission. The voltage gradient at the negative element is governed by the
following parameters:

·

The applied voltage between elements

·

The spacing between elements

© 1999 CRC Press LLC

·

The surface contour of the negative element

A large voltage gradient can exist in the following areas:

·

In front of a point on the negative element

·

In front of a particle adhering to the negative element

·

In front of a clump of gas molecules on the surface of the negative element

Field emission occurs readily from a cold surface if the operating conditions provide
the necessary voltage gradient.

Ionization of free gas can result from bombardment by field-emitted electrons.

Arcing is likely to occur as a result of field emission in operating equipment because
the plate voltage is maximum during that part of the signal cycle when ordinary plate
current from the cathode is shut off by the control grid. For this reason, an important
part of tube processing is high-voltage conditioning to remove sharp points or small
particles from tube elements. This part of tube processing may, and sometimes should,
be repeated in the field after shipment or storage if the tube is intended for use at a plate
voltage in excess of 10 kV.

High-voltage conditioning (sometimes called spot-knocking or debarnacling) con-

sists of applying successively higher voltages between tube elements, permitting the
tube to spark internally at each voltage level until stable (no arcing). The voltage then is
raised to the next higher potential until the tube is stable at a voltage approximately 15
percent higher than the peak signal voltage it will encounter in service.

The equipment used for tube conditioning is simple but specialized. It may provide

dc or ac voltage, or both. The current required is small. The voltage should be continu-
ously variable from practically zero to the highest value required for proper condition-
ing of the tube. The energy per spark is controlled by the internal resistance of the sup-
ply plus any external series resistor used. In dc conditioning, a milliammeter typically
is included to measure the level of field emission current prior to sparking, or simply to
determine whether the field emission current is within the specified range for a tube be-
ing tested. Also in dc conditioning, a capacitor may be placed across the tube under test
to closely control the energy released for each spark. If the conditioning is to proceed
effectively, the amount of energy delivered by the capacitor must be great enough to
condition the surfaces, but not so large as to cause permanent damage.

Power Supply

The power supply is the heart of any high-voltage tube conditioning station [6]. Cur-
rent-limiting resistance is desirable; the power supply may have an internal resistance
on the order of 100 k

. In addition to the power supply, a current-limiting resistor of

100 to 300 k

should be provided in the circuit to the tube being conditioned. In dc

processing, a storage capacitor of 1000 to 3000 pF can be placed directly across the
device under test (DUT), with a high-voltage noninductive resistor of approximately
50

connected between the tube and the capacitor. Naturally, this shunt capacitor

must be capable of withstanding the full applied potential of the power supply.

© 1999 CRC Press LLC

Using both polarities on the tube elements is advantageous in high-voltage condi-

tioning, suggesting that an ac supply might be the best choice for field tube condition-
ing. The advantage of the dc system, on the other hand, is that exact levels of field emis-
sion current and breakdown voltage may be determined for comparison with known
"clean" tubes, and with the manufacturer's specification. The polarity of the applied
voltage may be reversed by mounting the DUT on an insulating frame so that the posi-
tive and negative high-voltage leads may be interchanged during processing. Also, the
power supply rectifier and dc meters may be bypassed with suitable switching so that ac
is obtained from the supply for initial tube cleanup. The storage capacitor across the
tube should be disconnected if ac is used.

Conditioning Procedure

Manufacturers of large power tubes typically specify an upper limit for field emission
between the filament, control grid, and screen grid connected together, and the anode
[6]. Such devices should be allowed to spark until stable at the voltage where sparking
first occurs. The voltage then is increased in steps determined by the voltage that initi-
ates sparking again. This process is continued until the tube is stable at the maximum
test voltage. It is advisable to reverse the polarity of the supply if dc testing is em-
ployed.

Usually, no field emission specifications exist for applied voltage between grids and

between grid and filament, but these elements normally should hold off 10 kV inter-
nally. The need for conditioning between these elements is indicated if internal spark-
ing occurs at a significantly lower voltage. Consult the tube manufacturer for specific
recommendations on conditioning voltages.

During high-voltage processing, particularly between grids and between grid and

filament, some of the redistributed gas molecules may be deposited on the cold fila-
ment, causing a temporary loss of emission. If this happens, operate the tube for ap-
proximately 1 hour, with normal filament power only, to drive off the volatile material.
Normal electron emission from the filament will be reestablished by this procedure. It
is recommended that routine practice include operation of the filament for approxi-
mately 1 hour as an immediate follow-up to high-voltage conditioning.

After conditioning, the tube should be stored in the same position (orientation) as

that used for high-voltage conditioning and installation in the equipment. Movement of
the tube after conditioning should be minimized to prevent any redistribution of parti-
cles within the device.

Considerations for Very Large Tubes

High-voltage ac conditioning is not recommended for very large tubes, such as the
8971/X2177, 8972/X2176, 8973/X2170, 8974/2159, 4CM400000A, and similar de-
vices [6]. High-potential 50 or 60 Hz voltage can excite natural mechanical reso-
nances in the large filament and grid structures of these tube types. Excitation of these
resonances can break the filament and/or grids, thus destroying the device. If ac con-
ditioning must be used, the following precautions should be taken:

© 1999 CRC Press LLC

·

Connect all filament terminals together (external to the tube).

·

Connect all terminals of the tube to a terminal of the high-voltage power supply.
For example, if high voltage is to be applied between the screen and anode, the fil-
ament, grid, and screen should be connected together, then to one terminal of the
supply. If high voltage is to be applied between the filament and grid, connect the
grid, screen, and anode together and to one terminal of the supply. No terminal of
the tube should be allowed to float electrically during high-voltage conditioning.

·

As the test voltage is increased, listen carefully for vibration inside the tube. If
any is detected, immediately remove the ac high voltage.

These precautions will reduce the risk of damage from ac conditioning of very large
tubes. However, because of the special procedures required and the risk of damage re-
sulting from mechanical resonances, it is recommended that only dc conditioning be
used for such devices.

Safety

Personnel protection dictates that the tube being processed, as well as the power sup-
ply, be enclosed in a metal cabinet with access doors interlocked so that no high volt-
age can be applied until the doors are closed [6]. The enclosure material, wall thick-
ness, and dimensions must be chosen to limit X-ray radiation from the DUT to safe
levels. A steel-wall thickness of in has been found adequate for dc testing up to ap-
proximately 70 kV. For conditioning up to 100 kV dc, it is necessary to add 0.05 in of
lead sheet as a lining to the steel enclosure.

The field of health standards regarding x-radiation dosage for a given period of time

is complex and should be discussed with experts on the subject. Calibrated film badges
or other reliable measuring devices normally are used to determine human exposure to
X rays over time.

9.4.4 Filament Voltage

A true reading rms voltmeter is required for accurate measurement of filament volt-
age. The measurement is taken directly from the tube socket connections. A
true-reading rms meter, instead of the more common average responding rms meter,
is suggested because it can accurately measure a voltage despite an input waveform
that is not a pure sine wave. Some filament voltage regulators use silicon controlled
rectifiers (SCRs) to regulate the output voltage. Saturated core transformers, which
often deliver nonsine waveforms, also are commonly used to regulate filament input
power.

Long tube life requires filament voltage regulation. A tube whose filament voltage

is allowed to vary along with the primary line voltage will not achieve the life expec-
tancy possible with a tightly regulated supply. This problem is particularly acute at
mountain-top installations, where utility regulation is generally poor.

To extend tube life, some end-users leave the filaments on at all times, not shutting

down at sign-off. If the sign-off period is 3 hours or less, this practice can be beneficial.

© 1999 CRC Press LLC

Filament voltage regulation is a must in such situations because the primary line volt-
ages may vary substantially from the carrier-on to carrier-off value.

When a cold filament is turned-on, damage can be caused by two effects [7]:

·

Current inrush into the cold filament, which can be up to 10 times the normal op-
erating current if the filament supply is of low impedance.

·

Grain reorientation, which occurs at about 600 to 700 ºC, known as the
Miller-Larson effect. This grain reorientation will result in a momentary plastic
state that can cause the filament wire to grow in length and, therefore, become
thinner.

The Miller-Larson effect is aggravated by variations in the cross-sectional area of the
filament wires along their length. This will cause hot spots in the thinner sections be-
cause of the greater current densities there. The higher temperatures at the hot spots
cause increased growth during warmup through the 600 ºC temperature range, when
they have a higher current density than the rest of the wire, and a much higher power
dissipation per unit length as a result of the higher resistance. Each time the filament
is turned-on, the wire becomes thinner until the hot spot temperature enters a runaway
condition.

Filament voltage for a klystron should not be left on for a period of more than 2 hours

if no beam voltage is applied. The net rate of evaporation of emissive material from the
cathode surface of a klystron is greater without beam voltage. Subsequent condensa-
tion of the material on gun components may lead to voltage holdoff problems and an in-
crease in body current.

Black Heat

One practice used occasionally to extend the life of the thoriated tungsten power tube
filament is to operate the device at a low percentage of rated voltage during sign-off
periods [7]. This technique, known as black heat, involves operating the filament at
25 to 40 percent of the rated filament voltage. This range produces a temperature high
enough to maintain the filament above the 600 ºC region where the Miller-Larson ef-
fect occurs, and sometimes low enough so that forced cooling is not required during
stand-by.

Operating histories of several black heat installations indicate that the trauma of

raising the filament voltage from 25 percent up to the operating voltage is still poten-
tially damaging. For this reason, a higher stand-by voltage of about 80 to 90 percent of
the full rated value (orange heat) has been used with some success.

The voltage range of 40 to 80 percent should be avoided because the filament can act

as a getter within this range, and absorb gases that will raise the work function of the
thoriated tungsten filament. For orange heat operation, cooling system operation is re-
quired. Before attempting to implement a black heat or orange heat filament manage-
ment program, consult with the tube and transmitter manufacturers.

© 1999 CRC Press LLC

9.4.5 Filament Voltage Management

As outlined in Chapter 3, the thoriated-tungsten power grid tube lends itself to life ex-
tension through management of the applied filament voltage. Accurate adjustment of
the filament voltage can extend the useful life of the device considerably, sometimes
to twice the normal life expectancy. The following procedure typically is recom-
mended:

·

Install the tube, and tune for proper operation.

·

Operate the filament at its full rated voltage for the first 200 hours following in-
stallation.

·

Following the burn-in period, reduce the filament voltage by 0.1 V per step until
system power output begins to fall (for frequency-modulated systems) or until
modulating waveform distortion begins to increase (for amplitude-modulated
systems).

·

When the emissions floor has been reached, increase the filament voltage approx-
imately 0.2 V peak-to-peak (P-P).

Long-term operation at this voltage can result in a substantial extension of the useful
life of the tube, as illustrated in

Figure 9.12

.

The filament voltage should not, in any event, be operated at or below 90 percent of

its rated value. At regular intervals, about every 3 months, the filament voltage is
checked and increased if power output begins to fall or waveform distortion begins to
rise. Filament voltage never should be increased to more than 105 percent of the rated
voltage.

Note that some tube manufacturers place the minimum operating point at 94 per-

cent. Others recommend that the tube be set for 100 percent filament voltage and left
there. The choice is left to the user, after consultation with the manufacturer.

The filament current should be checked when the tube is installed, and at annual in-

tervals thereafter, to ensure that the filament draws the expected current. Tubes can fail
early in life because of an open filament bar--a problem that could be discovered dur-
ing the warranty period if a current check is made upon installation.

For 1 week of each year of tube operation, the filament should be run at full rated

voltage. This will operate the getter and clean the tube of gas.

Klystron Devices

Filament voltage is an equally important factor in achieving long life in a klystron.
The voltage recommended by the manufacturer must be accurately set and checked
on a regular basis. Attempts have been made to implement a filament voltage man-
agement program--as outlined previously for a thoriated-tungsten tube--on a kly-
stron, with varying degrees of success. If filament management is attempted, the fol-
lowing guidelines should be observed:

© 1999 CRC Press LLC

Figure 9.12 The effects of filament voltage management on a thoriated-tungsten fila-
ment power tube: (

a) useful life as a function of filament voltage, (b) available power as a

function of filament voltage. Note the dramatic increase in emission hours when filament
voltage management is practiced.

(

b)

(

a)

© 1999 CRC Press LLC

·

Install the tube, and tune for proper operation.

·

Operate the filament at its full rated voltage for the first 200 hours following in-
stallation.

·

Following the burn-in period, reduce the filament voltage by 0.1 V per step until
collector current begins to decrease. After each filament voltage change, wait at
least 5 minutes to allow for cathode cooling.

·

When the emissions floor has been reached, increase the filament voltage approx-
imately 0.25 V P-P.

Figure 9.13

charts the relationship between emitter temperature and beam current.

The filament should be adjusted to a point between E

f1

and E

f2

, which will vary with

each klystron. Note that continued operation below the E

f1

level, or at a point that

causes emission instability, can result in beam defocusing, leading to increased modu-
lating anode current and body current.

For operation at a reduced filament voltage, it may be necessary to allow a longer

warm-up period when bringing the tube up from a cold start. If loss of emission is noted
at turn-on, it will be necessary to either extend the warm-up time or raise the filament
voltage to overcome the loss.

9.4.6 PA Stage Tuning

The PA stage of an RF generator may be tuned in several ways. Experience is the best
teacher when it comes to adjusting for peak efficiency and performance. Compro-
mises often must be made among various operating parameters.

Figure 9.13 The effect of klystron emitter temperature on beam current.

© 1999 CRC Press LLC

Tuning can be affected by any number of changes in the PA stage. Replacing the fi-

nal tube in a low- to medium-frequency RF generator usually does not significantly al-
ter stage tuning. It is advisable, however, to run through a touch-up tuning procedure
just to be sure. Replacing a tube in a high-frequency RF generator, on the other hand,
can significantly alter stage tuning. At high frequencies, normal tolerances and varia-
tions in tube construction result in changes in element capacitance and inductance.
Likewise, replacing a component in the PA stage may cause tuning changes because of
normal device tolerances.

Stability is one of the primary objectives of tuning. Adjust for broad peaks or dips, as

required. Tune so that the system is stable from a cold startup to normal operating tem-
perature. Readings should not vary measurably after the first minute of operation.

Tuning is adjusted not only for peak efficiency, but also for peak performance.

These two elements, unfortunately, do not always coincide. Tradeoffs must sometimes
be made to ensure proper operation of the system.

9.4.7 Fault Protection

An arc is a self-sustained discharge of electricity between electrodes with a voltage
drop at the cathode on the order of the minimum ionizing potential of the environment
[8]. The arc supports large currents by providing its own mechanism of electron emis-
sion from the negative terminal, plus space-charge neutralization.

All power vacuum tubes operate at voltages that can cause severe damage in the

event of an internal arc unless the tube is properly protected. This damage can be rela-
tively minor in tubes operating from low-energy power supplies, but it will become cat-
astrophic in cases where a large amount of stored energy or follow-on current is in-
volved. Because any high-voltage vacuum device may arc at one time or another, some
means of protection is advised in all cases; it is mandatory in those instances where de-
structive quantities of energy are involved. In addition to protecting the tube in the
event of a tube or circuit malfunction, such measures also protect the external circuitry.

Most power grid tubes employing thoriated-tungsten filaments are capable of with-

standing relatively high energy arcs. However, while employing substantial materials,
such devices usually incorporate fine wire grids that must be protected. In large tube
applications, it is often found that the screen grid or bias supply, in addition to the anode
supply, is capable of delivering destructive energy to the tube structure in the event of an
internal arc. Consequently, protection must be provided in all cases where potentially
destructive energy is involved. Large tubes usually are capable of withstanding up to 50
joules (J) total energy without permanent damage.

In the design of a power amplifier, the stage should be tested to ensure that the 50 J

limit is not exceeded by any of the power supplies feeding the tube. Each supply,
whether anode or grid, should be short-circuit-tested through a 6-in length of
0.255-mm-diameter (no. 30 AWG) soft copper wire. A similar test should be made
from plate-to-screen grid to ensure that the tube is protected in the event of a
plate-to-screen arc. A protective spark gap between the screen grid and cathode is rec-
ommended so that, if an arc occurs, the resulting spark from screen to cathode is exter-
nal to the tube structure. The test wire will remain intact if the energy dissipated is less

© 1999 CRC Press LLC

than 50 J. The 50 J test includes total energy delivered from the energy storage device
and/or from the follow-on current from the power supply prior to the opening of the pri-
mary circuit.

Each power supply in the system must pass this test, or some form of energy-divert-

ing or energy-limiting circuit must be employed.

Figure 9.14

shows an energy-divert-

ing circuit for a high-power tube. Energy diverters at high-power levels are complex.
They typically employ large high-speed diverting elements such as thyratrons,
ignitrons, triggered spark gaps, and, in some cases, a high-speed vacuum switch. These
devices may be required to handle thousands of amperes of peak current. Resistor R1 in

Figure 9.14

is used to limit the peak current into the energy-diverting switch V1, as well

as to limit peak current demand on the transformer and rectifiers in the power supply.
Resistor R2 must be large enough to keep the voltage across V1 from dropping so low
under load-arc conditions as to inhibit the proper firing of the fault-diverting elements.
R1 and R2 must also damp circuit ringing sufficiently to prevent current reversal in V1,
which could cause V1 to open before the fault conditions are cleared. R1 and R2 are
sized to minimize average power losses and to handle the required voltage and power
transients while properly performing their functions. The sensing element CT1 detects
any abnormal peak-current levels in the system. Signals from the current transformer
are used to trigger V1 in the event of an overcurrent condition. A crowbar test switch,
S1, can be used to perform the wire test outlined previously, to ensure that the total en-
ergy dissipated in an arc is less than 50 J. Typically, S1 is a remote-controlled high-volt-
age switch of either air or vacuum construction.

In a typical test, when S1 is closed, current transformer CT1 senses the fast rising

current and fires the energy-diverting element V1; it simultaneously sends a fast trip
signal to the main circuit breaker. Most energy diverters fire within a few microseconds
of the time the fault is detected; all voltages and currents are removed from the pro-
tected device in tens of microseconds.

Figure 9.14 Arc protection circuit utilizing an energy-diverting element.

© 1999 CRC Press LLC

By using this technique, it is possible to limit the energy delivered during an arc to

much less than the recommended maximum. In a properly operating system, essen-
tially no damage will occur within the tube even under severe arcing conditions. The
test wire must be connected at the tube itself to take into account all stored energy (L
and C) in the system.

9.4.8 Vacuum Tube Life

The vacuum tube suffers wear-out because of a predictable chemical reaction. Life
expectancy is one of the most important factors to be considered in the use of vacuum
tubes. In general, manufacturers specify maximum operating parameters for power
grid tubes so that operation within the ratings will provide for a minimum useful life
of 4000 hours. Considerable variation can be found with microwave power tubes, but
15,000 hours is probably a fair average useful life. It should be noted that some micro-
wave devices, such as certain klystrons, have an expected useful life of 30,000 hours
or more.

The cathode is the heart of a power tube. The device is said to wear out when fila-

ment emissions are inadequate for full power output or acceptable waveform distortion.
In the case of the common thoriated-tungsten filament tube, three primary factors de-
termine the number of hours a device will operate before reaching wear-out:

·

The rate of evaporation of thorium from the cathode

·

The quality of the tube vacuum

·

The operating temperature of the filament

In the preparation of thoriated tungsten, 1 to 2 percent of thorium oxide (thoria) is

added to the tungsten powder before it is sintered and drawn into wire form. After being
mounted in the tube, the filament usually is carburized by being heated to a tempera-
ture of about 2000 K in a low-pressure atmosphere of hydrocarbon gas or vapor until its
resistance increases by 10 to 25 percent. This process allows the reduction of the thoria
to metallic thorium. The life of the filament as an emitter is increased because the rate
of evaporation of thorium from the carburized surface is several times smaller than
from a surface of pure tungsten.

Despite the improved performance obtained by carburization of a thoriated-tung-

sten filament, the element is susceptible to deactivation by the action of positive ions.
Although the deactivation process is negligible for anode voltages below a critical
value, a trace of residual gas pressure too small to affect the emission from a pure tung-
sten filament can cause rapid deactivation of a thoriated-tungsten filament. This re-
striction places stringent requirements on vacuum processing of the tube.

These factors, taken together, determine the wear-out rate of the device. Cata-

strophic failures due to interelectrode short circuits or failure of the vacuum envelope
are considered abnormal and are usually the result of some external influence. Cata-
strophic failures not the result of the operating environment usually are caused by a de-
fect in the manufacturing process. Such failures generally occur early in the life of the
component.

© 1999 CRC Press LLC

The design of the RF equipment can have a substantial impact on the life expectancy

of a vacuum tube. Protection circuitry must remove applied voltages rapidly to prevent
damage to the tube in the event of a failure external to the device. The filament turn-on
circuit also can affect PA tube life expectancy. The surge current of the filament circuit
must be maintained below a certain level to prevent thermal cycling problems. This
consideration is particularly important in medium- and high-power PA tubes. When the
heater voltage is applied to a cage-type cathode, the tungsten wires expand immedi-
ately because of their low thermal inertia. However, the cathode support, which is made
of massive parts (relative to the tungsten wires), expands more slowly. The resulting
differential expansion can cause permanent damage to the cathode wires. It also can
cause a modification of the tube operating characteristics and, occasionally, arcs be-
tween the cathode and the control grid.

Catastrophic Failures

Catastrophic failures can be divided into two primary categories:

·

The short-circuiting of broken or warped elements within the tube

·

A loss of vacuum in the device

Air in the tube causes a loss of dielectric standoff between the internal elements. It is
not possible to distinguish between an interelectrode short circuit and a loss of vac-
uum in the operating equipment.

Catastrophic failures that occur during initial installation are usually the result of

broken elements. Failures that occur after an initial burn-in period are more likely to be
caused by a loss of vacuum. In either case, continued efforts to operate the tube can re-
sult in considerable damage to the device and to other components in the system.

Note that a loss of vacuum can result from any of several anomalies. The two most

common are:

·

A latent manufacturing defect in one of the ceramic-to-metal seals

·

Circulating currents resulting from parasitic oscillations and/or improper tuning

Intermittent overloads are the most difficult to identify. They can be caused by cir-

cuit operating conditions or internal tube failures. A broken or warped filament ele-
ment, for example, can move around inside the device and occasionally short circuit to
the grid, causing a loss of grid bias and subsequent plate overload. Intermittent over-
loads also can be caused by a short circuit or high VSWR across the load.

9.4.9 Examining Tube Performance

Examination of a power tube after it has been removed from a transmitter or other
type of RF generator can reveal a great deal about how well the equipment-tube com-
bination is working. Contrast the appearance of a new power tube, shown in

Figure

9.15

, with a component at the end of its useful life. If a power tube fails prematurely,

© 1999 CRC Press LLC

the device should be examined to determine whether an abnormal operating condition
exists within the transmitter. Consider the following examples:

·

Figure 9.16

. Two 4CX15000A power tubes with differing anode heat-dissipation

patterns. Tube (a) experienced excessive heating because of a lack of PA com-
partment cooling air or excessive dissipation because of poor tuning. Tube (b)
shows a normal thermal pattern for a silver-plated 4CX15000A. Nickel-plated
tubes do not show signs of heating because of the high heat resistance of nickel.

·

Figure 9.17

. Base-heating patterns on two 4CX15000A tubes. Tube (a) shows ev-

idence of excessive heating because of high filament voltage or lack of cooling air
directed toward the base of the device. Tube (b) shows a typical heating pattern
with normal filament voltage.

·

Figure 9.18

. A 4CX5000A tube with burning on the screen-to-anode ceramic.

Exterior arcing of this type generally indicates a socketing problem, or another
condition external to the tube.

·

Figure 9.19

. The stem portion of a 4CX15000A tube that had gone down to air

while the filament was on. Note the deposits of tungsten oxide formed when the
filament burned up. The grids are burned and melted because of the ionization
arcs that subsequently occurred. A failure of this type will trip overload breakers
in the RF generator. It is indistinguishable from a short-circuited tube in opera-
tion.

·

Figure 9.20

. A 4CX15000A tube that experienced arcing typical of a bent

fingerstock, or exterior arcing caused by components other than the tube.

F i g u r e 9 . 1 5

A n e w , u n u s e d

4CX15000A tube. Contrast the ap-
pearance of this device with the tubes
shown in the following figures. (

Cour-

tesy of Varian.)

© 1999 CRC Press LLC

Figure 9.16 Anode dissipation patterns on two 4CX15000A tubes: (

a) excessive heat-

ing, (

b) normal wear. (Courtesy of Econco Broadcast Service.)

(

a) (b)

Figure 9.17 Base heating patterns on two 4CX15000A tubes: (

a) excessive heating, (b)

normal wear. (

Courtesy of Econco.)

(

a) (b)

© 1999 CRC Press LLC

Note that the previous examples apply to silver-plated power grid tubes.

The external metal elements of a tube typically are plated with nickel or silver. Tubes

that go into sockets are usually silver-plated. The soft silver provides a better contact in-
terface than the much harder nickel, and it deforms slightly under contact pressure, pro-
viding greater contact area. Silver plating has a dull, whitish cast, whereas nickel has a
hard, metallic appearance.

Nickel is resistant to discoloration due to heat at normal tube operating tempera-

tures, but silver will tarnish easily. As demonstrated in this section, the heat patterns on

Figure 9.18 A 4CX5000A tube that appears to
have suffered socketing problems. (

Courtesy of

Econco.)

Figure 9.19

The interior elements of a

4CX15000A tube that had gone to air while the fila-
ment was lit. (

Courtesy of Econco.)

© 1999 CRC Press LLC

silver-plated tubes are helpful in problem analysis. If a nickel-plated tube shows signs
of heat discoloration, a significant cooling or operating problem exists. Nickel will not
discolor until it reaches a temperature much higher than a tube normally achieves under
typical operating conditions.

9.4.10 Shipping and Handling Vacuum Tubes

Because of their fragile nature, vacuum tubes are packaged for shipment in
foam-filled or spring-supported shipping containers. When it is necessary to trans-
port a tube from one location to another, use the original packing material and box or
crate. The tube should be removed whenever the RF equipment is relocated. Never
leave a tube installed in its socket during an equipment move.

During installation or service, always keep the tube in its typical operating position

(most commonly horizontal). Never allow a tube to roll along a surface. Damage to the
filament or to other elements in the device may result.

It is occasionally necessary for an RF generator to be moved from one location to an-

other within a plant. Equipment used in this manner must be equipped with air-filled
casters; do not use solid casters.

When it is necessary or desirable to store a tube for an extended period of time, place

it in a plastic bag to protect it from moisture, and return the device to its shipping box.
Maintenance engineers are discouraged from placing markings on a power tube. Some
engineers note service dates using ink or pencil on tubes. This practice is not recom-
mended. Instead, attach a separate service record to the tube using a tie-wrap or similar
removable device. Do not use self-adhesive labels for this purpose. When removed, ad-
hesive labels usually leave some residue on the attachment surface.

Although it is not recommended that a power tube be stored indefinitely, most mod-

ern devices utilizing metal and ceramic envelopes can be stored for long periods of time

Figure 9.20 A 4CX15000A tube showing
signs of external arcing. (

Courtesy of

Econco.)

© 1999 CRC Press LLC

without deterioration. It usually is not necessary to rotate a spare tube through an oper-
ating socket to degas the device.

Older vacuum tubes utilizing glass as the insulating medium tend to leak gas over

time. The glass is not the problem, but, instead, the Kovar alloy used to seal the
glass-to-metal parts of the tube. Kovar also is subject to rusting when moisture is pres-
ent. Glass-insulated vacuum tubes should be sealed in a plastic bag for storage and ro-
tated through the equipment at least once every 12 months. The larger the physical size
of the tube, the greater the problem of gassing.

9.5

Klystron Reliability

Klystron amplifiers are well known as reliable, long-life microwave devices that pro-
vide the user thousands of hours of trouble-free service. Many of the guidelines dis-
cussed in Section 9.4 for power vacuum tubes in general also apply to klystrons and
related microwave devices. However, several considerations are specific to the kly-
stron, including:

·

Cooling system maintenance

·

Ceramic element cleanings

·

Gun element reconditioning

·

Focusing electromagnet maintenance

·

Protection during prime power interruption

·

Shutoff precautions

9.5.1 Cleaning and Flushing the Cooling System

One of the major problems encountered in field operation of a klystron cooling sys-
tem is the formation of corrosion products [9]. Corrosion frequently occurs in the
cooling channels of the klystron body on the surface of the water/vapor-cooled collec-
tor, and in the cooling channels of the electromagnet. It is the result of a chemical re-
action between free oxygen-laden coolant and the hot copper channel wall. Corro-
sion, as described here, is different from scale formation, which can be caused by the
use of a coolant other than clean distilled water, or may be the by-product of electroly-
sis between dissimilar metals in the heat-exchange system. Scale formation can be
controlled through the use of on-line purification loops.

New transmitter water lines frequently contain contaminants, including:

·

Solder

·

Soldering or brazing flux

·

Oils

·

Metal chips or burrs

© 1999 CRC Press LLC

·

Pieces of Teflon sealing tape

When the water lines are installed, these contaminants must be flushed and cleaned
from the system before the klystron and magnet are connected.

The following guidelines are suggested for cleaning the cooling passages of water-

and vapor-cooled klystrons and associated RF generation equipment. The procedures
should be performed before a klystron is installed, then repeated on a periodic basis
while the tube is in service.

Transmitter Flushing

Proper cleaning of the circulating water system in an RF generator is vital to
long-term reliability. Although each design is unique, the following general proce-
dures are appropriate for most installations [9]:

·

Disconnect the tube and the magnet.

·

Add jumper hoses between the input and output of the klystron and the electro-
magnet water lines.

·

Disconnect or bypass the pump motor.

·

Fill the system with hot tap water, if available.

·

Open the drain in the transmitter cabinet and flush for 15 minutes or until it is
clean.

·

Flush the water lines between the tank and pump separately with hot tap water, if
available, until they are clean.

·

Connect all water lines, and fill the system with hot tap water, if available, and
one cup of nonsudsing detergent. Trisodium phosphate is recommended.

·

Operate the water system with hot tap water for 30 minutes. An immersion heater
may be used to maintain hot water in the system.

·

Drain the system, and flush with hot tap water for 30 minutes.

·

Remove and clean the filter element.

·

Refill the water system with tap water (at ambient temperature).

·

Operate the water system. Maintain the water level while draining and flushing
the system until no detergent, foam, or foreign objects or particles are visible in
the drained or filtered element. To test for detergents in the water, drain a sample
of water into a small glass test tube, and allow it to stand for 5 minutes. To gener-
ate foam, shake the test tube vigorously for 15 s, then allow it to stand 15 s. A
completely foam-free surface indicates no foam-producing impurities.

·

Repeat the two previous steps if detergent foam is still present.

·

Drain the system and refill with distilled water when the tube and transmitter wa-
ter lines are clean.

© 1999 CRC Press LLC

·

Remove, clean, and replace the filter before using.

Flushing Klystron Water Lines

Although most klystrons are shipped from the manufacturer with water passages
clean and dry, it is good engineering practice to flush all cooling passages before in-
stalling the tube. Occasionally, tubes that have been in service for some time develop
scale on the collector and also must be flushed. Contaminated water further contrib-
utes to dirty water lines, requiring flushing to clear. The following general procedure
is recommended for tubes with contaminated water lines, corrosion, scale, or blocked
passages [9]:

·

Remove the input water fitting (Hansen type), and add a straight pipe extension,
approximately 1 to 2 in long, to the tube.

·

Attach a hose to this fitting, using a hose clamp to secure it. This connection will
serve as the drain line; it should be discharged into a convenient outlet.

·

On some tube types, the normal body-cooling output line is fed to the base of the
vapor-phase boiler. Remove the hose at the base of the boiler. Do not damage this
fitting; it must be reused and must seal tightly.

·

Attach a 2 to 3 in-long extension pipe that will fit the small hose at one end and a
garden hose at the other end. Secure the extension with hose clamps.

·

Connect the garden hose to a tap water faucet. Hot tap water is preferable, if avail-
able.

·

Back-flush the klystron-body cooling passages for 10 to 15 minutes at full pres-
sure until they are clean.

·

Reconnect the input and output water lines to the klystron.

·

If scale is present on the vapor-phase collector, use a solution of trisodium phos-
phate for the first cleaning. Perform this step after cleaning the transmitter water
system (as described previously).

·

Connect the klystron to the transmitter water lines, and fill the system with tap
water (hot, if available) and one cup of nonsudsing detergent.

·

Operate the water system for 15 minutes. Make certain that the water level com-
pletely covers the collector core.

·

Drain the system, and flush for 30 minutes or until no detergent foam is present.

·

Remove and clean the filter element.

·

Fill the system with distilled water.

·

Drain the system, and refill with distilled water.

© 1999 CRC Press LLC

Cleaning Klystron Water Lines

If heavy scaling has formed on the vapor-phase collector and/or blocked water pas-
sages, they may be cleaned using a stronger cleaning agent. A solution of Ty-Sol

1

may

be used to remove scale and corrosion. The following procedure is recommended [9]:

·

Clean the transmitter water system (as described previously).

·

Connect the tube to the transmitter water system.

·

Fill the system with hot tap water (if available) and add 2 gallons of cleaning solu-
tion for every 50 gallons of water.

·

Operate the water system for 15 minutes, or until scale has been removed and the
collector has a clean copper color.

·

Make certain that the water level covers the top of the collector during cleaning.
Evidence of cleaning action may be observed in the flow meters.

·

Drain the system, and flush with tap water.

·

Remove and clean the filter element, then replace.

·

Refill the system with distilled water and flush for 30 minutes to 1 hour.

·

Check for detergent foam at the end of the flushing period; also check the pH fac-
tor. Continue flushing with distilled water if foam is present or if the pH factor is
not within the specified range.

·

Refill the system with clean distilled water.

Chlorine present in common tap water is harmful to the water passages of the kly-
stron. Thorough flushing with distilled water will remove all chlorine traces. Never
use tap water for final refill or for make-up water.

Flushing and Cleaning Magnet Water Lines

The magnet water lines may be flushed and cleaned in the same manner as described
for the klystron.

General Cleaning

Two remaining items must be clean to achieve efficient operation: the sight glass and
float of the water-flow indicators. The water-flow indicators often become contami-
nated during use. This contamination collects on the sight glass and on the float, mak-
ing flow-reading difficult. If too much contamination is present on the glass and float,
sticking or erroneous readings may result. The detergent and cleaning solutions may

1

Chemical Research Association, Chicago, IL.

© 1999 CRC Press LLC

not remove all of this contamination. If this is the case, remove and clean the flow me-
ter using a brush to clean the glass surface.

9.5.2 Cleaning Ceramic Elements

Foreign deposits on the ceramic elements of a klystron should be removed periodi-
cally to prevent ceramic heating or arc paths that can lead to klystron failure. The rec-
ommended procedure is as follows:

·

Remove the klystron from the RF generator, and mount it to a stand or other suit-
able fixture.

·

Remove loose debris with a soft-fiber bristle brush.

·

Clean the ceramic with a mild abrasive cleanser, preferably one that includes a
bleaching agent. (Household cleansers such as Comet and Ajax are acceptable.)
While cleaning, do not apply excessive force to the ceramic. Normal hand pres-
sure is sufficient.

·

Flush the ceramic with clean water to remove the cleanser.

·

Flush a second time with clean isopropyl alcohol.

·

Air dry the device. Be certain that all moisture has evaporated from the tube be-
fore returning it to service.

9.5.3 Reconditioning Klystron Gun Elements

During the operational life of a klystron, voltage standoff problems may be experi-
enced when the operating mode of the device is changed from one collector voltage to
a higher voltage, or from continuous to pulsed service [10]. In both of these cases, the
difference of potential between the internal elements is increased over values used in
the previous mode of operation. In some instances, this elevated difference of poten-
tial can lead to internal arcing as a result of long-term buildup of electrical leakage
between elements. This is a part of the normal aging process in klystrons. In many
cases, however, this particular process can be reversed through high-voltage condi-
tioning. Maintenance engineers have found it useful to clear any residual leakage be-
tween the various elements before pulsing an older klystron, or when increasing the
operating potentials applied to an older klystron.

Klystron electron gun leakage can be checked, and, in many cases, the gun elements

reconditioned, by applying a high dc potential between these elements. The process is
termed hi-potting. It requires a current-limited (5 mA maximum), continuously-vari-
able dc supply with a range of 0 to 30 kV. The procedure is as follows:

·

Turn off the heater, and allow the cathode sufficient time to cool completely (ap-
proximately 1 to 2 hours).

·

Remove all electrical connections from the klystron, except for the body and col-
lector grounds.

© 1999 CRC Press LLC

·

Connect the negative high-voltage output of the hi-potter to the heater-cathode
terminal, and connect the positive ground lead to the mod-anode as shown in

Fig-

ure 9.21

. Increase the voltage output gradually, noting the leakage current.

·

If high levels of current are observed, reduce the voltage and allow it to stand at
the 1 to 2 mA level until the leakage current reduces (burns off).

·

Continue to raise the hi-potter output until the voltage applied is equal to the nor-
mal operating value for the klystron being checked. The typical leakage level is
200 to 400

µA.

·

Check the mod-anode-to-body ground leakage; reduce, if necessary, in the same
fashion. For this purpose, the negative high-voltage output is connected to the
mod-anode terminal and the positive ground lead to the klystron body.

The leakage level indicated as optimum for this procedure (200 to 400

µA) is a typical

value. Tubes still should operate normally above this level, but they may be more sus-
ceptible to internal arcing.

9.5.4 Focusing Electromagnet Maintenance

The focusing electromagnet of a klystron, given sufficient preventive maintenance,
will provide many years of service [11]. Maintenance personnel can extend the life-
time of the device by reducing the operating temperature of the electromagnet wind-

Figure 9.21 Electrical connections for reconditioning the gun elements of a klystron.

© 1999 CRC Press LLC

ings and by minimizing thermal cycling. Field experience has shown that the major
failure modes for a focusing electromagnet are short-circuited turns in the coil wind-
ings and/or short circuits from the windings to the coolant tubes.

Short-circuited turns result in lower magnetic flux density available to focus the kly-

stron beam and are indicated by higher klystron body current. To counteract this effect,
the transmitter operator often will increase the electromagnet current to a point above
the rating necessary to obtain the magnetic field strength needed for rated klystron
body current. This increases the heat dissipation in the device, which may lead to fur-
ther problems. Such failures are the result of insulation breakdown between turns in the
windings, and/or between the windings and the coolant tubes. Insulation breakdown
usually is caused by excessive operating temperatures in the individual windings. Ex-
cessive temperature is the result of one or more of the following operational conditions:

·

Excessive inlet coolant temperature

·

Insufficient coolant flow

·

Excessive ambient temperature

·

Excessive operating current

Insulation breakdown also may result from excessive thermal cycling of the coil
windings. Abrasion of internal insulation may occur during expansion and contrac-
tion of the winding assembly. Generally speaking, the life of a focusing electromagnet
can be doubled for every 10°C drop in operating temperature (up to a point).

To minimize the effects of thermal cycling, the electromagnet temperature should

be maintained at or below its operating value during standby periods by running at rated
or reduced current, and at rated coolant flow. If neither running nor reduced current is
available during standby conditions, heated coolant should be circulated through the
electromagnet.

9.5.5 Power Control Considerations

Under certain conditions, high-voltage transients that exceed the specified maximum
ratings of a klystron can occur in an RF generator. These transients can cause
high-voltage arcing external to the tube across the klystron modulating anode and
cathode seals. In some cases, such arcs can puncture sealing rings, letting the tube
down to air. High-voltage transient arcing is not common to all installations, but when
it does occur, it can result in a costly klystron failure. Overvoltage transients in power
circuits can be traced to any of several causes, including lightning.

Collection of moisture on the tube is another potential cause of arc-induced failure.

Integral-cavity klystrons, which are installed with the cathode down, may collect mois-
ture in the high-voltage (cathode) area because of water leakage or condensation, lead-
ing to external arcing and loss of vacuum.

To provide a measure of protection against these types of failures, arc shields may be

used on some tubes to protect the gun components that are the most likely to be dam-
aged by arcs. The shield typically consists of an aluminum cover fitted on a portion of

© 1999 CRC Press LLC

the tube. Although arc shields provide considerable protection, they cannot prevent all
failures; the magnitude of a potential arc is an unknown quantity. It must be understood
that an arc shield does not eliminate the cause of arcing, which should be addressed sep-
arately by the maintenance engineer. It is, instead, intended as a stop-gap measure to
prevent destruction of the tube until the root cause of the arcing is eliminated.

Installation of a gas-filled spark-gap transient voltage suppressor also can serve to

protect the klystron in the event of an arc or other transient discharge. The rating of the
suppressor is determined by the type of tube being protected. The device is connected
between ground and the cathode, installed as close as possible to the cathode. The sup-
pressor will prevent the application of transient voltages that exceed the standoff rating
of the klystron ceramic.

Power-line suppression components are useful for protecting transmission equip-

ment from incoming line transients and other disturbances, but may not be effective in
protecting against a system-induced arc within the transmitter or power supply that
could damage or destroy the klystron. Therefore, protection of the klystron itself is rec-
ommended.

Primary Power Interruption

Unexpected interruptions of primary power, which occur seasonally during winter
storms and summer lightning strikes, can damage or destroy a klystron unless ade-
quate protection measures are taken [12]. There are three main factors involved in
such failures:

·

Mechanical and/or electrical malfunction of the main circuit breaker of the trans-
mitter. This condition results when the breaker remains closed after the primary
power is interrupted. With the loss of primary power, all excitation to the trans-
mitter is removed. If the main circuit breaker remains closed when the power is
restored, high voltage can be applied to the klystron before the magnet supply can
provide adequate focusing. This allows full beam power dissipation in the body
of the klystron, which can melt the drift tube.

·

Loss of focusing during shutdown resulting from a power interruption at the
transmitter in a system that does not employ magnet power supply filtering.
When a loss of primary power occurs, the various power supplies start to dis-
charge. The time constant of the associated filter circuit determines the rate of dis-
charge. If there is no filter capacitance in the magnet supply, it will decay almost
instantaneously while the beam filtering allows the high voltage to decay com-
paratively slowly. This permits defocusing of the electron beam and dissipation
of the beam power in the drift tube region. Melting of the drift tube will result.

·

Loss of one phase of the primary power supply. In some transmitters, the magnet
supply and the protection circuits are fed from a common leg of the 3-phase in-
coming power line. A loss of this phase will cause a loss of focusing and disable
the protection circuits, including body-current protection. If the low-volt-
age-sensing relay in the main breaker is misadjusted or nonoperational, the

© 1999 CRC Press LLC

high-voltage supply will remain on (although somewhat reduced) and, without
focusing or protection, may melt the tube body.

To prevent these types of failures, the following steps are recommended:

·

Conduct regular tests of the main circuit breaker in accordance with the manufac-
turer's instructions. Note that turning the breaker on and off using the transmitter
control circuitry checks only half of the system. Tests also should be conducted
by interrupting the primary feed to the breaker.

·

Compute or measure the discharge time constant of the filtering circuits of both
the beam and magnet supplies. Capacitance should be added to the magnet supply
if the time constant is less than that of the beam supply. Be careful, however, not
to increase the magnet supply filtering too much because it may also increase the
ramp-up time of the magnet voltage. This could result in the beam supply being
applied before the magnet supply is at its full potential. A simple diode-steering
arrangement, such as the one shown in

Figure 9.22

, will prevent such problems in

most cases.

·

Consider installing a multipole vacuum relay at the high-voltage input power
lines (between the main breaker and the high-power supply) to prevent instanta-
neous application of high voltage in the event of a malfunction of the main
breaker. The primary or low-voltage circuit configuration of the vacuum relays
should be such that it will prevent reenergizing of the relays before the magnetic
field has built up when primary power is restored after a line loss.

·

Install a solid-state phase-loss sensor if this protection is not already provided.

·

Install an electromagnet undercurrent protection circuit that will remove the high
voltage if the magnet current falls below a preset level.

Figure 9.22 Diode-steering filtered power supply for a klystron focusing electromag-
net.

© 1999 CRC Press LLC

An end user who is contemplating any of these changes should consult the original
equipment manufacturer before proceeding.

9.6

References

1. Capitano, J., and J. Feinstein, "Environmental Stress Screening Demonstrates Its

Value in the Field," Proceedings of the IEEE Reliability and Maintainability Sym-
posium, IEEE, New York, 1986.

2. Wong, Kam L., "Demonstrating Reliability and Reliability Growth with Environ-

mental Stress Screening Data," Proceedings of the IEEE Reliability and Maintain-
ability Symposium, IEEE, New York, 1990.

3. Tustin, Wayne, "Recipe for Reliability: Shake and Bake," IEEE Spectrum, IEEE,

New York, December 1986.

4. Hobbs, Gregg K., "Development of Stress Screens," Proceedings of the IEEE Reli-

ability and Maintainability Symposium, IEEE, New York, 1987.

5. Smith, William B., "Integrated Product and Process Design to Achieve High Reli-

ability in Both Early and Useful Life of the Product," Proceedings of the IEEE Reli-
ability and Maintainability Symposium, IEEE, New York, 1987.

6. "Conditioning of Large Radio-Frequency Power Tubes," Application Bulletin no.

21, Varian/Eimac, San Carlos, CA, 1984.

7. Johnson, Walter, "Techniques to Extend the Service Life of High-Power Vacuum

Tubes," NAB Engineering Conference Proceedings, National Association of
Broadcasters, Washington, DC, pp. 285293, 1994.

8. "Fault Protection," Application Bulletin no. 17, Varian/Eimac, San Carlos, CA,

1987.

9. "Cleaning and Flushing Klystron Water and Vapor Cooling Systems," Application

Engineering Bulletin no. AEB-32, Varian Associates, Palo Alto, CA, 1982.

10. "Reconditioning of UHF-TV Klystron Electron Gun Elements," Technical Bulle-

tin no. 4867, Varian Associates, Palo Alto, CA, July 1985.

11. "UHF-TV Klystron Focusing Electromagnets," Technical Bulletin no. 3122,

Varian Associates, Palo Alto, CA, November 1973.

12. "Recommendations for Protection of 4KM Series Klystrons During Prime Power

Interruptions," Technical Bulletin no. 3542, Varian Associates, Palo Alto, CA.

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1986.

Clark, Richard J., "Electronic Packaging and Interconnect Technology Working

Group Report (IDA/OSD R&M Study)," IDA Record Document D-39, Institute of
Defense Analysis, August 1983.

© 1999 CRC Press LLC

"Cleaning UHF-TV Klystron Ceramics," Technical Bulletin no. 4537, Varian Associ-

ates, Palo Alto, CA, May 1982.

DesPlas, Edward, "Reliability in the Manufacturing Cycle," Proceedings of the IEEE

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© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

10 Device Performance Criteria"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

10

Device Performance Criteria

10.1 Introduction

Many shortcomings in the received audio, video, and/or data stream of a transmission
resulting from noise, reflections, and terrain limitations cannot be controlled by the
equipment design engineer or the operating maintenance engineer. However, other
problems can be prevented through proper application of vacuum tube devices and
proper equipment adjustment. The critical performance parameters for a transmission
system include:

·

Output power

·

Transmission line/antenna VSWR

·

Frequency stability/accuracy

·

Occupied bandwidth

·

Distortion mechanisms, including total harmonic distortion (THD), intercarrier
phase distortion, and quadrature distortion

·

Differential phase and gain

·

Frequency response errors

·

Envelope delay

The adjustment schedule required to maintain a satisfactory level of performance

depends upon a number of factors, including:

·

Operating environment

·

Equipment quality

·

Age of the equipment

·

Type of service being performed

© 1999 CRC Press LLC

10.2 Measurement Parameters

Most measurements in the RF field involve characterizing fundamental parameters.
These include signal level, phase, and frequency. Most other tests consist of measur-
ing these fundamental parameters and displaying the results in combination by using
some convenient format. For example, signal-to-noise ratio (S/N) consists of a pair of
level measurements made under different conditions and expressed as a logarithmic
ratio. Modern test instruments perform the necessary manipulation of the measured
parameters to yield a direct readout of the quantity of interest.

When characterizing a vacuum tube device or stage, it is common to view it as a box

with input and output terminals. In normal use, a signal is applied to the input of the
box, and the signal--modified in some way--appears at the output. Some measure-
ments are one-port tests, such as impedance or noise level, and are not concerned with
input/output signals, but with only one or the other.

Measurements are made on equipment to check performance under specified condi-

tions and to assess suitability for use in a particular application. The measurements may
be used to verify specified system performance or as a way of comparing several pieces
of equipment for use in a system. Measurements also may be used to identify compo-
nents in need of adjustment or repair.

Many parameters are important in vacuum tube devices and merit attention in the

measurement process. Some common measurements include frequency response, gain
or loss, harmonic distortion, intermodulation distortion, noise level, phase response,
and transient response.

Measurement of level is fundamental to most RF specifications. Level can be mea-

sured either in absolute terms or in relative terms. Power output is an example of an ab-
solute level measurement; it does not require any reference. S/N and gain or loss are ex-
amples of relative, or ratio, measurements; the result is expressed as a ratio of two mea-
surements. Although it may not appear so at first, frequency response is also a relative
measurement. It expresses the gain of the device under test as a function of frequency,
with the midband gain (typically) as a reference.

Distortion measurements are a way of quantifying the unwanted components added

to a signal by a piece of equipment. The most common technique is total harmonic dis-
tortion, but others are often used. Distortion measurements express the unwanted sig-
nal components relative to the desired signal, usually as a percentage or decibel value.
This is another example of multiple level measurements that are combined to give a
new measurement figure.

10.2.1 Power Measurements

The simplest definition of a level measurement is the alternating current amplitude at
a particular place in the system under test. However, in contrast to direct current mea-
surements, there are many ways of specifying ac voltage in a circuit. The most com-
mon methods include:

·

Average response

© 1999 CRC Press LLC

·

Root mean square (rms)

·

Peak

Strictly speaking, the term level refers to a logarithmic, or decibel, measurement.
However, common parlance employs the term for an ac amplitude measurement, and
that convention will be followed in this chapter.

Root Mean Square

The root-mean-square technique measures the effective power of the ac signal. It
specifies the value of the dc equivalent that would dissipate the same power if either
were applied to a load resistor. This process is illustrated in

Figure 10.1

for voltage

measurements. The input signal is squared, and the average value is found. This is
equivalent to finding the average power. The square root of this value is taken to trans-
fer the signal from a power value back to a voltage. For the case of a sine wave, the
rms value is 0.707 of its maximum value.

Assume that the signal is no longer a sine wave but rather a sine wave and several

harmonics. If the rms amplitude of each harmonic is measured individually and added,

Figure 10.1 Root-mean-square (rms) voltage measurements: (

a) the relationship of

rms and average values, (

b) the rms measurement circuit.

(

b)

(

a)

© 1999 CRC Press LLC

the resulting value will be the same as an rms measurement of the signals together. Be-
cause rms voltages cannot be added directly, it is necessary to perform an rms addition.
Each voltage is squared, and the squared values are added as follows:

V

V

V

V

rms total

rms

rms

rms n

=

+

+

1

2

2

2

2

...

(10.1)

Note that the result is not dependent on the phase relationship of the signal and its har-
monics. The rms value is determined completely by the amplitude of the components.
This mathematical predictability is useful in practical applications of level measure-
ment, enabling correlation of measurements made at different places in a system. It is
also important in correlating measurements with theoretical calculations.

Average-Response Measurement

The average-responding voltmeter measures ac voltage by rectifying it and filtering
the resulting waveform to its average value, as shown in

Figure 10.2

. This results in a

dc voltage that can be read on a standard dc voltmeter. As shown in the figure, the av-
erage value of a sine wave is 0.637 of its maximum amplitude. Average-responding

Figure 10.2 Average voltage measurements: (

a) illustration of average detection, (b)

average measurement circuit.

(

b)

(

a)

© 1999 CRC Press LLC

meters usually are calibrated to read the same as an rms meter for the case of a single
sine wave signal. This results in the measurement being scaled by a constant K of
0.707/0.637, or 1.11. Meters of this type are called average-responding, rms cali-
brated. For signals other than sine waves, the response will be different and difficult
to predict.

If multiple sine waves are applied, the reading will depend on the phase shift be-

tween the components and will no longer match the rms measurement. A comparison
of rms and average-response measurements is made in

Figure 10.3

for various wave-

forms. If the average readings are adjusted as described previously to make the average
and rms values equal for a sine wave, all the numbers in the average column would be
increased by 11.1 percent, and the rms-average numbers would be reduced by 11.1 per-
cent.

Peak-Response Measurement

Peak-responding meters measure the maximum value that the ac signal reaches as a
function of time. This approach is illustrated in

Figure 10.4

. The signal is

full-wave-rectified to find its absolute value, then passed through a diode to a storage
capacitor. When the absolute value of the voltage rises above the value stored on the
capacitor, the diode will conduct and increase the stored voltage. When the voltage
decreases, the capacitor will maintain the old value. Some method of discharging the
capacitor is required so that a new peak value can be measured. In a true peak detec-
tor, this is accomplished by a solid-state switch. Practical peak detectors usually in-

Figure 10.3 Comparison of rms and average voltage characteristics.

© 1999 CRC Press LLC

clude a large-value resistor to discharge the capacitor gradually after the user has had
a chance to read the meter.

The ratio of the true peak to the rms value is called the crest factor. For any signal but

an ideal square wave, the crest factor will be greater than 1, as demonstrated in

Figure

10.5

. As the measured signal becomes more peaked, the crest factor increases.

By introducing a controlled charge and discharge time, a quasi-peak detector is

achieved. The charge and discharge times may be selected, for example, to simulate the
transmission pattern for a digital carrier. The gain of a quasi-peak detector is normally
calibrated so that it reads the same as an rms detector for sine waves.

The peak-equivalent sine is another method of specifying signal amplitude. This

value is the rms level of a sine wave having the same peak-to-peak amplitude as the sig-
nal under consideration. This is the peak value of the waveform scaled by the correction
factor 1.414, corresponding to the peak-to-rms ratio of a sine wave. Peak-equivalent
sine is useful when specifying test levels of waveforms in distortion measurements. If
the distortion of a device is measured as a function of amplitude, a point will be reached
at which the output level cannot increase any further. At this point the peaks of the
waveform will be clipped, and the distortion will rise rapidly with further increases in
level. If another signal is used for distortion testing on the same device, it is desirable

Figure 10.4 Peak voltage measurements: (

a) illustration of peak detection, (b) peak

measurement circuit.

(

b)

(

a)

© 1999 CRC Press LLC

that the levels at which clipping is reached correspond. Signal generators are normally
calibrated in this way to allow changing between waveforms without clipping or read-
justing levels.

Measurement Bandwidth

The bandwidth of the level-measuring instrument can have a significant effect on the
accuracy of the reading. For a meter with a single-pole rolloff (one bandwidth-limit-
ing component in the signal path), significant measurement errors can occur. Such a
meter with a specified bandwidth of 1 MHz, for example, will register a 10 percent
error in the measurement of signals at 500 kHz. To obtain 1 percent accurate measure-
ments (disregarding other error sources in the meter), the signal frequency must be
less than 100 kHz.

Figure 10.6

illustrates another problem associated with limited-bandwidth measur-

ing devices. In the figure, a distorted sine wave is measured by two meters with differ-
ent bandwidths. The meter with the narrower bandwidth does not respond to all the har-
monics and gives a lower reading. The severity of this effect varies with the frequency
being measured and the bandwidth of the meter; it can be especially severe in the mea-
surement of wideband noise. Peak measurements are particularly sensitive to band-
width effects. Systems with restricted low-frequency bandwidth will produce tilt in a
square wave, and bumps in the high-frequency response will produce an overshoot. The
effect of either will be an increase in the peak reading.

Meter Accuracy

Accuracy is a measure of how well an instrument quantifies a signal at a midband fre-
quency. This sets a basic limit on the performance of the meter in establishing the ab-
solute amplitude of a signal. It is also important to look at the flatness specification to

Figure 10.5 Illustration of the crest factor in voltage measurements.

© 1999 CRC Press LLC

see how well this performance is maintained with changes in frequency. Flatness de-
scribes how well the measurements at any other frequency track those at the refer-
ence. If a meter has an accuracy of 2 percent at 1 MHz and a flatness of 1 dB (10 per-
cent) from 20 kHz to 20 MHz, the inaccuracy can be as great as 12 percent at 20 MHz.

Meters often have a specification of accuracy that changes with voltage range, being

most accurate only in the range in which the instrument was calibrated. A meter with 1
percent accuracy on the 2 V range and 1 percent accuracy per step would be 3 percent
accurate on the 200 V scale. Using the flatness specification given previously, the over-
all accuracy for a 100 V, 20 MHz sine wave is 14 percent. In many instruments, an addi-
tional accuracy derating is given for readings as a percentage of full scale, making read-
ings at less than full scale less accurate.

However, the accuracy specification is not normally as important as the flatness.

When performing frequency response or gain measurements, the results are relative
and are not affected by the absolute voltage used. When measuring gain, however, the
attenuator accuracy of the instrument is a direct error source. Similar comments apply
to the accuracy and flatness specifications for signal generators. Most are specified in
the same manner as voltmeters, with the inaccuracies adding in much the same manner.

RF Power Measurement

Measurement of RF power output typically is performed with a directional wattmeter,
calibrated for use over a specified range of frequencies and power levels. Various
grades of accuracy are available, depending upon the requirements of the application.
The most accurate measurements usually are made by determining the temperature
rise of the cooling through a calibrated dummy load. The power absorbed by the cool-
ant, usually water, can be calculated from the following equation:

P

K

Q

T

= × ×

(10.2)

Where:

Figure 10.6 The effects of instrument bandwidth on voltage measurements.

© 1999 CRC Press LLC

P = power dissipated in kilowatts
K = a constant, determined by the coolant (for pure water at 30°C, K = 0.264)
Q = coolant flow in gallons per minute
T = difference between inlet and outlet water temperature in degrees Celsius

This procedure often is used to verify the accuracy of in-line directional wattmeters,
which are more convenient to use but typically offer less accuracy.

10.2.2 Decibel Measurement

Measurements in RF work often are expressed in decibels. Radio frequency signals
span a wide range of levels, too wide to be accommodated on a linear scale. The deci-
bel is a logarithmic unit that compresses this wide range down to one that is easier to
handle. Order-of-magnitude (factor of 10) changes result in equal increments on a
decibel scale. A decibel may be defined as the logarithmic ratio of two power mea-
surements or as the logarithmic ratio of two voltages:

db = 20

1

2

log

E

E

(10.3)

db = 10

1

2

log

P

P

(10.4)

Decibel values from power measurements and decibel values from voltage measure-

ments are equal if the impedances are equal. In both equations, the denominator vari-
able is usually a stated reference, as illustrated by the example in

Figure 10.7

. Whether

the decibel value is computed from the power-based equation or from the voltage-based
equation, the same result is obtained.

Figure 10.7 Example of the equiv-
alence of voltage and power deci-
bels.

© 1999 CRC Press LLC

A doubling of voltage will yield a value of 6.02 dB, and a doubling of power will

yield 3.01 dB. This is true because doubling of voltage results in an increase in power by
a factor of 4.

Table 10.1

lists the decibel values for some common voltage and power ra-

tios.

RF engineers often express the decibel value of a signal relative to some standard

reference instead of another signal. The reference for decibel measurements may be
predefined as a power level, as in dBk (decibels above 1 kW), or it may be a voltage ref-
erence. Often, it is desirable to specify levels in terms of a reference transmission level
somewhere in the system under test. These measurements are designated dBr, where
the reference point or level must be separately conveyed.

10.2.3 Noise Measurement

Noise measurements are specialized level measurements. Noise may be expressed as
an absolute level by simply measuring the voltage at the desired point in the system.
This approach, however, is often not very meaningful. Specifying the noise perfor-
mance as the signal-to-noise ratio is a better approach. S/N is a decibel measurement
of the noise level using the signal level measured at the same point as a reference.
This makes measurements at different points in a system or in different systems di-
rectly comparable. A signal with a given S/N can be amplified with a perfect ampli-
fier or attenuated with no change in the S/N. Any degradation in S/N at a later point in
the system is the result of limitations of the equipment that follows.

dB Value

Voltage Ratio

Power Ratio

0

1

1

+1

1.122

1.259

+2

1.259

1.586

+3

1.412

1.995

+6

1.995

3.981

+10

3.162

10

+20

10

100

+40

100

10,000

1

0.891

0.794

2

0.794

0.631

3

0.707

0.501

6

0.501

0.251

10

0.3163

0.1

20

0.1

0.01

40

0.01

0.0001

Table 10.1 Common Decibel Values and Conversion Ratios

© 1999 CRC Press LLC

Noise performance is an important parameter in the operation of any power vacuum

tube amplifier or oscillator. All electric conductors contain free electrons that are in
continuous random motion. It can be expected that, by pure chance, more electrons will
be moving in one direction than in another at any instant. The result is that a voltage will
be developed across the terminals of the conductor if it is an open circuit, or a current
will be delivered to any connected circuit. Because this voltage (or current) varies in a
random manner, it represents noise energy distributed throughout the frequency spec-
trum, from the lowest frequencies well into the microwave range. This effect is com-
monly referred to as thermal-agitation noise, because the motion of electrons results
from thermal action. It is also referred to as resistance noise. The magnitude of the
noise depends upon the following:

·

The resistance across which the noise is developed

·

Absolute temperature of the resistance

·

Bandwidth of the system involved

Random noise, similar in character to that produced in a resistance, is generated in

vacuum tubes as a result of irregularities in electron flow. Tube noise can be divided
into the following general classes:

·

Shot effect, representing random variations in the rate of electron emission from
the cathode

·

Partition noise, arising from chance variations in the division of current between
two or more positive electrodes

·

Induced grid noise, produced as a result of variations in the electron stream pass-
ing adjacent to a grid

·

Gas noise, generated by random variations in the rate of ion production by colli-
sion

·

Secondary emission noise, arising from random variations in the rate of produc-
tion of secondary electrons

·

Flicker effect, a low-frequency variation in emission that occurs with ox-
ide-coated cathodes

Shot effect in the presence of space charge, partition noise, and induced grid noise is
the principal source of tube noise that must be considered in RF work.

10.2.4 Phase Measurement

When a signal is applied to the input of a device, the output will appear at some later
point in time. For sine wave excitation, this delay between input and output may be ex-
pressed as a proportion of the sine wave cycle, usually in degrees. One cycle is 360°,
one half-cycle is 180°, and so on. This measurement is illustrated in

Figure 10.8

. The

© 1999 CRC Press LLC

phasemeter input signal number 2 is delayed from, or is said to be lagging, input num-
ber 1 by 45°.

Most RF test equipment checks phase directly by measuring the proportion of one

signal cycle between zero crossings of the signals. Phase typically is measured and re-
corded as a function of frequency over a specified range. For most vacuum tube de-
vices, phase and amplitude responses are closely coupled. Any change in amplitude
that varies with frequency will produce a corresponding phase shift.

Relation to Frequency

When dealing with complex signals, the meaning of phase can become unclear.
Viewing the signal as the sum of its components according to Fourier theory, a differ-
ent value of phase shift is found at each frequency. With a different phase value on
each component, the one to be used as the reference is unclear. If the signal is periodic
and the waveshape is unchanged passing through the device under test, a phase value
still may be defined. This may be done by using the shift of the zero crossings as a
fraction of the waveform period. Indeed, most commercial phase-measuring instru-
ments will display this value. However, in the case of differential phase shift with fre-
quency, the waveshape will be changed. It is then impossible to define any phase-shift
value, and phase must be expressed as a function of frequency.

Group delay is another useful expression of the phase characteristics of an RF de-

vice. Group delay is the slope of the phase response. It expresses the relative delay of
the spectral components of a complex waveform. If the group delay is flat, all compo-
nents will arrive at a given point together. A peak or rise in the group delay indicates that
those components will arrive later by the amount of the peak or rise. Group delay is
computed by taking the derivative of the phase response vs. frequency. Mathematically:

Figure 10.8 Illustration of the measurement of a phase shift between two signals.

© 1999 CRC Press LLC

(

)

Group delay

phase at f

phase at f

f

f

= -

-
-

2

1

2

1

(10.5)

This definition requires that phase be measured over a range of frequencies to give a
curve that can be differentiated. It also requires that the phase measurements be per-
formed at frequencies sufficiently close to provide a smooth and accurate derivative.

10.2.5 Nonlinear Distortion

Distortion is a measure of signal impurity, a deviation from ideal performance of a de-
vice, stage, or system. Distortion usually is expressed as a percentage or decibel ratio
of the undesired components to the desired components of a signal. Distortion of a
vacuum tube device or stage is measured by inputting one or more sine waves of vari-
ous amplitudes and frequencies. In simplistic terms, any frequencies at the output that
were not present at the input are distortion. However, strictly speaking, components
caused by power line interference or another spurious signal are not distortion but,
rather, noise. Many methods of measuring distortion are in common use, including
harmonic distortion and several types of intermodulation distortion.

Harmonic Distortion

The transfer characteristic of a typical vacuum tube amplifier is shown in

Figure 10.9

.

The transfer characteristic represents the output voltage at any point in the signal
waveform for a given input voltage; ideally this is a straight line. The output wave-
form is the projection of the input sine wave on the device transfer characteristic. A

Figure 10.9 Illustration of total harmonic distortion (THD) measurement of an amplifier
transfer characteristic.

© 1999 CRC Press LLC

change in the input produces a proportional change in the output. Because the actual
transfer characteristic is nonlinear, a distorted version of the input waveshape appears
at the output.

Harmonic distortion measurements excite the device under test with a sine wave and

measure the spectrum of the output. Because of the nonlinearity of the transfer charac-
teristic, the output is not sinusoidal. By using Fourier series, it can be shown that the
output waveform consists of the original input sine wave plus sine waves at integer mul-
tiples (harmonics) of the input frequency. The spectrum of a distorted signal is shown in

Figure 10.10

. The harmonic amplitudes are proportional to the amount of distortion in

the device under test. The percentage of harmonic distortion is the rms sum of the har-
monic amplitudes divided by the rms amplitude of the fundamental.

Harmonic distortion may be measured with a spectrum analyzer or a distortion test

set.

Figure 10.11

shows the setup for a spectrum analyzer. As shown in

Figure 10.10

,

the fundamental amplitude is adjusted to the 0 dB mark on the display. The amplitudes
of the harmonics are then read and converted to linear scale. The rms sum of these val-
ues is taken, which represents the THD.

A simpler approach to the measurement of harmonic distortion can be found in the

notch-filter distortion analyzer, illustrated in

Figure 10.12

. This device, commonly re-

ferred to as simply a distortion analyzer, removes the fundamental of the signal to be in-
vestigated and measures the remainder.

Figure 10.13

shows the notch-filter approach

applied to a spectrum analyzer for distortion measurements.

The correct method of representing percentage distortion is to express the level of

the harmonics as a fraction of the fundamental level. However, many commercial dis-
tortion analyzers use the total signal level as the reference voltage. For small amounts of
distortion, these two quantities are essentially equivalent. At large values of distortion,

Figure 10.10 Example of reading THD from a spectrum analyzer.

© 1999 CRC Press LLC

however, the total signal level will be greater than the fundamental level. This makes
distortion measurements on such units lower than the actual value. The relationship be-
tween the measured distortion and true distortion is given in

Figure 10.14

. The errors

are not significant until about 20 percent measured distortion.

Because of the notch-filter response, any signal other than the fundamental will in-

fluence the results, not just harmonics. Some of these interfering signals are illustrated
in

Figure 10.15

. Any practical signal contains some noise, and the distortion analyzer

will include this noise in the reading. Because of these added components, the correct
term for this measurement is total harmonic distortion and noise (THD+N). Additional
filters are included on most distortion analyzers to reduce unwanted noise and permit a
more accurate reading.

The use of a sine wave test signal and a notch-type distortion analyzer provides the

distinct advantage of simplicity in both design and use. This simplicity has an addi-

Figure 10.11 Common test setup to measure harmonic distortion with a spectrum ana-
lyzer.

Figure 10.12 Simplified block diagram of a harmonic distortion analyzer.

© 1999 CRC Press LLC

tional benefit in ease of interpretation. The shape of the output waveform from a
notch-type analyzer indicates the slope of the nonlinearity. Displaying the residual
components on the vertical axis of an oscilloscope and the input signal on the horizon-
tal axis provides a plot of the deviation of the transfer characteristic from a best-fit
straight line. This technique is diagrammed in

Figure 10.16

. The trace will be a hori-

zontal line for a perfectly linear device. If the transfer characteristic curves upward on
positive input voltages, the trace will bend upward at the right-hand side.

Figure 10.13 Typical test setup for measuring the harmonic and spurious output of a
transmitter. The notch filter is used to remove the fundamental frequency to prevent
overdriving the spectrum analyzer input and to aid in evaluation of distortion compo-
nents.

Figure 10.14 Conversion graph for indicated distortion and true distortion.

© 1999 CRC Press LLC

Examination of the distortion components in real time on an oscilloscope allow ob-

servation of oscillation on the peaks of the signal and clipping. This is a valuable tool in
the design and development of RF circuits, and one that no other distortion measure-
ment method can fully match. Viewing the residual components in the frequency do-

Figure 10.15 Example of interference sources in distortion and noise measurements.

Figure 10.16 Transfer-function monitoring configuration using an oscilloscope and dis-
tortion analyzer.

© 1999 CRC Press LLC

main by using a spectrum analyzer also reveals much information about the distortion
mechanism inside the device or stage under test.

When measuring distortion at high frequencies, bandwidth limitations are an im-

portant consideration, as illustrated in

Figure 10.17

. Because the components being

measured are harmonics of the input frequency, they may fall outside the passband of
the device under test.

Intermodulation Distortion

Intermodulation distortion (IMD) describes the presence of unwanted signals that are
caused by interactions between two or more desired signals. IMD, as it applies to an
RF amplifier, refers to the generation of spurious products in a nonlinear amplifying
stage. IMD can be measured using a distortion monitor or a spectrum analyzer.

Nonlinearities in a circuit can cause it to act like a mixer, generating the sum and dif-

ference frequencies of two signals that are present. These same imperfections generate
harmonics of the signals, which then can be mixed with other fundamental or harmonic
frequencies.

Figure 10.18

shows the relationship of these signals in the frequency do-

main. The order of a particular product of IMD is defined as the number of "steps" be-
tween it and a single fundamental signal. Harmonics add steps. For example, the second
harmonic = 2f

1

= f

1

+ f

1

, which is two steps, making it a second-order product. The prod-

uct resulting from 2f

1

f

2

is, then, a third-order product. Some of the IMD products re-

sulting from two fundamental frequencies include:

Figure 10.17 Illustration of problems that occur when measuring harmonic distortion in
band-limited systems.

© 1999 CRC Press LLC

second-order = 2f

1

(10.6)

= 2f

2

(10.7)

= f

1

+ f

2

(10.8)

= f

1

f

2

(10.9)

third-order = 2f

1

+ f

2

(10.10)

= 2f

1

f

2

(10.11)

= 2f

2

+ f

1

(10.12)

= 2f

2

f

1

(10.13)

and so on.

The order is important because, in general, the amplitudes of IMD products fall off

as the order increases. Therefore, low-order IMD products have the greatest potential to
cause problems if they fall within the band of interest, or on frequencies used by other
services or equipment. Odd-order IMD products are particularly troublesome because
they fall closest to the signals that cause them, usually within the operating frequency
band.

Measurement Techniques

A number of techniques have been devised to measure the intermodulation (IM) of
two or more signals passing through a device simultaneously. The most common of

Figure 10.18 Frequency-domain relationships of the desired signals and low-order
IMD products.

© 1999 CRC Press LLC

these involves the application of a test waveform consisting of a low-frequency signal
and a high-frequency signal mixed in a specified amplitude ratio. The signal is ap-
plied to the device under test, and the output waveform is examined for modulation of
the upper frequency by the low-frequency signal. The amount by which the low-fre-
quency signal modulates the high-frequency signal indicates the degree of
nonlinearity. As with harmonic distortion measurement, this test may be done with a
spectrum analyzer or a dedicated distortion analyzer. The test setup for a spectrum an-
alyzer is shown in

Figure 10.19

. Two independent signal sources are connected using

a power combiner to drive the device under test. The sources are set at the same output
level, but at different frequencies. A typical spectrum analyzer display of the two-tone
distortion test is shown in

Figure 10.20

. As shown, the third-order products that fall

close to the original two tones are measured. This measurement is a common, and im-
portant, one because such products are difficult to remove by filtering.

The mechanisms of intermodulation as applied to interfering radiated signals are

discussed later in this chapter.

Addition and Cancellation of Distortion Components

The addition and cancellation of distortion components in test equipment or the de-
vice under test is an often-overlooked problem in measurement. Consider the exam-
ples in

Figure 10.21

and

10.22

. Assume that one device under test has a transfer char-

acteristic similar to that diagrammed at the top left of

Figure 10.21a

, and another has

the characteristic diagrammed at the center. If the devices are cascaded, the resulting
transfer-characteristic nonlinearity will be magnified as shown. The effect on sine
waves in the time domain is illustrated in

Figure 10.21b

. The distortion components

generated by each nonlinearity are in phase and will sum to a component of twice the
original magnitude. However, if the second device under test has a complementary
transfer characteristic, as shown in

Figure 10.22

, quite a different result is obtained.

When the devices are cascaded, the effects of the two curves cancel, yielding a
straight line for the transfer characteristic (

Figure 10.22a

). The corresponding distor-

Figure 10.19 Test setup for measuring the intermodulation distortion of an RF amplifier
using a spectrum analyzer.

© 1999 CRC Press LLC

Figure 10.20 Typical two-tone IMD measurement, which evaluates the third-order
products within the bandpass of the original tones.

Figure 10.21 Illustration of the addition of distortion components: (

a) addition of trans-

fer-function nonlinearities, (

b) addition of distortion components.

(

b)

(

a)

© 1999 CRC Press LLC

tion products are out of phase, resulting in no measured distortion components in the
final output (

Figure 10.22b

).

This problem is common at low levels of distortion, especially between the test

equipment and the device under test. For example, if the test equipment has a residual of
0.002 percent when connected to itself, and readings of 0.001 percent are obtained from
the circuit under test, cancellations are occurring. It is also possible for cancellations to
occur in the test equipment itself, with the combined analyzer and signal generator giv-
ing readings lower than the sum of their individual residuals. If the distortion is the re-
sult of even-order (asymmetrical) nonlinearity, reversing the phase of the signal be-
tween the offending devices will change a cancellation to an addition. If the distortion is
the result of an odd-order (symmetrical) nonlinearity, phase inversions will not affect
the cancellation.

Intermodulation Precorrection Techniques

In a practical system, amplifiers cannot be absolutely linear; all linear amplifiers ex-

hibit intermodulation distortion to some extent, relative to the amplitudes of the par-
ent signals [1]. Adequate linearity is judged by the application, but in general, the
in-band odd-order products should be as low as the state of the art permits because
they are in-band and thus cannot be filtered out.

All intermodulation products are the result of undesirable time domain multiplica-

tion of the matrix of input signals to a linear amplifier. All harmonic components are
the result of the same process, only here, an individual signal is multiplied by itself. For

Figure 10.22 Illustration of the cancellation of distortion components: (

a) cancellation of

transfer-characteristic nonlinearities, (

b) cancellation of the distortion waveform.

(

b)

(

a)

© 1999 CRC Press LLC

second order intermodulation, the ideal linear amplifier is modified by modeling a sin-
gle mixer feeding a time domain multiplied signal around the amplifier. For third order
intermodulation, the ideal linear amplifier is modified by modeling two mixers feeding
a time domain multiplied signal around the amplifier, as shown in

Figure 10.23

.

Because conventional amplitude modulation contains at least three parent signals

(the carrier plus two sidebands), AM amplification has always been fraught with
in-band intermodulation distortion. A common technique to correct for these distortion
products is to precorrect the power amplifier using the following process:

·

Intentionally generate an anti-intermodulation signal by mixing the three input
signals together

·

Adjust the amplitude of the anti-IM product to match that of the real IM product

·

Invert the phase

·

Delay the correction signal or the parent signals in time so as to permit proper
alignment

·

Add the correction signal into the main parent signal path to cause cancellation in
the output circuit of the amplifier

This is a well-known technique used in countless applications.

Each intermodulation product may have its own anti-intermodulation product inten-

tionally generated to cause targeted cancellation. Even if one or more parent signals
contains modulation of some type--AM, FM, or PM--the legitimate modulation sig-
nal is often included in the intended mixing circuit so that even the sidebands produced
around an intermodulation distortion product are canceled.

The precorrection circuit typically consists of mixer devices with a local oscillator

(LO) port, an RF port, and an IF port. The LO port is fed with the parent signal, such as
the carrier, which contains no modulation. Its purpose is to hard-switch the mixer di-

Figure 10.23 Basic model of an intermodulation correction circuit. (

After [1].)

© 1999 CRC Press LLC

odes, causing frequency translation of the signal fed to the RF port to the sum and dif-
ference frequencies of the LO and RF input signals exiting at the IF port. The
intermodulation product to be canceled is either a result of the sum or the difference sig-
nal so that the product of no concern is filtered out. The resulting intended mixer prod-
uct is fed to the IF port of a second mixer where LO injection is from the third parent
signal. The third order intended mixing product is then available at the second mixer RF
port, and will be at the frequency of the offending IM and coherent with it.

Digital signals suitable for radio transmission actually remain analog, but are

termed digital because of the nature of the information that they carry. A modern digital
radio transmission using quadrature amplitude modulation (QAM) or n-level vestigial
sideband modulation (e.g., 8-VSB) consists of a vector signal constantly moving in
phase angle and magnitude relative to some reference. The vector signal is smooth and
without discontinuity so that it is well contained within a finite bandwidth. Its digital
nature stems from decoding the proper point or spatial area around a point (QAM) of a

Figure 10.24 Digital decoding process: (

a) 16-QAM, (b) 8-VSB. (After [1].)

(

b)

(

a)

© 1999 CRC Press LLC

finite number of legitimate points, or the proper line (VSB) or spatial area around a line
of a finite number of legitimate lines, that the vector is passing though.

Figure 10.24

il-

lustrates this process. In either case, the area is assigned a discrete digital value, which
represents a portion of the information being transmitted. Spatial location is generally
expressed in terms of rectangular coordinates with I (in phase) and Q (quadrature
phase) components.

Because the vector appears to move at random from point to point, especially result-

ing from the use of an intentional randomizer circuit, the frequency distribution within
the passband of the signal is constant, flat, and favors no particular frequency.

Figure

10.25

illustrates the frequency distribution with the sidebands shown before and after

amplification. As a result, the transmitted signal is said to be "noise-like." As such, it
takes on the nature of noise and need not be periodic.

For this case, there is no identifiable set of parent components upon which to obtain

signals for the purpose of mixing to generate an anti-intermodulation signal for cancel-
lation purposes. There is no carrier signal to feed into the LO port of a mixer for hard
switching to generate sum and difference frequencies. Because the signal is noise-like,
there is no correlation of any piece of passband spectrum with any other. Generating a
precorrection signal, thus, is not possible using the process described for the discrete
signal analog case.

10.3 Vacuum Tube Operating Parameters

Direct current input power to the final RF output stage is a function of the applied
plate or collector voltage and the plate or collector current of the final tube(s) [2]. Test
points or built-in meters usually are provided for checking the dc voltage and current
levels of the final amplifier.

Transmitter deviation or modulation must be set properly. A deviation level set too

high or too low adversely affects radio range. If the deviation is set too low, the sig-

Figure 10.25 Digitally modulated RF signal spectrum before and after amplification.
(

After [1].)

© 1999 CRC Press LLC

nal-to-noise ratio of the overall system is diminished. If the deviation is set too high, the
transmitter signal may exceed the receiver's modulation acceptance bandwidth, caus-
ing severe distortion on modulation peaks and out-of-band spurious radiation.

The difference between the positive and negative excursions of the deviated wave-

form is defined as modulation dissymmetry. Theoretically, positive and negative excur-
sions should be perfectly symmetrical. In practice, however, there is usually some
dissymmetry. In a multichannel phase-modulated transmitter, modulation
dissymmetry may be worse on some channels than others as a result of modulator tun-
ing. If the modulator tuning is peaked on one channel, for example, response on another
channel may be compromised.

10.3.1 Stage Tuning

A power output stage can be tuned in any number of ways [2]. Determining factors in-
clude the type of device, circuit design, and operational objectives. A PA can be tuned
to optimize bandwidth, linearity, efficiency, and other parameters. It is often neces-
sary to compromise one parameter to enhance another. This tradeoff is dictated by the
application.

Power Grid Tubes

When optimizing a tetrode or pentode RF amplifier for proper excitation and loading,
the general tuning procedure varies depending upon whether the screen voltage is
taken from a fixed supply or a dropping resistor supply with poor regulation [2]. If
both the screen supply and grid bias are from fixed sources with good regulation, the
plate current is almost entirely controlled by RF excitation. The loading is then varied
until the maximum power output is obtained. Following these adjustments, the excita-
tion is trimmed, along with the loading, until the desired control and screen grid cur-
rents are obtained.

In the case of an RF amplifier where both the screen and grid bias are taken from

sources with poor regulation, the stage will tune in a manner similar to a triode RF
power amplifier. The plate current will be controlled principally by varying the load-
ing, and the excitation trimmed to yield the desired control grid current. In this case, the
screen current will be almost entirely set by the choice of the dropping resistor. Excita-
tion and loading will vary the screen voltage considerably, and these should be trimmed
to yield a desired screen voltage.

The grounded-grid amplifier has been used for many years, and, with the advent of

high-power zero-bias triodes, this configuration has become more common. Adjust-
ment of the excitation and loading of a grounded-grid RF amplifier requires a slightly
different procedure. The plate voltage (plate and screen voltage in the case of a tetrode
or pentode) must be applied before the excitation. If this precaution is not followed, the
control grid may sustain damage. The loading is increased as the excitation is in-
creased. When the desired plate current is reached, note the power output. The loading
can be reduced slightly and the excitation increased until the plate current is the same as
before. If the power output is less than before, a check can be made with increased load-

© 1999 CRC Press LLC

ing and less excitation. By proper trimming, the proper grid current, plate current, and
optimum power output can be obtained. In a grounded-grid circuit, the cathode or input
circuit is in series with the plate circuit. Because of this arrangement, any change made
in the plate circuit will affect the input circuit. Therefore, the driver amplifier does not
see its designed load until the driven stage is up to full plate current.

Typical operating values provide a starting point for tuning efforts. As long as the

maximum limits are not exceeded, a wide range of tuning options is available.

Screen Voltage

Typical operating values given on the data sheet for tetrode and pentode devices in-
clude recommended screen voltages [2]. These values are not critical for most appli-
cations; they are chosen as a convenient value consistent with low driving power and
reasonable screen dissipation. If lower values of screen voltage are used, more driving
voltage will be required on the grid to obtain the same plate current. If higher values
of screen voltage are used, less driving voltage will be required. Thus, high power
gain can be achieved, provided the circuit exhibits adequate stability. Be careful not to
exceed the screen dissipation limit. The value of screen voltage can be chosen to suit
available power supplies or amplifier conditions.

The published characteristic curves of tetrodes and pentodes are shown for com-

monly used screen voltages. Occasionally, it is desirable to operate the tetrode or
pentode at some screen voltage other than that shown on the characteristic curves. It is a
relatively simple matter to convert the published curves to corresponding curves at a
different screen voltage. This conversion is based on the fact that if all interelectrode
voltages are either raised or lowered by the same relative amount, the shape of the volt-
age field pattern is not altered, and neither is the current distribution. The current lines
simply take on new proportionate values in accordance with the three-halves power
law. This method fails only where insufficient cathode emission or high secondary
emission affects the current values.

For example, if the characteristic curves are shown at a screen voltage of 250 V and it

is desirable to determine conditions at 500 V on the screen, all voltage scales should be
multiplied by the same factor that is applied to the screen voltage (2 in this case). The 1
kV plate voltage line thus becomes 2 kV, the 50 V grid voltage line becomes 100 V, and
so on. All current lines then assume new values in accordance with the 3/2 power law.
Because the voltage was increased by a factor of 2, the current lines all will be increased
in value by a factor of 2

3/2

or 2.8. All current values are then multiplied by the factor of

2.8. The 100 mA line thus becomes a 280 mA line, and so on.

Likewise, if the screen voltage given on the characteristic curve is higher than the

conditions desired, the voltage should be reduced by the same factor that is used to ob-
tain the desired screen voltage. Correspondingly, the current values all will be reduced
by an amount equal to the 3/2 power of this factor. Commonly used factors are given in

Table 10.2

.

© 1999 CRC Press LLC

Back-Heating by Electrons

Back-heating of the cathode is an undesirable action involving the motion of electrons
within the tube at VHF and UHF [2]. Because the time of flight of electrons from the
cathode through the grid structure to the plate becomes an appreciable part of the cy-
cle at high frequencies, electrons can be stopped in flight and turned back by the rap-
idly changing grid voltage. Under these conditions, the electrons are deflected from
their normal paths and given excess energy with which they bombard the cathode and
other portions of the tube structure. This effect can be aggravated by the choice of op-
erating conditions to the extent that destructive effects occur. The tube can be de-
stroyed within a few minutes under severe conditions.

Fortunately, the conditions that tend to minimize this back-bombardment by elec-

trons also characterize minimum driving voltage. The tendency of electrons to turn
back in flight is reduced by using the lowest possible RF grid voltage on the tube. This is
achieved by using the lowest possible dc grid bias. In tetrodes, this effect is inherently
lower because of the action of the screen in accelerating the electrons toward the anode.
A tetrode also permits the use of smaller grid voltages. Consequently, under favorable
conditions, the number of electrons turned back to heat the cathode and tube structure
can be kept to a practical, low level. In addition to low dc grid bias, high screen voltage
is desirable.

The plate circuit should always operate with heavy loading (low external plate im-

pedance) so that the minimum instantaneous value of plate voltage will stay suffi-
ciently positive to continue accelerating electrons to the anode. For this reason, the lon-
gest tube life is realized when the tetrode amplifier is heavily loaded, as indicated by
having small values of dc screen and dc control grid current.

Voltage Factor

Current Factor

0.25

0.125

0.50

0.35

0.75

0.65

1.00

1.00

1.25

1.40

1.50

1.84

1.75

2.30

2.00

2.80

2.25

3.40

2.50

4.00

2.75

4.60

3.00

5.20

Table 10.2 Three-Halves Current Values of Commonly Used Factors

© 1999 CRC Press LLC

Never operate a tube with light plate loading. If the plate load is removed so that the

minimum instantaneous plate voltage tends to fall to values approximating cathode po-
tential (as it must do when the loading is removed completely and excitation is present),
the number of electrons turned back can be destructive to the tube. It has been found
that under conditions of no loading, the electron bombardment and increased electric
field heating of the insulating portion of the tube is often sufficient to cause suck-in of
the glass or cracking of the ceramic envelope. Automatic protection should be provided
to remove all voltages from the tube if the plate circuit loading becomes too light for the
amount of excitation applied.

Note that parasitic oscillations seldom are loaded heavily, as indicated by the high

grid currents often observed during such self-oscillation. Thus, excessive RF plate
voltages are developed that, at VHF, can be as damaging as unloaded operation on a
VHF fundamental frequency. If unloaded VHF parasitic oscillations are present simul-
taneously with apparently satisfactory operation on the fundamental, unexplained re-
duction of tube life may result.

Occasionally, an output line circuit may resonate simultaneously to a harmonic fre-

quency as well as to the fundamental frequency. The higher resonant modes of practical
line circuits are not normally harmonically related. Sometimes, however, the tuning
curve of a mode will cross the fundamental tuning curve and, at that point, the circuit
will build up resonant voltages at both the harmonic and the fundamental frequency.
The harmonic resonance usually is lightly loaded and the damaging action is similar to
that of lightly loaded parasitic or fundamental. Again, while operation of the tube and
circuit on the fundamental may appear normal, but with lower than expected efficiency,
damage can occur to the tube.

In addition to operating the tube with minimum bias, high screen voltage, and heavy

loading on the plate circuit, some degree of compensation for the remaining back-heat-
ing of the cathode may be required. This can be accomplished by lowering the filament
voltage or heater voltage until the cathode operates at a normal temperature. In the case
of tetrodes and pentodes, by taking precautions necessary to minimize back-bombard-
ment by electrons, the required compensation for back-heating of the cathode is not sig-
nificant and may often be neglected.

10.3.2 Amplifier Balance

In a push-pull RF amplifier, lack of balance in the plate circuit or unequal plate dissi-
pation usually is caused by a lack of symmetry in the RF circuit [2]. Normally, the
tubes are similar enough that such imbalance is not associated with the tubes and their
characteristics. This can readily be checked by interchanging the tubes in the sockets
(provided both tubes have common dc voltages to plate, screen, and grid) and observ-
ing whether the unbalanced condition remains at the socket location or moves with
the tube. If it remains at the socket location, the circuit requires adjustment. If appre-
ciable imbalance is associated with the tube, it is likely that one tube is abnormal and
should be replaced.

The basic indicators of balance are the plate current and plate dissipation of each

tube. It is assumed that the circuit applies the same dc plate voltage, dc screen voltage

© 1999 CRC Press LLC

(in the case of a tetrode or pentode), and dc grid bias to each tube from common sup-
plies. Also, it is initially assumed that the plate circuit is mechanically and electrically
symmetrical (or approximately so).

Imbalance in a push-pull RF amplifier is usually the result of unequal RF voltages

applied to the grids of the tubes, or the RF plate circuit applying unequal RF voltages to
the plates of the tubes. Correction of this situation involves first balancing grid excita-
tion until equal dc plate currents flow in each tube. Next, the RF plate circuit is adjusted
until equal plate dissipation appears on each tube, or equal RF plate voltage is ob-
served. The balance of plate current is a more important criterion than equality of
screen current (in a tetrode or pentode) or grid current. This results from the fact that
tubes tend to be more uniform in plate current characteristics. However, the screen cur-
rent also is sensitive to a lack of voltage balance in the RF plate circuit and may be used
as an indicator. If the tubes differ somewhat in screen current characteristics, and the
circuit has common dc supply voltages, final trimming of the plate circuit balance may
be made by interchanging tubes and adjusting the circuit to give the same screen current
for each tube, regardless of its location.

Note that the dc grid current has not been used as an indicator of balance of the RF

power amplifier. It is probable that after following the foregoing procedure, the grid
currents will be fairly well balanced, but this condition, in itself, is not a safe indicator
of grid excitation balance.

10.3.3 Parallel Tube Amplifiers

The previous discussion has been oriented toward the RF push-pull amplifier. The
same comments can be directed to parallel tube RF amplifiers [2]. Balance--the con-
dition in which each tube carries its fair share of the load--still must be considered. In
low-frequency amplifiers operating in class AB

1

or class AB

2

, the idle dc plate current

per tube should be balanced by separate bias adjustments for each tube. In many
cases, some lack of balance of the plate currents will have negligible effect on the
overall performance of the amplifier.

When tubes are operating in the idle position close to cutoff, an exact balance is dif-

ficult to achieve because plate current cannot be held to a close percentage tolerance.
At this point, the action of the plate and screen voltages is in delicate balance with the
opposing negative grid voltage. The state of balance is indicated by the plate current.
Minor variations of individual grid wires or variations in the diameter of grid wires can
upset the balance. It is practically impossible to control such minor variations in manu-
facture.

10.3.4 Harmonic Energy

A pulse of plate current delivered by the tube to the output circuit contains compo-
nents of the fundamental and most harmonic frequencies [2]. To generate output
power that is an harmonic of the exciting voltage applied to the control grid, it is
merely necessary to resonate the plate circuit to the desired harmonic frequency. To
optimize the performance of the amplifier, it is necessary to adjust the angle of plate

© 1999 CRC Press LLC

current flow to maximize the desired harmonic. The shorter the length of the current
pulse, in the case of a particular harmonic, the higher the plate efficiency. However,
the bias, exciting voltage, and driving power must be increased. Also, if the pulse is
too long or too short, the output power will decrease appreciably.

The plate circuit efficiency of tetrode and pentode harmonic amplifiers is quite

high. In triode amplifiers, if feedback of the output harmonic occurs, the phase of the
voltage feedback usually reduces the harmonic content of the plate pulse, and thereby
lowers the plate circuit efficiency. Because tetrodes and pentodes have negligible feed-
back, the efficiency of a harmonic amplifier usually is comparable to that of other am-
plifier types. Also, the high amplification factor of a tetrode or pentode causes the plate
voltage to have little effect on the flow of plate current. A well-designed tetrode or
pentode also permits large RF voltages to be developed in the plate circuit while still
passing high peaks of plate current in the RF pulse. These two factors help to further in-
crease plate efficiency.

Controlling Harmonics

The previous discussion of harmonics has been for the case where harmonic power in
the load is the objective. The generation and radiation of harmonic energy must be
kept to a minimum in a fundamental frequency RF amplifier [2]. The ability of the
tetrode and pentode to isolate the output circuit from the input circuit over a wide
range of frequencies is important in avoiding feedthrough of harmonic voltages from
the grid. An important part of this shielding is the result of the basic physical design
of the devices. The following steps permit reduction of unwanted harmonic energy in
the output circuit:

·

Minimize the circuit impedance between plate and cathode. This objective may
be achieved by having some or all of the tuning capacitance of the resonant circuit
close to the tube.

·

Completely shield the input and output compartments of the stage.

·

Use inductive output coupling from the resonant plate circuit, and possibly a ca-
pacitive or Faraday shield between the coupling coil and the tank coil. A high-fre-
quency attenuating circuit, such as a Pi or Pi-L network, also may be used.

·

Incorporate low-pass filters on all supply leads and wires leading to the output
and input compartments.

·

Use resonant traps for particular frequencies as required.

·

Install a low-pass filter in series with the output transmission line.

10.3.5 Klystron Tuning Considerations

The output power stability of a klystron is sensitive to changes in the principal operat-
ing parameters, including:

·

Beam input power

© 1999 CRC Press LLC

·

RF drive power

·

RF drive frequency

·

Device tuning

·

Magnetic fields in the vicinity of the tube

·

VSWR of the load

Varying any of these parameters will affect the RF output power response.

Figure 10.26

illustrates how changes in the dc beam input power affect the RF output

power of a klystron under constant RF drive conditions. Small changes in dc input
power produce marked changes in RF output power.

Figure 10.27

shows RF output

power as a function of RF drive power applied to the tube. From this curve, it can be
seen that when the RF drive power is low, the RF output power is low. As the level of RF
drive power increases, RF output power increases until an optimum operating point is
reached. Beyond this point, further increases in RF drive power result in reduced RF
output power.

Klystrons are said to be saturated at the point where a further increase in RF drive

only decreases the RF output power. To obtain maximum RF output from a klystron,
sufficient RF drive must be applied to the tube to reach the point of saturation. Oper-
ating at RF drive levels beyond saturation will only overdrive the klystron, decrease RF
output power, and increase the amount of beam interception (body current) at the drift
tube.

Figure 10.26 RF output power variation as a function of dc beam input power for a kly-
stron.

© 1999 CRC Press LLC

Tuning of any cavity of a multicavity klystron will affect the overall shape of the am-

plitude-response curve. There are two common methods of tuning klystrons:

·

High-efficiency narrowband tuning

·

Broadband tuning, yielding lower gain and efficiency

Figure 10.28

shows how RF output power changes with various levels of RF drive ap-

plied to a klystron under different tuning conditions. The RF power at point A repre-
sents the drive saturation point for a synchronously tuned tube. Point B shows a new
point of saturation that is reached by tuning the penultimate (next to the last) cavity to
a somewhat higher frequency. By tuning the penultimate cavity still further, point C is
reached. At point D, increasing the penultimate cavity frequency no longer increases
RF output power; instead, it reduces the power delivered by the tube (curve E).

10.3.6 Intermodulation Distortion

Intermodulation distortion products in linear power amplifier circuits can be caused
by either amplitude gain nonlinearity or phase shift with change in input signal level.
Intermodulation distortion products appear when the RF signal has a varying enve-
lope amplitude. A single continuous-frequency wave will be amplified by a fixed
amount and shifted in phase by a fixed amount. The nonlinearity of the amplifier will

Figure 10.27 RF output power of a klystron as a function of RF drive power.

© 1999 CRC Press LLC

produce only harmonics of the input wave. As the input RF wave changes, the
nonlinearity of the amplifier will cause undesirable intermodulation distortion.

When an RF signal with varying amplitude is passed through a nonlinear device,

many new products are produced. The frequency and amplitude of each component can
be determined mathematically. The nonlinear device can be represented by a power se-
ries expanded about the zero-signal operating point. Intermodulation distortion energy
is wasted and serves no purpose other than to cause interference to adjacent channels.
Intermodulation distortion in a power amplifier tube is principally the result of its trans-
fer characteristics. An ideal transfer characteristic curve is shown in

Figure 10.29

.

Even-order products do not contribute to the intermodulation distortion problem be-

cause they usually fall outside the amplifier passband. Therefore, if the transfer charac-
teristic produces an even-order curvature at the small-signal end of the curve (from
point A to point B) and the remaining portion of the curve (point B to point C) is linear,
the tube is considered to have an ideal transfer characteristic. If the operating point of
the amplifier is set at point 0 (located midway horizontally between point A and point
B), there will be no distortion in a class AB amplifier. However, no tube has this ideal-
ized transfer characteristic. It is possible, by manipulation of the electron ballistics
within a given tube structure, to alter the transfer characteristic and minimize the dis-
tortion products.

Figure 10.28 Output power variation of a klystron as a function of drive power under dif-
ferent tuning conditions.

© 1999 CRC Press LLC

Generation of RF Intermodulation Products

When two or more transmitters are coupled to each other, new spectral components
are produced by mixing of the fundamental and harmonic terms of each of the desired
output frequencies.

1

The most common form of coupling results from transmitting an-

tennas that are physically closely spaced. If sufficient filtering is not provided at the
output stages and/or on one or both of the transmitter feedlines, energy from one
transmitter may travel back to the final power amplifier of the other. In the PA stage,
the desired and undesired signals mix, producing spurious intermodulation products.

Consider the following example involving two transmitters. The third-order

intermodulation products (IM3) are generated in the following manner:

·

The output of transmitter 1 (f

1

) is coupled into the nonlinear output stage of trans-

mitter 2 (f

2

). This condition is the result of insufficient isolation between the two

output signals.

·

Signal f

1

mixes with the second harmonic of f

2

, producing an in-band third-order

term with a frequency of 2f

2

f

1

at the output of transmitter 2.

·

In a similar manner, another third-order term will be produced at a frequency of
2f

1

f

2

at the output of transmitter 1.

Figure 10.29 Ideal grid-plate transfer curve for class AB operation.

1

Portions of this section were contributed by Geoffrey N. Mendenhall, P. E., Cincinnati, OH.

© 1999 CRC Press LLC

This implies that the second-harmonic content within the output stage of each trans-
mitter, along with the specific nonlinear characteristics of the output stage, will have
an effect on the value of the mixing loss.

It is also possible to generate these same third-order terms in another way. If the dif-

ference frequency between the two transmitters (f

2

f

1

), which is an out-of-band fre-

quency, remixes with either f

1

or f

2

, the same third-order intermodulation frequencies

will be produced.

Empirical measurements [3] indicate that the 2f

2

f

1

type of mechanism is the domi-

nant mode generating third-order IM products in RF generators using a tuned cavity for
the output network.

Figure 10.30

illustrates the mechanism of third-order IM for the

two-signal condition.

IM As a Function of Turnaround Loss

2

Turnaround or mixing loss describes the phenomenon whereby the interfering signal
mixes with the fundamental and its harmonics within the nonlinear output device.
This mixing occurs with a net conversion loss; hence, the term turnaround loss has
become widely used to quantify the ratio of the interfering level to the resulting IM3
level. A turn around loss of 10 dB means that the IM3 product fed back to the antenna
system will be 10 dB below the interfering signal fed into the transmitter output stage.

Turnaround loss will increase if the interfering signal falls outside the passband of

the transmitter output circuit, varying with frequency separation of the desired signal

Figure 10.30 Frequency spectrum of third-order IM with the interfering level equal to
the carrier level.

2

Mendenhall, Geoffrey N., "A Study of RF Intermodulation Between FM Broadcast
Transmitters Sharing Filterplexed or Colocated Antenna Systems, Broadcast Electronics,
Quincy, Il, 1983.

© 1999 CRC Press LLC

and the interfering signal. This is because the interfering signal is first attenuated by the
output circuit selectivity going into the nonlinear device, then the IM3 product is fur-
ther attenuated as it comes back through the frequency-selective circuit. Turnaround
loss, thus, can be divided into three principal components:

·

The basic in-band conversion loss of the nonlinear device

·

Attenuation of the out-of-band interfering signal resulting from the selectivity of
the output stage

·

Attenuation of the resulting out-of-band IM3 products resulting from the selec-
tivity of the output circuit

As the turnaround loss increases, the level of undesirable intermodulation products is
reduced, and the amount of isolation required between the transmitters also is re-
duced.

Transmitter output circuit loading directly affects the power amplifier source im-

pedance and, therefore, also affects the efficiency of coupling the interfering signal into
the output circuit where it mixes with the other frequencies present to yield IM3 prod-
ucts. Light loading reduces the amount of interference that enters the output circuit
with a resulting increase in turnaround loss. In addition, output loading will change the
output circuit bandwidth (loaded Q) and, therefore, also affect the amount of attenua-
tion that out-of-band signals will encounter in passing into and out of the output circuit.

Second-harmonic traps or low-pass filters in the transmission line of either trans-

mitter have little effect on the generation of IM products. This is because the harmonic
content of the interfering signal entering the output circuit of the transmitter has much
less effect on IM3 generation than the harmonic content within the nonlinear device it-
self. The interfering signal and the resulting IM3 products thus fall within the passband
of the low-pass filters and outside the reject band of the second-harmonic traps; these
devices typically offer little, if any, attenuation to RF intermodulation products.

Site-related intermodulation problems and typical solutions are discussed later in

this chapter.

10.3.7 VSWR

Among all the possible load measurements, VSWR is the most common. An in-line
directional wattmeter commonly is used to measure both the forward and reflected
power in a feedline. It is the comparison of reflected power (P

r

) to forward power (P

f

)

that provides a qualitative measure of the match between a transmitter and its
feedline/load.

The standing wave ratio (SWR) on a transmission line is related to the ratio of the for-

ward to reflected power by the following:

© 1999 CRC Press LLC

SWR =

1

1

+

-

P

P

P

P

r

f

r

f

(10.14)

The standing wave is the result of the presence of two components of power: one

traveling toward the load and the other reflected by a load mismatch, traveling back to-
ward the generator. These components are defined by the following:

( )

P

E

Z

I

Z

f

f

o

f

o

=

=

2

2

(10.15)

( )

P

E

Z

I

Z

r

r

o

r

o

=

=

2

2

(10.16)

(

)

P

P

P

n

f

r

=

-

(10.17)

Where:
P

f

= forward power

P

r

= reflected power

E

f

= forward voltage

E

r

= reflected voltage

I

f

= forward current

I

r

= reflected current

Z

o

= characteristic impedance of the line

P

n

= net power absorbed by the load (transmission line loss and antenna radiation)

Because the forward and reflected voltages and currents are traveling in opposite di-

rections, they will add in phase at some point along the line to produce a voltage maxi-
mum. At wavelength along the line in either direction from this maximum, the forward
and reflected components will be out of phase, and will produce a voltage minimum.
The forward and reflected components of current also add vectorally to produce a cur-
rent standing wave. The magnitude of this wave is defined by:

SWR =

E

E

I

I

max

min

max

min

=

(10.18)

The ratio of the highest voltage point on the line to the lowest voltage point on the

line commonly is used to evaluate system performance. This property is known as the
voltage standing wave ratio (VSWR). A VSWR of 1.0:1.0 means that a perfect match
has been achieved. A VSWR of 2.0:1.0 means that a mismatch is causing power to be
reflected back from the load, approximately 11 percent in this particular case.

© 1999 CRC Press LLC

At the point of reflection (the load mismatch), the phase of the reflected current is re-

versed 180° from the forward current. The reflected voltage does not exhibit this phase
reversal. This displaces the voltage and current standing waves by 90° along the line so
that the E

max

and I

min

occur at the same points, while E

min

and I

max

occur 90° in either direc-

tion from E

max

and I

min

.

The fact that the reflected current is reversed in phase makes it possible to measure

forward and reflected power separately using a directional coupler device. A small volt-
age is obtained by inductive coupling that represents the current in the transmission
line. To this signal is added a sample of the voltage across the line, simultaneously ob-
tained by capacitive coupling. These two samples are adjusted to be exactly equal when
the line is terminated with its characteristic impedance. The two RF samples, when
added, yield a resultant RF voltage proportional to the forward components of voltage
and current. Reflected power is measured by installing a second coupling section physi-
cally turned in the opposite direction. The phase of the current sample is reversed, and
the reflected components add while the forward components balance out.

On a theoretical, lossless feedline, VSWR is the same at all points along the line.

Actually, all feedlines have some attenuation distributed uniformly along their entire
lengths. Any attenuation between the reflection point and the source reduces VSWR at
the source. This concept is illustrated in

Figure 10.31

. Standing waves on the feedline

increase feedline attenuation to a level greater than normal attenuation under
matched-line conditions. The amount of feedline attenuation increase depends on the
VSWR and the extent of losses under matched-line conditions. The higher the
matched-line attenuation, the greater the additional attenuation caused by a given
VSWR.

Figure 10.32

correlates additional attenuation with matched-line attenuation

for various VSWR levels.

When a feedline is terminated by an impedance equal to the feedline characteristic

impedance, the feedline impedance at any point is equal to the feedline characteristic
impedance. When the feedline is terminated by an impedance not equal to the feedline
characteristic impedance, the feedline input impedance will vary with changes in
feedline length. The greater the impedance mismatch at the load, the wider the feedline
input impedance variation with changes in line length. The largest feedline impedance
change occurs with the worst mismatches: open circuit or short circuit.

A short circuit at the end of a feedline appears as an open circuit at a point wave-

length down the feedline. It is the greatest impedance transformation that can occur on
a feedline. Less severe mismatches cause smaller impedance transformations. At a
point wavelength down the line, the impedance equals the load impedance. Feedlines
with high VSWR sometimes are called resonant or tuned feedlines. Matched
lines--those with little VSWR--are referred to as nonresonant or untuned feedlines.
This phenomenon means that high feedline VSWR can cause the feedline input to pres-
ent a load impedance to the output tube that differs greatly from the feedline character-
istic impedance. Some RF generators incorporate an output matching network that pro-
vides a conjugate match to a broad range of impedances.

© 1999 CRC Press LLC

10.4 RF System Performance

Assessing the performance of an RF generator has been greatly simplified by recent
advancements in system design. The overall performance of most transmitters can be
assessed by measuring the audio, video, or data performance of the system as a
whole. If these "intelligence signal" tests indicate good performance, chances are
good that all of the RF parameters also meet specifications. Both AM and FM trans-
mission systems have certain limitations that prevent them from ever being a com-
pletely transparent medium. Through proper system design and regular equipment
maintenance, however, overall performance can be maintained at acceptable levels in
most operational environments.

Figure 10.31 Illustration of how the VSWR on a transmission line affects the attenua-
tion performance of the line. Scale

A represents the VSWR at the transmitter; scale B

represents the corresponding reflected power in watts for an example system.

© 1999 CRC Press LLC

10.4.1 Key System Measurements

The procedures for measuring transmitter performance vary widely, depending on the
type of equipment being used. Certain basic measurements, however, can be applied
to characterize the overall performance of most systems:

·

Frequency response: the actual deviation from a constant amplitude across a
given span of frequencies.

·

Total harmonic distortion: the creation, by a nonlinear device, of spurious signals
harmonically related to the applied waveform. THD is sensitive to the noise floor
of the system under test. If the system has a signal-to-noise ratio of 60 dB, the dis-
tortion analyzer's best possible reading will be higher than 0.1 percent (60 dB =
0.001 = 0.1 percent).

·

Intermodulation distortion: the creation, by a nonlinear device, of spurious sig-
nals not harmonically related to the audio waveform. These distortion compo-
nents are sum-and-difference (beat note) mixing products. The IM measurement
is relatively impervious to the noise floor of the system under test.

Figure 10.32 Illustration of how VSWR causes additional loss in a length of feedline.

© 1999 CRC Press LLC

·

Signal-to-noise ratio (S/N): the amplitude difference, expressed in decibels, be-
tween a reference-level audio signal and the residual noise and hum of the overall
system.

·

Crosstalk: the amplitude difference, expressed in decibels, between two or more
channels of a multichannel transmission system.

10.4.2 Synchronous AM in FM Systems

3

The output spectrum of an FM transmitter contains many sideband frequency compo-
nents, theoretically an infinite number. The practical considerations of transmitter de-
sign and frequency allocation make it necessary to restrict the bandwidth of FM sig-
nals. Bandwidth restriction brings with it a number of undesirable side effects, includ-
ing:

·

Phase shifts through the transmission chain

·

Generation of synchronous AM components

·

Distortion in the demodulated output of certain receivers

In most medium- and high-power FM transmitters, the primary out-of-band filtering
is performed in the output cavity of the final stage. Previous stages in the transmitter
(the exciter and IPA) are designed to be broadband, or at least more broadband than
the PA. As the bandwidth of an FM transmission system is reduced, synchronous am-
plitude modulation increases for a given carrier deviation (modulation). Synchronous
AM is produced as tuned circuits with finite bandwidth are swept by the frequency of
modulation. The amount of synchronous AM generated is dependent on tuning,
which determines (to a large extent) the bandwidth of the system.

Figure 10.33

illus-

trates how synchronous AM is generated through modulation of the FM carrier.

Bandwidth Limiting

The design goal of most medium- and high-power FM transmission equipment is to
limit the bandwidth of the transmitted RF signal at one stage only. In this way, control
over system bandwidth can be tightly maintained, and the tradeoffs required in any
practical FM system can be optimized. If, on the other hand, more than one
narrowband stage exists within the transmitter, adjustment for peak efficiency and
performance can be a difficult proposition. The following factors can affect the band-
width of an FM system:

·

Total number of tuned circuits in the system

3

Mendenhall, Geoffrey N., "Techniques for Measuring Synchronous AM Noise in FM
Transmitters," Broadcast Electronics, Quincy, IL, 1988.

© 1999 CRC Press LLC

·

Amplitude and phase response of the total combination of all tuned circuits in the
RF path

·

Amount of drive to each RF stage (saturation effects)

·

Nonlinear transfer function within each RF stage

The following techniques can be used to improve the bandwidth of a given system:

·

Maintain wide bandwidth until the final RF stage. This can be accomplished
through the use of a broadband exciter and broadband (usually solid-state) IPA.

·

Minimize the number of interactive-tuned networks within the system. Through
proper matching network design, greater bandwidth and simplified tuning can be
accomplished.

Bandwidth affects the gain and efficiency of any FM RF amplifier. The bandwidth

is determined by the load resistance across the tuned circuit and the output or input ca-
pacitance of the RF stage. Because grid-driven PA tubes typically exhibit high input ca-
pacitance, the PA input circuit generally has the greatest effect on bandwidth limiting in
an FM transmitter. The grid circuit can be broadened through resistive swamping in the
input circuit, or through the use of a broadband input impedance matching network.
One approach involves a combination of series inductor and shunt capacitor elements

Figure 10.33 The generation of synchronous AM in a bandwidth-limited FM system.
Note that minimum synchronous AM occurs when the system operates in the center of
its passband.

© 1999 CRC Press LLC

implemented on a printed wiring board, as illustrated in

Figure 10.34

. In this design, the

inductors and capacitors are etched into the copper-clad laminate of the PWB. This ap-
proach provides the necessary impedance transformation from a 50

solid-state driver

to a high-impedance PA grid in a series of small steps, rather than more conventional L,
Pi, or Tee matching networks.

Effects of Synchronous AM

The effect of bandwidth on synchronous AM performance for an example FM RF
amplifier is plotted in

Figure 10.35

. Notice that as the 3 dB points of the RF system

passband are narrowed, a dramatic increase in synchronous AM occurs. The effect
shown in the figure is applicable to any bandwidth-limited FM system.

Bandwidth also affects the distortion floor of the demodulated signal of an FM sys-

tem, as plotted in

Figure 10.36

. The data shown apply to a test setup involving a sin-

gle-tuned circuit fed by an FM generator with a deviation of ±75 kHz at a modulating
frequency of 15 kHz (no de-emphasis is applied to the output signal). Remember that
FM is a nonlinear process and that interpretation of distortion numbers for multi-
ple-carrier signals is not accurate. The example shown, however, illustrates how the
bandwidth of the RF channel can set a minimum performance limit on system total har-
monic distortion. By understanding the mechanics of synchronous AM, adjustment for
optimum performance of an FM system can be accomplished.

10.4.3 Incidental Phase Modulation

Incidental phase modulation (IPM) is phase modulation produced by an AM trans-
mitter as a result of amplitude modulation. In other words, as an AM transmitter de-
velops the amplitude-modulated signal, it also produces a phase-modulated, or PM,
version of the input intelligence. In theory, IPM is of little consequence for classic
AM applications because FM and PM do not affect the carrier amplitude. An enve-
lope detector will, in theory at least, ignore IPM. However, a number of applications
utilize manipulation of the carrier phase to convey information in addition to ampli-
tude modulation of the carrier. Receivers designed to decode such AM/PM wave-
forms will not perform properly if excessive IPM is produced by the transmitter. Al-

Figure 10.34 IPA plate-to-PA-grid matching network designed for wide bandwidth op-
eration.

© 1999 CRC Press LLC

though the effects of IPM vary from one system to another, optimum performance is
realized when IPM is minimized.

As a general rule, because IPM is a direct result of the modulation process, it can be

generated in any stage that is influenced by modulation. The most common cause of
IPM in plate-modulated and pulse-modulated transmitters is improper neutralization
of the final RF amplifier. Adjusting the transmitter for minimum IPM is an accurate
way of achieving proper neutralization. The reverse is not always true, however, be-
cause some neutralization methods will not necessarily result in the lowest IPM.

Improper tuning of the IPA stage is the second most common cause of IPM. As mod-

ulation changes the driver loading to the PA grid, the driver output also may vary. The
circuits that feed the driver stage usually are sufficiently isolated from the PA that they
do not produce IPM. An exception is if the power supply for the IPA is influenced by
modulation. Such a problem could be caused by a lack of adequate capacitance in the
filters of the high-voltage power supply.

Bandwidth Considerations

When a perfect AM transmitter is modulated with a tone, two sidebands are created,
one above the carrier and one below it. Phase modulation of the same carrier, how-
ever, produces an infinite number of sidebands at intervals above and below the car-
rier. The number of significant sidebands depends on the magnitude of the modula-
tion. With moderate amounts of IPM, the transmitted spectrum can be increased to the
point at which the legal channel limits are exceeded.

Figure 10.35 Synchronous AM content of an example FM system as a function of sys-
tem bandwidth.

© 1999 CRC Press LLC

Corrective Procedures

The spectrum analyzer is generally the best instrument for identifying IPM compo-
nents. The transmitter is first modulated with a pure sine wave, and a THD measure-
ment is taken with a distortion analyzer. These data will provide an idea of what the
spectrum analyzer display should look like when it is connected to the transmitter. If
the transmitter has an excessive amount of harmonic distortion, it will be impossible
to tell the difference between the normal distortion sidebands and the IPM sidebands.

Figure 10.37

shows the trace of an ideal transmitted spectrum on a spectrum ana-

lyzer when modulated with a 1 kHz tone at 100 percent modulation. Notice that there
are just three components: one carrier and two sidebands. This transmitter would mea-
sure 0 percent distortion, because no harmonic sidebands are present. Also, no IPM is
present, because it would show up as extra sidebands in the spectrum.

If the transmitter had an IPM problem, the spectrum analyzer display might look like

Figure 10.38

. The distortion analyzer still would read 0 percent, but the sidebands on

the spectrum analyzer would not agree. The key is to start with a low harmonic distor-
tion reading on the transmitter so that any sidebands will be recognizable immediately
as the result of IPM.

Figure 10.36 Total harmonic distortion (THD) content of an example demodulated FM
signal as a function of transmission-channel bandwidth.

© 1999 CRC Press LLC

10.4.4 Carrier Amplitude Regulation

Carrier amplitude regulation (carrier level shift) is the effective shift in apparent car-
rier level as a result of the amplitude modulation process [4]. Carrier level shift can be
caused by the following:

·

Poor power supply regulation

·

Modulation-related even order harmonic distortion that generates a dc offset
component in the modulated RF envelope

·

A combination of the previous two mechanisms

Carrier level shift can be either positive or negative, although is usually negative be-
cause power supply regulation is most often the major source of carrier level shift, and
power supply regulation is generally negative in sign (i.e., lower voltage output at
higher current loads). Strictly speaking, carrier level shift is defined as the shift in ef-
fective carrier voltage or current resulting from modulation. (See.

Figure 10.39

.)

10.4.5 Site-Related Intermodulation Products

As discussed previously in this chapter, an intermodulation product (intermod) is a
signal created when two or more signals are unintentionally mixed to produce a new
signal or group of signals. Mixing can occur in transmitter circuitry, in receiver cir-
cuitry, or elsewhere. The apparent effect is the same: one or more low-level products
strong enough to interfere with desired signals and prevent proper reception. Some-

Figure 10.37 The spectrum of an ideal AM transmitter. THD measures 0 percent; there
are no harmonic sidebands.

© 1999 CRC Press LLC

times the intermod falls on unused frequencies. Sometimes it falls on a channel in use
or near enough to the channel to cause interference.

It is possible to calculate the intermod that may result from frequencies in use at a

given transmission site. Normally produced by a computerized analysis, this intermod
study is helpful in several ways. If the study is conducted as one or more new frequen-
cies are under consideration for future use at an existing site, it may reveal which fre-
quencies are more likely to produce a harmful intermod and which are not. If the study
is conducted as part of an investigation into an interference problem, it may help in
identifying transmitters that play a role in generating the intermod.

Intermod can be generated by any nonlinear device or circuit. The most common

place for intermod to be generated is in the power amplifier of an FM transmitter. Such
an amplifier does not need to preserve waveform integrity; the waveform is restored by
the tuned output circuit. Thus, the amplifier is operated in an efficient but nonlinear
mode, class C. In contrast, many amplifiers for AM service operate in a linear mode.
Such amplifiers normally do not act as mixers.

Generally, solid-state amplifiers are better mixers than their electron tube counter-

parts, a fact that may explain why intermod may appear suddenly when older equip-
ment is replaced. A mix occurs when RF energy from one transmitter (Tx 1) finds its
way down the transmission line and into the final amplifier of another transmitter (Tx
2), as illustrated in

Figure 10.40

. The energy mixes in the final amplifier with the

prefiltered second harmonic of Tx 2 to produce another in-band signal, a third-order
intermod product. The reverse mix also may occur in the final amplifier of Tx 1, pro-
ducing another intermod signal.

After the cause is discovered, a mix generated in a transmitter final amplifier is rela-

tively easy to eliminate: Increase the isolation between the offending transmitters. This

Figure 10.38 The spectrum of an ideal transmitter with the addition of IPM.

© 1999 CRC Press LLC

may be accomplished by increasing the space between the antennas, or by placing an
isolator at the output of each transmitter.

Receiver Intermod

Intermod also may be generated in the front end of a receiver. High-level RF energy
from transmitters at a site can generate a bias voltage in the receiver sufficient to
place the receiver RF amplifier into a nonlinear state. In such a mode, the stage acts as
a mixer and creates unwanted frequencies. Filters may be used to solve this problem.
The filters should notch out the offending transmitted signals. If another transmitter is
added to the site, another notch filter usually must be added to the receiver transmis-
sion line.

Hardware Intermod

At a large transmission site, the entire area--including towers, guy wires, antennas,
clamps, and obstruction lighting hardware--is saturated by high-level RF. Any loose
or moving hardware can cause broadband noise. Corroded connections, tower joints,
or clamps can form semiconductor junctions that may generate harmful intermod.
Because the antenna system is exposed to a changing external environment, the de-

Figure 10.39 Carrier amplitude regulation (carrier level shift) defined. (

From [4]. Used

with permission.)

© 1999 CRC Press LLC

gree of semiconductivity can vary with moisture or temperature. Changing
semiconductivity causes intermittent problems, at changing levels of severity.

10.5 References

1. Hulick, Timothy P., "Very Simple Out-of-Band IMD Correctors for Adjacent Chan-

nel NTSC/DTV Transmitters," Digital Television `98, Intertec Publishing, Over-
land Park, KS, 1998.

2. Laboratory Staff, Care and Feeding of Power Grid Tubes, Varian/Eimac, San

Carlos, CA, 1984.

3. Mendenhall, Geoffrey N., M. Shrestha, and E. Anthony, "FM Broadcast Transmit-

ters," NAB Engineering Handbook, 8th Ed., E. B. Crutchfield (ed.), National Asso-
ciation of Broadcasters, Washington, DC, pp. 463 - 464, 1992.

4. George Woodard, "AM Transmitters," in NAB Engineering Handbook, 9th Ed.,

Jerry C. Whitaker (ed.), National Association of Broadcasters, Washington, DC,
1998.

10.6 Bibliography

Benson, K. B., and Jerry C. Whitaker, Television and Audio Handbook for Engineers

and Technicians, McGraw-Hill, New York, 1989.

Bordonaro, Dominic, "Reducing IPM in AM Transmitters," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, May 1988.

Crutchfield, E. B. (ed.), NAB Engineering Handbook, 8th Ed., National Association of

Broadcasters, Washington, DC, 1992.

Figure 10.40 The mechanics of intermod generation at a two-transmitter installation.

© 1999 CRC Press LLC

Dolgosh, Al, and Ron Jakubowski, "Controlling Interference at Communications

Sites," Mobile Radio Technology, Intertec Publishing, Overland Park, KS, Febru-
ary 1990.

Hershberger, David, and Robert Weirather, "Amplitude Bandwidth, Phase Bandwidth,

Incidental AM, and Saturation Characteristics of Power Tube Cavity Amplifiers for
FM," application note, Harris Corporation, Quincy, IL, 1982.

Kinley, Harold, "Base Station Feedline Performance Testing," Mobile Radio Technol-

ogy, Intertec Publishing, Overland Park, KS, April 1989.

Mendenhall, Geoffrey N., "A Study of RF Intermodulation Between FM Broadcast

Transmitters Sharing Filterplexed or Co-located Antenna Systems," Broadcast
Electronics, Quincy, IL, 1983.

Mendenhall, Geoffrey N., "A Systems Approach to Improving Subcarrier Perfor-

mance," Proceedings of the 1986 NAB Engineering Conference, National Associa-
tion of Broadcasters, Washington, DC, 1986.

Mendenhall, Geoffrey N., "Fine Tuning FM Final Stages," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, May 1987.

Mendenhall, Geoffrey N., "Improving FM Modulation Performance by Tuning for

Symmetrical Group Delay," Broadcast Electronics, Quincy IL, 1991.

Mendenhall, Geoffrey N., "Techniques for Measuring Synchronous AM Noise in FM

Transmitters," Broadcast Electronics, Quincy IL, 1988.

Mendenhall, Geoffrey N., "Testing Television Transmission Systems for Multichan-

nel Sound Compatibility," Broadcast Electronics, Quincy, IL, 1985.

Terman, F. E., Radio Engineering, 3rd Ed., McGraw-Hill, New York, 1947.
Whitaker, Jerry C., Maintaining Electronic Systems, CRC Press, Boca Raton, FL,

1992.

Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1990.

Witte, Robert A., "Distortion Measurements Using a Spectrum Analyzer," RF Design,

Cardiff Publishing, Denver, CO, pp. 75 - 84, September 1992.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

11 Safe Handling of Vacuum Tube Devices"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

11

Safe Handling of Vacuum Tube

Devices

11.1 Introduction

Electrical safety is important when working with any type of electronic hardware. Be-
cause vacuum tubes operate at high voltages and currents, safety is doubly important.
The primary areas of concern, from a safety standpoint, include:

·

Electric shock

·

Nonionizing radiation

·

Beryllium oxide (BeO) ceramic dust

·

Hot surfaces of vacuum tube devices

·

Polychlorinated biphenyls (PCBs)

11.2 Electric Shock

Surprisingly little current is required to injure a person. Studies at Underwriters Labo-
ratories (UL) show that the electrical resistance of the human body varies with the
amount of moisture on the skin, the muscular structure of the body, and the applied
voltage. The typical hand-to-hand resistance ranges between 500

and 600 k, de-

pending on the conditions. Higher voltages have the capability to break down the
outer layers of the skin, which can reduce the overall resistance value. UL uses the
lower value, 500

, as the standard resistance between major extremities, such as

from the hand to the foot. This value is generally considered the minimum that would
be encountered and, in fact, may not be unusual because wet conditions or a cut or
other break in the skin significantly reduces human body resistance.

© 1999 CRC Press LLC

11.2.1 Effects on the Human Body

Table 11.1

lists some effects that typically result when a person is connected across a

current source with a hand-to-hand resistance of 2.4 k

. The table shows that a cur-

rent of approximately 50 mA will flow between the hands, if one hand is in contact
with a 120 V ac source and the other hand is grounded. The table indicates that even
the relatively small current of 50 mA can produce ventricular fibrillation of the heart,
and perhaps death. Medical literature describes ventricular fibrillation as rapid, unco-
ordinated contractions of the ventricles of the heart, resulting in loss of synchroniza-
tion between heartbeat and pulse beat. The electrocardiograms shown in

Figure 11.1

compare a healthy heart rhythm with one in ventricular fibrillation. Unfortunately,
once ventricular fibrillation occurs, it will continue. Barring resuscitation techniques,
death will ensue within a few minutes.

The route taken by the current through the body has a significant effect on the degree

of injury. Even a small current, passing from one extremity through the heart to another
extremity, is dangerous and capable of causing severe injury or electrocution. There are
cases where a person has contacted extremely high current levels and lived to tell about
it. However, usually when this happens, the current passes only through a single limb
and not through the body. In these instances, the limb is often lost, but the person sur-
vives.

Current is not the only factor in electrocution.

Figure 11.2

summarizes the relation-

ship between current and time on the human body. The graph shows that 100 mA flow-
ing through a human adult body for 2 s will cause death by electrocution. An important
factor in electrocution, the let-go range, also is shown on the graph. This range is de-
scribed as the amount of current that causes "freezing", or the inability to let go of the
conductor. At 10 mA, 2.5 percent of the population will be unable to let go of a "live"
conductor. At 15 mA, 50 percent of the population will be unable to let go of an ener-
gized conductor. It is apparent from the graph that even a small amount of current can

Table 11.1 The Effects of Current on the Human Body

Current

Effect

1 mA or less

No sensation, not felt

More than 3 mA

Painful shock

More than 10 mA

Local muscle contractions, sufficient to cause "freez-
ing" to the circuit for 2.5 percent of the population

More than 15 mA

Local muscle contractions, sufficient to cause "freez-
ing" to the circuit for 50 percent of the population

More than 30 mA

Breathing is difficult, can cause unconsciousness

50 mA to 100 mA

Possible ventricular fibrillation

100 mA to 200 mA

Certain ventricular fibrillation

More than 200 mA

Severe burns and muscular contractions; heart more
apt to stop than to go into fibrillation

More than a few amperes

Irreparable damage to body tissue

© 1999 CRC Press LLC

"freeze" someone to a conductor. The objective for those who must work around elec-
tric equipment is how to protect themselves from electric shock.

Table 11.2

lists re-

quired precautions for personnel working around high voltages.

11.2.2 Circuit Protection Hardware

The typical primary panel or equipment circuit breaker or fuse will not protect a per-
son from electrocution. In the time it takes a fuse or circuit breaker to blow, someone
could die. However, there are protection devices that, properly used, may help prevent
electrocution. The ground-fault current interrupter (GFCI), shown in

Figure 11.3

,

works by monitoring the current being applied to the load. The GFI uses a differential
transformer and looks for an imbalance in load current. If a current (5 mA, ±1 mA)
begins to flow between the neutral and ground or between the hot and ground leads,
the differential transformer detects the leakage and opens up the primary circuit
within 2.5 ms.

GFIs will not protect a person from every type of electrocution. If the victim be-

comes connected to both the neutral and the hot wire, the GFI will not detect an imbal-
ance.

11.2.3 Working with High Voltage

Rubber gloves are commonly used by engineers working on high-voltage equipment.
These gloves are designed to provide protection from hazardous voltages or RF when

Figure 11.1 Electrocardiogram of a human heartbeat: (

a) healthy rhythm, (b) ventricu-

lar fibrillation.

(

a)

(

b)

© 1999 CRC Press LLC

the wearer is working on "hot" ac or RF circuits. Although the gloves may provide
some protection from these hazards, placing too much reliance on them can have di-
sastrous consequences. There are several reasons why gloves should be used with a
great deal of caution and respect. A common mistake made by engineers is to assume
that the gloves always provide complete protection. The gloves found in many facili-
ties may be old or untested. Some may show signs of user repair, perhaps with electri-
cal tape. Few tools could be more hazardous than such a pair of gloves.

Another mistake is not knowing the voltage rating of the gloves. Gloves are rated dif-

ferently for both ac and dc voltages. For example, a class 0 glove has a minimum dc
breakdown voltage of 35 kV; the minimum ac breakdown voltage, however, is only 6
kV. Furthermore, high-voltage rubber gloves are not usually tested at RF frequencies,

Figure 11.2 Effects of electric current and time on the human body. Note the "let-go"
range.

© 1999 CRC Press LLC

and RF can burn a hole in the best of them. It is possible to develop dangerous working
habits by assuming that gloves will offer the required protection.

Gloves alone may not be enough to protect an individual in certain situations. Recall

the axiom of keeping one hand in a pocket while working around a device with current
flowing? That advice is actually based on simple electricity. It is not the "hot" connec-
tion that causes the problem, but the ground connection that lets the current begin to
flow. Studies have shown that more than 90 percent of electric equipment fatalities oc-
curred when the grounded person contacted a live conductor. Line-to-line electrocution
accounted for less than 10 percent of the deaths.

When working around high voltages, always look for grounded surfaces. Keep

hands, feet, and other parts of the body away from any grounded surface. Even concrete

Table 11.2 Required Safety Practices for Engineers Working Around High-Voltage
Equipment

High-Voltage Precautions

3 Remove all ac power from the equipment. Do not rely on internal contactors or

SCRs to remove dangerous ac.

3 Trip the appropriate power distribution circuit breakers at the main breaker

panel.

3 Place signs as needed to indicate that the circuit is being serviced.
3 Switch the equipment being serviced to the local control mode as provided.
3 Discharge all capacitors using the discharge stick provided by the manufac-

turer.

3 Do not remove, short circuit, or tamper with interlock switches on access cov-

ers, doors, enclosures, gates, panels, or shields.

3 Keep away from live circuits.

Figure 11.3 Basic design of a ground-fault interrupter (GFI).

© 1999 CRC Press LLC

can act as a ground if the voltage is sufficiently high. If work must be performed in
"live" cabinets, then consider using, in addition to rubber gloves, a rubber floor mat,
rubber vest, and rubber sleeves. Although this may seem to be a lot of trouble, consider
the consequences of making a mistake. Of course, the best troubleshooting methodol-
ogy is never to work on any circuit without being certain that no hazardous voltages are
present. In addition, any circuits or contactors that normally contain hazardous volt-
ages should be firmly grounded before work begins.

RF Considerations

Engineers often rely on electrical gloves when making adjustments to live RF circuits.
This practice, however, can be extremely dangerous. Consider the typical load match-
ing unit shown in

Figure 11.4

. In this configuration, disconnecting the coil from either

L2 or L3 places the full RF output literally at the engineer's fingertips. Depending on
the impedances involved, the voltages can become quite high, even in a circuit that
normally is relatively tame.

In the

Figure 11.4

example, assume that the load impedance is approximately 106

+j202

. With 1 kW feeding into the load, the rms voltage at the matching output will be

approximately 700 V. The peak voltage (which determines insulating requirements)
will be close to 1 kV, and perhaps more than twice that if the carrier is being ampli-
tude-modulated. At the instant the output coil clip is disconnected, the current in the
shunt leg will increase rapidly, and the voltage easily could more than double.

11.2.4 First Aid Procedures

All engineers working around high-voltage equipment should be familiar with first
aid treatment for electric shock and burns. Always keep a first aid kit on hand at the
facility.

Figure 11.5

illustrates the basic treatment for victims of electric shock. Copy

Figure 11.4 Example of how high voltages can be generated in an RF load matching
unit.

© 1999 CRC Press LLC

Figure 11.5 Basic first aid treatment for electric shock.

© 1999 CRC Press LLC

the information, and post it in a prominent location. Better yet, obtain more detailed
information from the local heart association or Red Cross chapter. Personalized in-
struction on first aid usually is available locally.

11.3 Operating Hazards

A number of potential hazards exist in the operation and maintenance of high-power
vacuum tube RF equipment. Maintenance personnel must exercise extreme care
around such hardware. Consider the following guidelines:

·

Use caution around the high-voltage stages of the equipment. Many power tubes
operate at voltages high enough to kill through electrocution. Always break the
primary ac circuit of the power supply, and discharge all high-voltage capacitors.

·

Minimize exposure to RF radiation. Do not permit personnel to be in the vicinity
of open, energized RF generating circuits, RF transmission systems
(waveguides, cables, or connectors), or energized antennas. High levels of radia-
tion can result in severe bodily injury, including blindness. Cardiac pacemakers
may also be affected.

·

Avoid contact with beryllium oxide (BeO) ceramic dust and fumes. BeO ceramic
material may be used as a thermal link to carry heat from a tube to the heat sink.
Do not perform any operation on any BeO ceramic that might produce dust or
fumes, such as grinding, grit blasting, or acid cleaning. Beryllium oxide dust and
fumes are highly toxic, and breathing them can result in serious injury or death.
BeO ceramics must be disposed of as prescribed by the device manufacturer.

·

Avoid contact with hot surfaces within the equipment. The anode portion of most
power tubes is air-cooled. The external surface normally operates at a high tem-
perature (up to 250°C). Other portions of the tube also may reach high tempera-
tures, especially the cathode insulator and the cathode/heater surfaces. All hot
surfaces may remain hot for an extended time after the tube is shut off. To prevent
serious burns, avoid bodily contact with these surfaces during tube operation and
for a reasonable cool-down period afterward.

Table 11.3

lists basic first aid pro-

cedures for burns.

11.3.1 OSHA Safety Considerations

The U.S. government has taken a number of steps to help improve safety within the
workplace under the auspices of the Occupational Safety and Health Administration
(OSHA). The agency helps industries monitor and correct safety practices. OSHA
has developed a number of guidelines designed to help prevent accidents. OSHA re-
cords show that electrical standards are among the most frequently violated of all
safety standards.

Table 11.4

lists 16 of the most common electrical violations, includ-

ing exposure of live conductors, improperly labeled equipment, and faulty grounding.

© 1999 CRC Press LLC

Protective Covers

Exposure of live conductors is a common safety violation. All potentially dangerous
electric conductors should be covered with protective panels. The danger is that
someone may come into contact with the exposed current-carrying conductors. It is
also possible for metallic objects such as ladders, cable, or tools to contact a hazard-
ous voltage, creating a life-threatening condition. Open panels also present a fire haz-
ard.

Identification and Marking

Circuit breakers and switch panels should be properly identified and labeled. Labels
on breakers and equipment switches may be many years old and may no longer reflect
the equipment actually in use. This is a safety hazard. Casualties or unnecessary dam-
age can be the result of an improperly labeled circuit panel if no one who understands
the system is available in an emergency. If a number of devices are connected to a sin-
gle disconnect switch or breaker, a diagram should be provided for clarification. La-
bel with brief phrases, and use clear, permanent, and legible markings.

Equipment marking is a closely related area of concern. This is not the same thing as

equipment identification. Marking equipment means labeling the equipment breaker

Table 11.3 Basic First Aid Procedures for Burns (More detailed information can be ob-
tained from any Red Cross office.)

Extensively Burned and Broken Skin

3 Cover affected area with a clean sheet or cloth.
3 Do not break blisters, remove tissue, remove adhered particles of clothing, or

apply any salve or ointment.

3 Treat victim for shock as required.
3 Arrange for transportation to a hospital as quickly as possible.
3 If arms or legs are affected, keep them elevated.
3 If medical help will not be available within an hour and the victim is conscious

and not vomiting, prepare a weak solution of salt and soda: 1 level teaspoon of
salt and

1

2

level teaspoon of baking soda to each quart of tepid water. Allow the

victim to sip slowly about 4 ounces (half a glass) over a period of 15 minutes.
Discontinue fluid intake if vomiting occurs. (Do not offer alcohol.)

Less Severe Burns (First and Second Degree)

3 Apply cool (not ice-cold) compresses using the cleanest available cloth article.
3 Do not break blisters, remove tissue, remove adhered particles of clothing, or

apply salve or ointment.

3 Apply clean, dry dressing if necessary.
3 Treat victim for shock as required.
3 Arrange for transportation to a hospital as quickly as possible.
3 If arms or legs are affected, keep them elevated.

© 1999 CRC Press LLC

panels and ac disconnect switches according to device rating. Breaker boxes should
contain a nameplate showing the manufacturer, rating, and other pertinent electrical
factors. The intent is to prevent devices from being subjected to excessive loads or volt-
ages.

Grounding

OSHA regulations describe two types of grounding: system grounding and equipment
grounding. System grounding actually connects one of the current-carrying conduc-
tors (such as the terminals of a supply transformer) to ground. (See

Figure 11.6

.)

Equipment grounding connects all of the noncurrent-carrying metal surfaces together
and to ground. From a grounding standpoint, the only difference between a grounded
electrical system and an ungrounded electrical system is that the main bonding
jumper from the service equipment ground to a current-carrying conductor is omitted
in the ungrounded system. The system ground performs two tasks:

·

It provides the final connection from equipment-grounding conductors to the
grounded circuit conductor, thus completing the ground-fault loop.

Table 11.4 Sixteen Common OSHA Violations (

After [1].)

Fact Sheet

Subject

NEC

1

Reference

1

Guarding of live parts

110-17

2

Identification

110-22

3

Uses allowed for flexible cord

400-7

4

Prohibited uses of flexible cord

400-8

5

Pull at joints and terminals must be prevented

400-10

6.1

Effective grounding, Part 1

250-51

6.2

Effective grounding, Part 2

250-51

7

Grounding of fixed equipment, general

250-42

8

Grounding of fixed equipment, specific

250-43

9

Grounding of equipment connected by cord
and plug

250-45

10

Methods of grounding, cord and
plug-connected equipment

250-59

11

AC circuits and systems to be grounded

250-5

12

Location of overcurrent devices

240-24

13

Splices in flexible cords

400-9

14

Electrical connections

110-14

15

Marking equipment

110-21

16

Working clearances about electric equipment

110-16

1

National Electrical Code

© 1999 CRC Press LLC

·

It solidly ties the electrical system and its enclosures to their surroundings (usu-
ally earth, structural steel, and plumbing). This prevents voltages at any source
from rising to harmfully high voltage-to-ground levels.

Figure 11.6 AC service entrance bonding requirements: (

a) 120 V phase-to-neutral

(240 V phase-to-phase), (

b) 3-phase 208 V wye (120 V phase-to-neutral), (c) 3-phase

240 V (or 480 V) delta. Note that the main bonding jumper is required in only two of the
designs.

(

a)

(

b)

(

c)

© 1999 CRC Press LLC

Note that equipment grounding--bonding all electric equipment to ground--is re-
quired whether or not the system is grounded. Equipment grounding serves two im-
portant tasks:

·

It bonds all surfaces together so that there can be no voltage difference among
them.

·

It provides a ground-fault current path from a fault location back to the electrical
source, so that if a fault current develops, it will rise to a level high enough to op-
erate the breaker or fuse.

The National Electrical Code (NEC) is complex and contains numerous require-

ments concerning electrical safety. The fact sheets listed in

Table 11.4

are available

from OSHA.

11.3.2 Beryllium Oxide Ceramics

Some tubes, both power grid and microwave, contain beryllium oxide (BeO) ceram-
ics, typically at the output waveguide window or around the cathode. Never perform
any operations on BeO ceramics that produce dust or fumes, such as grinding, grit
blasting, or acid cleaning. Beryllium oxide dust and fumes are highly toxic, and
breathing them can result in serious personal injury or death.

If a broken window is suspected on a microwave tube, carefully remove the device

from its waveguide, and seal the output flange of the tube with tape. Because BeO
warning labels may be obliterated or missing, maintenance personnel should contact
the tube manufacturer before performing any work on the device. Some tubes have
BeO internal to the vacuum envelope.

Take precautions to protect personnel working in the disposal or salvage of tubes

containing BeO. All such personnel should be made aware of the deadly hazards in-
volved and the necessity for great care and attention to safety precautions. Some tube
manufacturers will dispose of tubes without charge, provided they are returned to the
manufacturer prepaid, with a written request for disposal.

11.3.3 Corrosive and Poisonous Compounds

The external output waveguides and cathode high-voltage bushings of microwave
tubes are sometimes operated in systems that use a dielectric gas to impede micro-
wave or high-voltage breakdown. If breakdown does occur, the gas may decompose
and combine with impurities, such as air or water vapor, to form highly toxic and cor-
rosive compounds. Examples include Freon gas, which may form lethal phosgene,
and sulfur hexafluoride (SF

6

) gas, which may form highly toxic and corrosive sulfur

or fluorine compounds such as beryllium fluoride. When breakdown does occur in the
presence of these gases, proceed as follows:

·

Ventilate the area to outside air

·

Avoid breathing any fumes or touching any liquids that develop

© 1999 CRC Press LLC

·

Take precautions appropriate for beryllium compounds and for other highly toxic
and corrosive substances

If a coolant other than pure water is used, follow the precautions supplied by the cool-
ant manufacturer.

11.3.4 FC-75 Toxic Vapor

The decomposition products of FC-75 are highly toxic. Decomposition may occur as
a result of any of the following:

·

Exposure to temperatures above 200°C

·

Exposure to liquid fluorine or alkali metals (lithium, potassium, or sodium)

·

Exposure to ionizing radiation

Known thermal decomposition products include perfluoroisobutylene (PFIB; [CF

3

]

2

C = CF

2

), which is highly toxic in small concentrations.

If FC-75 has been exposed to temperatures above 200°C through fire, electric heat-

ing, or prolonged electric arcs, or has been exposed to alkali metals or strong ionizing
radiation, take the following steps:

·

Strictly avoid breathing any fumes or vapors.

·

Thoroughly ventilate the area.

·

Strictly avoid any contact with the FC-75.

Under such conditions, promptly replace the FC-75 and handle and dispose of the
contaminated FC-75 as a toxic waste.

11.3.5 Nonionizing Radiation

Nonionizing radio frequency radiation (RFR) resulting from high-intensity RF fields
is a growing concern to engineers who must work around high-power transmission
equipment. The principal medical concern regarding nonionizing radiation involves
heating of various body tissues, which can have serious effects, particularly if there is
no mechanism for heat removal. Recent research has also noted, in some cases, subtle
psychological and physiological changes at radiation levels below the threshold for
heat-induced biological effects. However, the consensus is that most effects are ther-
mal in nature.

High levels of RFR can affect one or more body systems or organs. Areas identified

as potentially sensitive include the ocular (eye) system, reproductive system, and the
immune system. Nonionizing radiation also is thought to be responsible for metabolic
effects on the central nervous system and cardiac system.

In spite of these studies, many of which are ongoing, there is still no clear evidence

in Western literature that exposure to medium-level nonionizing radiation results in
detrimental effects. Russian findings, on the other hand, suggest that occupational ex-

© 1999 CRC Press LLC

posure to RFR at power densities above 1.0 mW/cm

2

does result in symptoms, particu-

larly in the central nervous system.

Clearly, the jury is still out as to the ultimate biological effects of RFR. Until the situ-

ation is better defined, however, the assumption must be made that potentially serious
effects can result from excessive exposure. Compliance with existing standards should
be the minimum goal, to protect members of the public as well as facility employees.

NEPA Mandate

The National Environmental Policy Act of 1969 required the Federal Communica-
tions Commission to place controls on nonionizing radiation. The purpose was to pre-
vent possible harm to the public at large and to those who must work near sources of
the radiation. Action was delayed because no hard and fast evidence existed that low-
and medium-level RF energy is harmful to human life. Also, there was no evidence
showing that radio waves from radio and TV stations did not constitute a health haz-
ard.

During the delay, many studies were carried out in an attempt to identify those levels

of radiation that might be harmful. From the research, suggested limits were developed
by the American National Standards Institute (ANSI) and stated in the document
known as ANSI C95.1-1982. The protection criteria outlined in the standard are shown
in

Figure 11.7

.

The energy-level criteria were developed by representatives from a number of in-

dustries and educational institutions after performing research on the possible effects
of nonionizing radiation. The projects focused on absorption of RF energy by the hu-
man body, based upon simulated human body models. In preparing the document,
ANSI attempted to determine those levels of incident radiation that would cause the

Figure 11.7 The power density limits for nonionizing radiation exposure for humans.

© 1999 CRC Press LLC

body to absorb less than 0.4 W/kg of mass (averaged over the whole body) or peak ab-
sorption values of 8 W/kg over any 1 gram of body tissue.

From the data, the researchers found that energy would be absorbed more readily at

some frequencies than at others. The absorption rates were found to be functions of the
size of a specific individual and the frequency of the signal being evaluated. It was the
result of these absorption rates that culminated in the shape of the safe curve shown in
the figure. ANSI concluded that no harm would come to individuals exposed to radio
energy fields, as long as specific values were not exceeded when averaged over a period
of 0.1 hour. It was also concluded that higher values for a brief period would not pose
difficulties if the levels shown in the standard document were not exceeded when aver-
aged over the 0.1-hour time period.

The FCC adopted ANSI C95.1-1982 as a standard that would ensure adequate pro-

tection to the public and to industry personnel who are involved in working around RF
equipment and antenna structures.

Revised Guidelines

The ANSI C95.1-1982 standard was intended to be reviewed at 5-year intervals. Ac-
cordingly, the 1982 standard was due for reaffirmation or revision in 1987. The pro-
cess was indeed begun by ANSI, but was handed off to the Institute of Electrical and
Electronics Engineers (IEEE) for completion. In 1991, the revised document was
completed and submitted to ANSI for acceptance as ANSI/IEEE C95.1-1992.

The IEEE standard incorporated changes from the 1982 ANSI document in four

major areas:

·

An additional safety factor was provided in certain situations. The most signifi-
cant change was the introduction of new uncontrolled (public) exposure guide-
lines, generally established at one-fifth of the controlled (occupational) exposure
guidelines.

Figure 11.8

illustrates the concept for the microwave frequency band.

·

For the first time, guidelines were included for body currents; examination of the
electric and magnetic fields were determined to be insufficient to determine com-
pliance.

·

Minor adjustments were made to occupational guidelines, including relaxation of
the guidelines at certain frequencies and the introduction of breakpoints at new
frequencies.

·

Measurement procedures were changed in several aspects, most notably with re-
spect to spatial averaging and to minimum separation from reradiating objects
and structures at the site.

The revised guidelines are complex and beyond the scope of this handbook. Refer to
the ANSI/IEEE document for details.

© 1999 CRC Press LLC

Multiple-User Sites

At a multiple-user site, the responsibility for assessing the RFR situation--although
officially triggered by either a new user or the license renewal of all site tenants--is,
in reality, the joint responsibility of all the site tenants. In a multiple-user environment
involving various frequencies, and various protection criteria, compliance is indicated
when the fraction of the RFR limit within each pertinent frequency band is estab-
lished and added to the sum of all the other fractional contributions. The sum must not
be greater than 1.0. Evaluating the multiple-user environment is not a simple matter,
and corrective actions, if indicated, may be quite complex.

Operator Safety Considerations

RF energy must be contained properly by shielding and transmission lines. All input
and output RF connections, cables, flanges, and gaskets must be RF leakproof. The
following guidelines should be followed at all times:

·

Never operate a power tube without a properly matched RF energy absorbing
load attached.

·

Never look into or expose any part of the body to an antenna or open RF generat-
ing tube, circuit, or RF transmission system that is energized.

·

Monitor the RF system for radiation leakage at regular intervals and after servic-
ing.

Figure 11.8 ANSI/IEEE exposure guidelines for microwave frequencies.

© 1999 CRC Press LLC

11.3.6 X-Ray Radiation Hazard

The voltages typically used in microwave tubes are capable of producing dangerous X
rays. As voltages increase beyond 15 kV, metal-body tubes are capable of producing
progressively more dangerous radiation. Adequate X-ray shielding must be provided
on all sides of such tubes, particularly at the cathode and collector ends, as well as at
the modulator and pulse transformer tanks (as appropriate). High-voltage tubes never
should be operated without adequate X-ray shielding in place. The X-ray radiation of
the device should be checked at regular intervals and after servicing.

11.3.7 Implosion Hazard

Because of the high internal vacuum in power grid and microwave tubes, the glass or
ceramic output window or envelope can shatter inward (implode) if struck with suffi-
cient force or exposed to sufficient mechanical shock. Flying debris could result in
bodily injury, including cuts and puncture wounds. If the device is made of beryllium
oxide ceramic, implosion may produce highly toxic dust or fumes.

In the event of such an implosion, assume that toxic BeO ceramic is involved unless

confirmed otherwise.

11.3.8 Hot Coolant and Surfaces

Extreme heat occurs in the electron collector of a microwave tube and the anode of a
power grid tube during operation. Coolant channels used for water or vapor cooling
also can reach high temperatures (boiling--100°C--and above), and the coolant is
typically under pressure (as high as 100 psi). Some devices are cooled by boiling the
coolant to form steam.

Contact with hot portions of the tube or its cooling system can scald or burn. Care-

fully check that all fittings and connections are secure, and monitor back pressure for
changes in cooling system performance. If back pressure is increased above normal op-
erating values, shut the system down and clear the restriction.

For a device whose anode or collector is air-cooled, the external surface normally

operates at a temperature of 200 to 300°C. Other parts of the tube also may reach high
temperatures, particularly the cathode insulator and the cathode/heater surfaces. All
hot surfaces remain hot for an extended time after the tube is shut off. To prevent serious
burns, take care to avoid bodily contact with these surfaces during operation and for a
reasonable cool-down period afterward.

11.3.9 Polychlorinated Biphenyls

PCBs belong to a family of organic compounds known as chlorinated hydrocarbons.
Virtually all PCBs in existence today have been synthetically manufactured. PCBs
have a heavy oil-like consistency, high boiling point, a high degree of chemical stabil-
ity, low flammability, and low electrical conductivity. These characteristics resulted in
the widespread use of PCBs in high-voltage capacitors and transformers. Commercial

© 1999 CRC Press LLC

products containing PCBs were widely distributed between 1957 and 1977 under sev-
eral trade names including:

·

Aroclor

·

Pyroclor

·

Sanotherm

·

Pyranol

·

Askarel

Askarel is also a generic name used for nonflammable dielectric fluids containing
PCBs.

Table 11.5

lists some common trade names used for Askarel. These trade

names typically will be listed on the nameplate of a PCB transformer or capacitor.

PCBs are harmful because once they are released into the environment, they tend not

to break apart into other substances. Instead, PCBs persist, taking several decades to
slowly decompose. By remaining in the environment, they can be taken up and stored in
the fatty tissues of all organisms, from which they are slowly released into the blood-
stream. Therefore, because of the storage in fat, the concentration of PCBs in body tis-
sues can increase with time, even though PCB exposure levels may be quite low. This
process is called bioaccumulation. Furthermore, as PCBs accumulate in the tissues of
simple organisms, and as they are consumed by progressively higher organisms, the
concentration increases. This process is called biomagnification. These two factors are
especially significant because PCBs are harmful even at low levels. Specifically, PCBs
have been shown to cause chronic (long-term) toxic effects in some species of animals
and aquatic life. Well-documented tests on laboratory animals show that various levels
of PCBs can cause reproductive effects, gastric disorders, skin lesions, and cancerous
tumors.

PCBs may enter the body through the lungs, the gastrointestinal tract, and the skin.

After absorption, PCBs are circulated in the blood throughout the body and stored in

PCB Trade Names

Apirolio

Abestol

Askarel

1

Aroclor B

Chlorexto

Chlophen

Chlorinol

Clorphon

Diaclor

DK

Dykanol

EEC-18

Elemex

Eucarel

Fenclor

Hyvol

Inclor

Inerteen

Kanechlor

No-Flamol

Phenodlor

Pydraul

Pyralene

Pyranol

Pyroclor

Sal-T-Kuhl

Santothern FR Santovac

Solvol

Thermin

1

Generic name used for nonflammable dielectric fluids containing PCBs.

Table 11.5 Commonly Used Names for PCB Insulating Material

© 1999 CRC Press LLC

fatty tissues and a variety of organs, including the liver, kidneys, lungs, adrenal glands,
brain, heart, and skin.

The health risk from PCBs lies not only in the PCB itself, but also in the chemicals

that develop when PCBs are heated. Laboratory studies have confirmed that PCB
by-products, including polychlorinated dibenzofurans (PCDFs) and polychlorinated
dibenzo-p-dioxins (PCDDs), are formed when PCBs or chlorobenzenes are heated to
temperatures ranging from approximately 900 to 1300°F. Unfortunately, these prod-
ucts are more toxic than PCBs themselves.

Governmental Action

The U.S. Congress took action to control PCBs in October 1975 by passing the Toxic
Substances Control Act (TSCA). A section of this law specifically directed the EPA
to regulate PCBs. Three years later the Environmental Protection Agency (EPA) is-
sued regulations to implement the congressional ban on the manufacture, processing,
distribution, and disposal of PCBs. Since that time, several revisions and updates have
been issued by the EPA. One of these revisions, issued in 1982, specifically addressed
the type of equipment used in industrial plants and transmitting stations. Failure to
properly follow the rules regarding the use and disposal of PCBs has resulted in high
fines and even jail sentences.

Although PCBs are no longer being produced for electrical products in the United

States, there are thousands of PCB transformers and millions of small PCB capacitors
still in use or in storage. The threat of widespread contamination from PCB fire-related
incidents is one reason behind the EPA's efforts to reduce the number of PCB products
in the environment. The users of high-power equipment are affected by the regulations
primarily because of the widespread use of PCB transformers and capacitors. These
components usually are located in older (pre-1979) systems, so this is the first place to
look for them. However, some facilities also maintain their own primary power trans-
formers. Unless these transformers are of recent vintage, it is quite likely that they too
contain a PCB dielectric.

Table 11.6

lists the primary classifications of PCB devices.

PCB Components

The two most common PCB components are transformers and capacitors. A PCB
transformer is one containing at least 500 ppm (parts per million) PCBs in the dielec-
tric fluid. An Askarel transformer generally has 600,000 ppm or more. A PCB trans-
former may be converted to a PCB-contaminated device (50 to 500 ppm) or a
non-PCB device (less than 50 ppm) by having it drained, refilled, and tested. The test-
ing must not take place until the transformer has been in service for a minimum of 90
days. Note that this is not something a maintenance technician can do. It is the exclu-
sive domain of specialized remanufacturing companies.

PCB transformers must be inspected quarterly for leaks. If an impervious dike is

built around the transformer sufficient to contain all of the liquid material, the inspec-
tions can be conducted yearly. Similarly, if the transformer is tested and found to con-

© 1999 CRC Press LLC

Table 11.6 Definition of PCB Terms as Identified by the EPA

Term

Definition

Examples

PCB

Any chemical substance that
is limited to the biphenyl mole-
cule that has been chlorinated
to varying degrees, or any
combination of substances
that contain such substances.

PCB dielectric fluids, PCB
heat-transfer fluids, PCB
hydraulic fluids,
2,2',4-trichlorobiphenyl

PCB article

Any manufactured article,
other than a PCB container,
that contains PCBs and
whose surface has been in
direct contact with PCBs.

Capacitors, transformers,
electric motors, pumps,
pipes

PCB container

A device used to contain
PCBs or PCB articles, and
whose surface has been in
direct contact with PCBs.

Packages, cans, bottles,
bags, barrels, drums,
tanks

PCB article container

A device used to contain PCB
articles or equipment, and
whose surface has not been in
direct contact with PCBs.

Packages, cans, bottles,
bags, barrels, drums,
tanks

PCB equipment

Any manufactured item, other
than a PCB container or PCB
article container, that contains
a PCB article or other PCB
equipment.

Microwave systems,
fluorescent light ballasts,
electronic equipment

PCB item

Any PCB article, PCB article
container, PCB container, or
PCB equipment that
deliberately or unintentionally
contains, or has as a part of it,
any PCBs.

PCB transformer

Any transformer that contains
PCBs in concentrations of 500
ppm or greater.

PCB contaminated

Any electric equipment that
contains more than 50, but
less than 500 ppm of PCBs.
(Oil-filled electric equipment
other than circuit breakers,
reclosers, and cable whose
PCB concentration is un-
known must be assumed to be
PCB-contaminated electric
equipment.)

Transformers, capacitors,
contaminated circuit break-
ers, reclosers, voltage reg-
ulators, switches, cable,
electromagnets

© 1999 CRC Press LLC

tain less than 60,000 ppm, a yearly inspection is sufficient. Failed PCB transformers
cannot be repaired; they must be properly disposed of.

If a leak develops, it must be contained and daily inspections begun. A cleanup must

be initiated as soon as possible, but no later than 48 hours after the leak is discovered.
Adequate records must be kept of all inspections, leaks, and actions taken for 3 years af-
ter disposal of the component. Combustible materials must be kept a minimum of 5 m
from a PCB transformer and its enclosure.

As of October 1, 1990, the use of PCB transformers (500 ppm or greater) was pro-

hibited in or near commercial buildings when the secondary voltages are 480 V ac or
higher.

The EPA regulations also require that the operator notify others of the possible dan-

gers. All PCB transformers (including PCB transformers in storage for reuse) must be
registered with the local fire department. The following information must be supplied:

·

The location of the PCB transformer(s).

·

Address(es) of the building(s) and, for outdoor PCB transformers, the location.

·

Principal constituent of the dielectric fluid in the transformer(s).

·

Name and telephone number of the contact person in the event of a fire involving
the equipment.

Any PCB transformers used in a commercial building must be registered with the
building owner. All building owners within 30 m of such PCB transformers also must
be notified. In the event of a fire-related incident involving the release of PCBs, the
Coast Guard National Spill Response Center (800-424-8802) must be notified imme-
diately. Appropriate measures also must be taken to contain and control any possible
PCB release into water.

Capacitors are divided into two size classes, large and small. A PCB small capacitor

contains less than 1.36 kg (3 lbs) of dielectric fluid. A capacitor having less than 100 in

3

also is considered to contain less than 3 lb of dielectric fluid. A PCB large capacitor has
a volume of more than 200 in

3

and is considered to contain more than 3 lb of dielectric

fluid. Any capacitor having a volume between 100 and 200 in

3

is considered to contain

3 lb of dielectric, provided the total weight is less than 9 lb. A PCB large high-voltage
capacitor contains 3 lb or more of dielectric fluid and operates at voltages of 2 kV or
greater. A large low-voltage capacitor also contains 3 lb or more of dielectric fluid but
operates below 2 kV.

The use and servicing of PCB small capacitors is not restricted by the EPA unless

there is a leak. In that event, the leak must be repaired or the capacitor disposed of. Dis-
posal may be handled by an approved incineration facility, or the component may be
placed in a specified container and buried in an approved chemical waste landfill. Items
such as capacitors that are leaking oil greater than 500 ppm PCBs should be taken to an
EPA-approved PCB disposal facility.

© 1999 CRC Press LLC

PCB Liability Management

Properly managing the PCB risk is not particularly difficult; the keys are understand-
ing the regulations and following them carefully. Any program should include the fol-
lowing steps:

·

Locate and identify all PCB devices. Check all stored or spare devices.

·

Properly label PCB transformers and capacitors according to EPA requirements.

·

Perform the required inspections and maintain an accurate log of PCB items, their
location, inspection results, and actions taken. These records must be maintained
for 3 years after disposal of the PCB component.

·

Complete the annual report of PCBs and PCB items by July 1 of each year. This
report must be retained for 5 years.

·

Arrange for necessary disposal through a company licensed to handle PCBs. If
there are any doubts about the company's license, contact the EPA.

·

Report the location of all PCB transformers to the local fire department and to the
owners of any nearby buildings.

The importance of following the EPA regulations cannot be overstated.

11.4 References

1. National Electrical Code, NFPA #70.

11.5 Bibliography

"Current Intelligence Bulletin #45," National Institute for Occupational Safety and

Health, Division of Standards Development and Technology Transfer, February 24,
1986.

Code of Federal Regulations, 40, Part 761.
"Electrical Standards Reference Manual," U.S. Department of Labor, Washington,

DC.

Hammett, William F., "Meeting IEEE C95.1-1991 Requirements," NAB 1993 Broad-

cast Engineering Conference Proceedings, National Association of Broadcasters,
Washington, D.C., pp. 471 - 476, April 1993.

Hammar, Willie, Occupational Safety Management and Engineering, Prentice Hall,

New York.

Markley, Donald, "Complying with RF Emission Standards," Broadcast Engineering,

Intertec Publishing, Overland Park, KS, May 1986.

"Occupational Injuries and Illnesses in the United States by Industry," OSHA Bulletin

2278, U.S. Department of Labor, Washington, DC, 1985.

OSHA, "Electrical Hazard Fact Sheets," U.S. Department of Labor, Washington, DC,

January 1987.

OSHA, "Handbook for Small Business," U.S. Department of Labor, Washington, DC.

© 1999 CRC Press LLC

Pfrimmer, Jack, "Identifying and Managing PCBs in Broadcast Facilities," 1987 NAB

Engineering Conference Proceedings, National Association of Broadcasters,
Washington, DC, 1987.

"Safety Precautions," Publication no. 3386A, Varian Associates, Palo Alto, CA,

March 1985.

Smith, Milford K., Jr., "RF Radiation Compliance," Proceedings of the Broadcast En-

gineering Conference, Society of Broadcast Engineers, Indianapolis, IN, 1989.

"Toxics Information Series," Office of Toxic Substances, July 1983.
Whitaker, Jerry C., AC Power Systems, 2nd Ed., CRC Press, Boca Raton, FL, 1998.
Whitaker, Jerry C., G. DeSantis, and C. Paulson, Interconnecting Electronic Systems,

CRC Press, Boca Raton, FL, 1993.

Whitaker, Jerry C., Maintaining Electronic Systems, CRC Press, Boca Raton, FL,

1991.

Whitaker, Jerry C., Radio Frequency Transmission Systems: Design and Operation,

McGraw-Hill, New York, 1990.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

12 Reference Data"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

© 1999 CRC Press LLC

Chapter

12

Reference Data

Standard Units

Name

Symbol

Quantity

ampere

A

electric current

ampere per meter

A/m

magnetic field strength

ampere per square meter

A/m

2

current density

Becquerel

Bq

activity (of a radionuclide)

candela

cd

luminous intensity

coulomb

C

electric charge

coulomb per kilogram

C/kg

exposure (x- and gamma-rays)

coulomb per sq. meter

C/m

2

electric flux density

cubic meter

m

3

volume

cubic meter per kilogram

m

3

/kg

specific volume

degree Celsius

°C

Celsius temperature

farad

F

capacitance

farad per meter

F/m

permittivity

henry

H

inductance

henry per meter

H/m

permeability

hertz

Hz

frequency

joule

J

energy, work, quantity of heat

joule per cubic meter

J/m

3

energy density

joule per kelvin

J/K

heat capacity

joule per kilogram K

J/(kg-K)

specific heat capacity

joule per mole

J/mol

molar energy

kelvin

K

thermodynamic temperature

kilogram

kg

mass

kilogram per cubic meter

kg/m

3

density, mass density

lumen

lm

luminous flux

lux

lx

luminance

meter

m

length

meter per second

m/s

speed, velocity

meter per second sq.

m/s

2

acceleration

(

Continued on next page)

© 1999 CRC Press LLC

Name

Symbol

Quantity

mole

mol

amount of substance

pascal

Pa

pressure, stress

pascal second

Pa

·s

dynamic viscosity

radian

rad

plane angle

radian per second

rad/s

angular velocity

radian per second squared

rad/s

2

angular acceleration

second

s

time

siemens

S

electric conductance

square meter

m

2

area

steradian

sr

solid angle

tesla

T

magnetic flux density

volt

V

electrical potential

volt per meter

V/m

electric field strength

watt

W

power, radiant flux

watt per meter kelvin

W/(m

·K)

thermal conductivity

watt per square meter

W/m

2

heat (power) flux density

weber

Wb

magnetic flux

Standard Units Continued

Standard Prefixes

Multiple

Prefix

Symbol

10

18

exa

E

10

15

peta

P

10

12

tera

T

10

9

giga

G

10

6

mega

M

10

3

kilo

k

10

2

hecto

h

10

deka

da

10

-1

deci

d

10

-2

centi

c

10

-3

milli

m

10

-6

micro

µ

10

-9

nano

n

10

-12

pico

p

10

-15

femto

f

10

-18

atto

a

© 1999 CRC Press LLC

Common Standard Units for Electronics Work

Unit

Symbol

centimeter

cm

cubic centimeter

cm

3

cubic meter per second

m

3

/s

gigahertz

GHz

gram

g

kilohertz

kHz

kilohm

k

kilojoule

kJ

kilometer

km

kilovolt

kV

kilovoltampere

kVA

kilowatt

kW

megahertz

MHz

megavolt

MV

megawatt

MW

megohm

M

microampere

µA

microfarad

µF

microgram

µg

microhenry

µH

microsecond

µs

microwatt

µW

milliampere

mA

milligram

mg

millihenry

mH

millimeter

mm

millisecond

ms

millivolt

mV

milliwatt

mW

nanoampere

nA

nanofarad

nF

nanometer

nm

nanosecond

ns

nanowatt

nW

picoampere

pA

picofarad

pF

picosecond

ps

picowatt

pW

© 1999 CRC Press LLC

Power Conversion Factors (Decibels to Watts)

dBm

dBw

Watts

Multiple

Prefix

+150

+120

1,000,000,000,000

10

12

1 terawatt

+140

+110

100,000,000,000

10

11

100 gigawatts

+130

+100

10,000,000,000

10

10

10 gigawatts

+120

+90

1,000,000,000

10

9

1 gigawatt

+110

+80

100,000,000

10

8

100 megawatts

+100

+70

10,000,000

10

7

10 megawatts

+90

+60

1,000,000

10

6

1 megawatt

+80

+50

100,000

10

5

100 kilowatts

+70

+40

10,000

10

4

10 kilowatts

+60

+30

1000

10

3

1 kilowatt

+50

+20

100

10

2

1 hectowatt

+40

+10

10

10

1 decawatt

+30

0

1

1

1 watt

+20

10

0.1

10

1

1 deciwatt

+10

20

0.01

10

2

1 centiwatt

0

30

0.001

10

3

1 milliwatt

10

40

0.0001

10

4

100 microwatts

20

50

0.00001

10

5

10 microwatts

30

60

0.000,001

10

6

1 microwatt

40

70

0.0,000,001

10

7

100 nanowatts

50

80

0.00,000,001

10

8

10 nanowatts

60

90

0.000,000,001

10

9

1 nanowatt

70

100

0.0,000,000,001

10

10

100 picowatts

80

110

0.00,000,000,001

10

11

10 picowatts

90

120

0.000,000,000,001

10

12

1 picowatt

© 1999 CRC Press LLC

VSWR

Reflection

Return

Power

Percent

Coefficient

Loss

Ratio

Reflected

1.01:1

0.0050

46.1 dB

0.00002

0.002

1.02:1

0.0099

40.1 dB

0.00010

0.010

1.04:1

0.0196

34.2 dB

0.00038

0.038

1.06:1

0.0291

30.7 dB

0.00085

0.085

1.08:1

0.0385

28.3 dB

0.00148

0.148

1.10:1

0.0476

26.4 dB

0.00227

0.227

1.20:1

0.0909

20.8 dB

0.00826

0.826

1.30:1

0.1304

17.7 dB

0.01701

1.7

1.40:1

0.1667

15.6 dB

0.02778

2.8

1.50:1

0.2000

14.0 dB

0.04000

4.0

1.60:1

0.2308

12.7 dB

0.05325

5.3

1.70:1

0.2593

11.7 dB

0.06722

6.7

1.80:1

0.2857

10.9 dB

0.08163

8.2

1.90:1

0.3103

10.2 dB

0.09631

9.6

2.00:1

0.3333

9.5 dB

0.11111

11.1

2.20:1

0.3750

8.5 dB

0.14063

14.1

2.40:1

0.4118

7.7 dB

0.16955

17.0

2.60:1

0.4444

7.0 dB

0.19753

19.8

2.80:1

0.4737

6.5 dB

0.22438

22.4

3.00:1

0.5000

6.0 dB

0.25000

25.0

3.50:1

0.5556

5.1 dB

0.30864

30.9

4.00:1

0.6000

4.4 dB

0.36000

36.0

4.50:1

0.6364

3.9 dB

0.40496

40.5

5.00:1

0.6667

3.5 dB

0.44444

44.4

6.00:1

0.7143

2.9 dB

0.51020

51.0

7.00:1

0.7500

2.5 dB

0.56250

56.3

8.00:1

0.7778

2.2 dB

0.60494

60.5

9.00:1

0.8000

1.9 dB

0.64000

64.0

10.00:1

0.8182

1.7 dB

0.66942

66.9

15.00:1

0.8750

1.2 dB

0.76563

76.6

20.00:1

0.9048

0.9 dB

0.81859

81.9

30.00:1

0.9355

0.6 dB

0.87513

97.5

40.00:1

0.9512

0.4 dB

0.90482

90.5

50.00:1

0.9608

0.3 dB

0.92311

92.3

Voltage Standing Wave Ratio and Key Parameters

© 1999 CRC Press LLC

Wire

Diam.

Circular Turns Per Linear Inch

1

Ohms

Current-

Diam.

Size

in

mil

Per

Carrying

in

AWG

mils

Area

enam. SCE

DCC

100 ft

2

Capacity

3

mm

1

289.3

83810

-

-

-

0.1239

119.6

7.348

2

257.6

05370

-

-

-

0.1563

94.8

6.544

3

229.4

62640

-

-

-

0.1970

75.2

5.827

4

204.3

41740

-

-

-

0.2485

59.6

5.189

5

181.9

33100

-

-

-

0.3133

47.3

4.621

6

162.0

26250

-

-

-

0.3951

37.5

4.115

7

144.3

20820

-

-

-

0.4982

29.7

3.665

8

128.5

16510

7.6

-

7.1

0.6282

23.6

3.264

9

114.4

13090

8.6

-

7.8

0.7921

18.7

2.906

10

101.9

10380

9.6

9.1

8.9

0.9989

14.8

2.588

11

90.7

8234

10.7

-

9.8

1.26

11.8

2.305

12

80.8

6530

12.0

11.3

10.9

1.588

9.33

2.063

13

72.0

5178

13.5

-

12.8

2.003

7.40

1.828

14

64.1

4107

15.0

14.0

13.8

2.525

5.87

1.628

15

57.1

3257

16.8

-

14.7

3.184

4.65

1.450

16

50.8

2583

18.9

17.3

16.4

4.016

3.69

1.291

17

45.3

2048

21.2

-

18.1

5.064

2.93

1.150

18

40.3

1624

23.6

21.2

19.8

6.386

2.32

1.024

19

35.9

1288

26.4

-

21.8

8.051

1.84

0.912

20

32.0

1022

29.4

25.8

23.8

10.15

1.46

0.812

21

28.5

810

33.1

-

26.0

12.8

1.16

0.723

22

25.3

642

37.0

31.3

30.0

16.14

0.918

0.644

23

22.6

510

41.3

-

37.6

20.36

0.728

0.573

24

20.1

404

46.3

37.6

35.6

25.67

0.577

0.511

25

17.9

320

51.7

-

38.6

32.37

0.458

0.455

26

15.9

254

58.0

46.1

41.8

40.81

0.363

0.406

27

14.2

202

64.9

-

45.0

51.47

0.288

0.361

28

12.6

160

72.7

54.6

48.5

64.9

0.228

0.321

29

11.3

127

81.6

-

51.8

81.83

0.181

0.286

30

10.0

101

90.5

64.1

55.5

103.2

0.144

0.255

31

8.9

50

101

-

59.2

130.1

0.114

0.227

32

8.0

63

113

74.1

61.6

164.1

0.090

0.202

33

7.1

50

127

-

66.3

206.9

0.072

0.180

34

6.3

40

143

86.2

70.0

260.9

0.057

0.160

35

5.6

32

158

-

73.5

329.0

0.045

0.143

36

5.0

25

175

103.1

77.0

414.8

0.036

0.127

37

4.5

20

198

-

80.3

523.1

0.028

0.113

38

4.0

16

224

116.3

83.6

659.6

0.022

0.101

39

3.5

12

248

-

86.6

831.8

0.018

0.090

1. Based on 25.4 mm.
2. Ohms per 1000 ft measured at 20°C.
3. Current-carrying capacity at 700 circular mils per amp.

Specifications of Standard Copper Wire

© 1999 CRC Press LLC

°Celsius

°Fahrenheit

°Celsius

°Fahrenheit

50

58

125

257

45

49

130

266

40

40

135

275

35

31

140

284

30

22

145

293

25

13

150

302

20

4

155

311

15

5

160

320

10

14

165

329

5

23

170

338

0

32

175

347

5

41

180

356

10

50

185

365

15

59

190

374

20

68

195

383

25

77

200

392

30

86

205

401

35

95

210

410

40

104

215

419

45

113

220

428

50

122

225

437

55

131

230

446

60

140

235

455

65

149

240

464

70

158

245

473

75

167

250

482

80

176

255

491

85

185

260

500

90

194

265

509

95

203

270

518

100

212

275

527

105

221

280

536

110

230

285

545

115

239

290

554

120

248

295

563

Celsius-to-Fahrenheit Conversion

© 1999 CRC Press LLC

Inch 0

1/8

1/4

3/8

1/2

5/8

3/4

7/8

Inch

0

0.0

3.18

6.35

9.52

12.70

15.88

19.05

22.22

0

1

25.40

28.58

31.75

34.92

38.10

41.28

44.45

47.62

1

2

50.80

53.98

57.15

60.32

63.50

66.68

69.85

73.02

2

3

76.20

79.38

82.55

85.72

88.90

92.08

95.25

98.42

3

4

101.6

104.8

108.0

111.1

114.3

117.5

120.6

123.8

4

5

127.0

130.2

133.4

136.5

139.7

142.9

146.0

149.2

5

6

152.4

155.6

158.8

161.9

165.1

168.3

171.4

174.6

6

7

177.8

181.0

184.2

187.3

190.5

193.7

196.8

200.0

7

8

203.2

206.4

209.6

212.7

215.9

219.1

222.2

225.4

8

9

228.6

231.8

235.0

238.1

241.3

244.5

247.6

250.8

9

10

254.0

257.2

260.4

263.5

266.7

269.9

273.0

276.2

10

11

279

283

286

289

292

295

298

302

11

12

305

308

311

314

317

321

324

327

12

13

330

333

337

340

343

346

349

352

13

14

356

359

362

365

368

371

375

378

14

15

381

384

387

391

394

397

400

403

15

16

406

410

413

416

419

422

425

429

16

17

432

435

438

441

445

448

451

454

17

18

457

460

464

467

470

473

476

479

18

19

483

486

489

492

495

498

502

505

19

20

508

511

514

518

521

524

527

530

20

Inch-to-Millimeter Conversion

© 1999 CRC Press LLC

mm

Inches

mm

Inches

mm

Inches

1

0.039370

46

1.811020

91

3.582670

2

0.078740

47

1.850390

92

3.622040

3

0.118110

48

1.889760

93

3.661410

4

0.157480

49

1.929130

94

3.700780

5

0.196850

50

1.968500

95

3.740150

6

0.236220

51

2.007870

96

3.779520

7

0.275590

52

2.047240

97

3.818890

8

0.314960

53

2.086610

98

3.858260

9

0.354330

54

2.125980

99

3.897630

10

0.393700

55

2.165350

100

3.937000

11

0.433070

56

2.204720

105

4.133848

12

0.472440

57

2.244090

110

4.330700

13

0.511810

58

2.283460

115

4.527550

14

0.551180

59

2.322830

120

4.724400

15

0.590550

60

2.362200

125

4.921250

16

0.629920

61

2.401570

210

8.267700

17

0.669290

62

2.440940

220

8.661400

18

0.708660

63

2.480310

230

9.055100

19

0.748030

64

2.519680

240

9.448800

20

0.787400

65

2.559050

250

9.842500

21

0.826770

66

2.598420

260

10.236200

22

0.866140

67

2.637790

270

10.629900

23

0.905510

68

2.677160

280

11.032600

24

0.944880

69

2.716530

290

11.417300

25

0.984250

70

2.755900

300

11.811000

26

1.023620

71

2.795270

310

12.204700

27

1.062990

72

2.834640

320

12.598400

28

1.102360

73

2.874010

330

12.992100

29

1.141730

74

2.913380

340

13.385800

30

1.181100

75

2.952750

350

13.779500

31

1.220470

76

2.992120

360

14.173200

32

1.259840

77

3.031490

370

14.566900

33

1.299210

78

3.070860

380

14.960600

34

1.338580

79

3.110230

390

15.354300

35

1.377949

80

3.149600

400

15.748000

36

1.417319

81

3.188970

500

19.685000

37

1.456689

82

3.228340

600

23.622000

38

1.496050

83

3.267710

700

27.559000

39

1.535430

84

3.307080

800

31.496000

40

1.574800

85

3.346450

900

35.433000

41

1.614170

86

3.385820

1000

39.370000

42

1.653540

87

3.425190

2000

78.740000

43

1.692910

88

3.464560

3000

118.110000

44

1.732280

89

3.503903

4000

157.480000

45

1.771650

90

3.543300

5000

196.850000

Millimeters-to-Decimal Inches Conversion

© 1999 CRC Press LLC

Common

Decimal

mm

Common

Decimal

mm

Fractions

1

Fractions

1

(approx.)

Fractions

Fractions

(appox.)

1/128

0.008

0.20

1/2

0.500

12.70

1/64

0.016

0.40

33/64

0.516

13.10

1/32

0.031

0.79

17/32

0.531

13.49

3/64

0.047

1.19

35/64

0.547

13.89

1/16

0.063

1.59

9/16

0.563

14.29

5/64

0.078

1.98

37/64

0.578

14.68

3/32

0.094

2.38

19/32

0.594

15.08

7/64

0.109

2.78

39/64

0.609

15.48

1/8

0.125

3.18

5/8

0.625

15.88

9/64

0.141

3.57

41/64

0.641

16.27

5/32

0.156

3.97

21/32

0.656

16.67

11/64

0.172

4.37

43/64

0.672

17.07

3/16

0.188

4.76

11/16

0.688

17.46

13/64

0.203

5.16

45/64

0.703

17.86

7/32

0.219

5.56

23/32

0.719

18.26

15/64

0.234

5.95

47/64

0.734

18.65

1/4

0.250

6.35

3/4

0.750

19.05

17/64

0.266

6.75

49/64

0.766

19.45

9/32

0.281

7.14

25/32

0.781

19.84

19/64

0.297

7.54

51/64

0.797

20.24

5/16

0.313

7.94

13/16

0.813

20.64

21/64

0.328

8.33

53/64

0.828

21.03

11/32

0.344

8.73

27/32

0.844

21.43

23/64

0.359

9.13

55/64

0.859

21.83

3/8

0.375

9.53

7/8

0.875

22.23

25/64

0.391

9.92

57/64

0.891

22.62

13/32

0.406

10.32

29/32

0.906

23.02

27/64

0.422

10.72

59/64

0.922

23.42

7/16

0.438

11.11

15/16

0.938

23.81

29/64

0.453

11.51

61/64

0.953

24.21

15/32

0.469

11.91

31/32

0.969

24.61

31/64

0.484

12.30

63/64

0.984

25.00

1

Units in inches.

Conversion of Common Fractions to Decimal and Millimeter Units

© 1999 CRC Press LLC

Known Quantity

Multiply By

Quantity to Find

inches (in)

2.54

centimeters (cm)

feet (ft)

30

centimeters (cm)

yards (yd)

0.9

meters (m)

miles (mi)

1.6

kilometers (km)

millimeters (mm)

0.04

inches (in)

centimeters (cm)

0.4

inches (in)

meters (m)

3.3

feet (ft)

meters (m)

1.1

yards (yd)

kilometers (km)

0.6

miles (mi)

centimeters (cm)

10

millimeters (mm)

decimeters (dm)

10

centimeters (cm)

decimeters (dm)

100

millimeters (mm)

meters (m)

10

decimeters (dm)

meters (m)

1000

millimeters (mm)

dekameters (dam)

10

meters (m)

hectometers (hm)

10

dekameters (dam)

hectometers (hm)

100

meters (m)

kilometers (km)

10

hectometers (hm)

kilometers (km)

1000

meters (m)

Conversion Ratios for Length

Known Quantity

Multiply By

Quantity to Find

square inches (in

2

)

6.5

square centimeters (cm

2

)

square feet (ft

2

)

0.09

square meters (m

2

)

square yards (yd

2

)

0.8

square meters (m

2

)

square miles (mi

2

)

2.6

square kilometers (km

2

)

acres

0.4

hectares (ha)

square centimeters (cm

2

)

0.16

square inches (in

2

)

square meters (m

2

)

1.2

square yards (yd

2

)

square kilometers (km

2

)

0.4

square miles (mi

2

)

hectares (ha)

2.5

acres

square centimeters (cm

2

)

100

square millimeters (mm

2

)

square meters (m

2

)

10,000

square centimeters (cm

2

)

square meters (m

2

)

1,000,000

square millimeters (mm

2

)

ares (a)

100

square meters (m

2

)

hectares (ha)

100

ares (a)

hectares (ha)

10,000

square meters (m

2

)

square kilometers (km

2

)

100

hectares (ha)

square kilometers (km

2

)

1,000

square meters (m

2

)

Conversion Ratios for Area

© 1999 CRC Press LLC

Known Quantity

Multiply By

Quantity to Find

ounces (oz)

28

grams (g)

pounds (lb)

0.45

kilograms (kg)

tons

0.9

tonnes (t)

grams (g)

0.035

ounces (oz)

kilograms (kg)

2.2

pounds (lb)

tonnes (t)

100

kilograms (kg)

tonnes (t)

1.1

tons

centigrams (cg)

10

milligrams (mg)

decigrams (dg)

10

centigrams (cg)

decigrams (dg)

100

milligrams (mg)

grams (g)

10

decigrams (dg)

grams (g)

1000

milligrams (mg)

dekagram (dag)

10

grams (g)

hectogram (hg)

10

dekagrams (dag)

hectogram (hg)

100

grams (g)

kilograms (kg)

10

hectograms (hg)

kilograms (kg)

1000

grams (g)

metric tons (t)

1000

kilograms (kg)

Conversion Ratios for Mass

Known Quantity

Multiply By

Quantity to Find

milliliters (mL)

0.03

fluid ounces (fl oz)

liters (L)

2.1

pints (pt)

liters (L)

1.06

quarts (qt)

liters (L)

0.26

gallons (gal)

gallons (gal)

3.8

liters (L)

quarts (qt)

0.95

liters (L)

pints (pt)

0.47

liters (L)

cups (c)

0.24

liters (L)

fluid ounces (fl oz)

30

milliliters (mL)

teaspoons (tsp)

5

milliliters (mL)

tablespoons (tbsp)

15

milliliters (mL)

liters (L)

100

milliliters (mL)

Conversion Ratios for Volume

© 1999 CRC Press LLC

Known Quantity

Multiply By

Quantity to Find

cubic meters (m

3

)

35

cubic feet (ft

3

)

cubic meters (m

3

)

1.3

cubic yards (yd

3

)

cubic yards (yd

3

)

0.76

cubic meters (m

3

)

cubic feet (ft

3

)

0.028

cubic meters (m

3

)

cubic centimeters (cm

3

)

1000

cubic millimeters (mm

3

)

cubic decimeters (dm

3

)

1000

cubic centimeters (cm

3

)

cubic decimeters (dm

3

)

1,000,000

cubic millimeters (mm

3

)

cubic meters (m

3

)

1000

cubic decimeters (dm

3

)

cubic meters (m

3

)

1

steres

cubic feet (ft

3

)

1728

cubic inches (in

3

)

cubic feet (ft

3

)

28.32

liters (L)

cubic inches (in

3

)

16.39

cubic centimeters (cm

3

)

cubic meters (m

3

)

264

gallons (gal)

cubic yards (yd

3

)

27

cubic feet (ft

3

)

cubic yards (yd

3

)

202

gallons (gal)

gallons (gal)

231

cubic inches (in

3

)

Conversion Ratios for Cubic Measure

Known Quantity

Multiply By

Quantity to Find

Btu per minute

0.024

horsepower (hp)

Btu per minute

17.57

watts (W)

horsepower (hp)

33,000

foot-pounds per min (ft-lb/min)

horsepower (hp)

746

watts (W)

kilowatts (kW)

57

Btu per minute

kilowatts (kW)

1.34

horsepower (hp)

Conversion Ratios for Electrical Quantities

© 1999 CRC Press LLC

Basic Electrical Units

Unit

Symbol

Definition

Ampere

A

One ampere is the constant current flowing in two parallel
conductors one meter apart that would produce a force of
2

× 10

7

newtons per meter of length.

Coulomb

C

One coulomb is the charge that is moved in one second by a
current of one ampere.

Farad

F

One farad is the capacitance of a capacitor that produces a
potential of one volt between the plates when charged to one
coulomb.

Henry

H

One henry is the inductance of a coil that has one volt in-
duced in it when the current varies uniformly at one ampere
per second.

Joule

J

One joule is the work done by a force of one newton acting
over a distance of one meter.

Ohm

One ohm is the resistance between two points of a conductor
that produces a current of one ampere when one volt is ap-
plied between these points.

Siemens

S

One siemens is the conductance between two points of a
conductor that produces a current of one ampere when one
volt is applied between these points. Reciprocal of the ohm.
(Formerly mho).

Volt

V

One volt is the potential difference between two points in a
wire carrying one ampere when the power dissipated be-
tween these points is one watt.

Watt

W

One watt is the power that produces energy at one joule per
second.

Weber

Wb

One weber is the magnetic flux that produces an electromo-
tive force of one volt in a circuit of one turn as the flux
changes uniformly from maximum to zero in one second.

© 1999 CRC Press LLC

Whitaker, Jerry C. "Chapter

13 Glossary"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

Chapter

13

Glossary

A

ab so lute de lay The amount of time a sig nal is de layed. The de lay may be ex pressed in

time or num ber of pulse events.

ab so lute zero The low est tem per a ture the o ret i cally pos si ble, 273.16°C. Ab so lute

zero is equal to zero de grees Kel vin.

ab sorp tion The trans fer ence of some or all of the en ergy con tained in an elec tro mag -

netic wave to the sub stance or me dium in which it is prop a gat ing or upon which it is
in ci dent.

ab sorp tion auroral The loss of en ergy in a ra dio wave pass ing through an area af -

fected by so lar auroral ac tiv ity.

ac cou pling A method of cou pling one cir cuit to an other through a ca pac i tor or trans -

former so as to trans mit the vary ing (ac) char ac ter is tics of the sig nal while block ing
the static (dc) char ac ter is tics.

ac/dc cou pling Cou pling be tween cir cuits that ac com mo dates the pass ing of both ac

and dc sig nals (may also be re ferred to as sim ply dc cou pling).

ac cel er ated life test A spe cial form of re li abil ity test ing per formed by an equip ment

man u fac turer. The unit un der test is sub jected to stresses that ex ceed those typ i cally
ex pe ri enced in nor mal op er a tion. The goal of an

ac cel er ated life test is to im prove

the re li abil ity of prod ucts shipped by forc ing la tent fail ures in com po nents to be -
come ev i dent be fore the unit leaves the fac tory.

ac cel er at ing elec trode The elec trode that causes elec trons emit ted from an elec tron

gun to ac cel er ate in their jour ney to the screen of a cath ode ray tube.

ac cel er at ing volt age The volt age ap plied to an elec trode that ac cel er ates a beam of

elec trons or other charged par ti cles.

ac cept able re li abil ity level The max i mum num ber of fail ures al lowed per thou sand

op er at ing hours of a given com po nent or sys tem.

ac cep tance test The pro cess of test ing newly pur chased equip ment to en sure that it is

fully com pli ant with con trac tual spec i fi ca tions.

ac cess The point at which en try is gained to a cir cuit or fa cil ity.

ac qui si tion time In a com mu ni ca tion sys tem, the amount of time re quired to at tain

syn chro nism.

ac tive Any de vice or cir cuit that in tro duces gain or uses a source of en ergy other than

that in her ent in the sig nal to per form its func tion.

adapter A fit ting or elec tri cal con nec tor that links equip ment that can not be con nected

di rectly.

adap tive A de vice able to ad just or re act to a con di tion or ap pli ca tion, as an adap tive

cir cuit. This term usu ally re fers to fil ter cir cuits.

adap tive sys tem A gen eral name for a sys tem that is ca pa ble of reconfiguring it self to

meet new re quire ments.

ad der A de vice whose out put rep re sents the sum of its in puts.

ad ja cent chan nel in ter fer ence In ter fer ence to com mu ni ca tions caused by a trans mit -

ter op er at ing on an ad ja cent ra dio chan nel. The side bands of the trans mit ter mix
with the car rier be ing re ceived on the de sired chan nel, re sult ing in noise.

ad mit tance A mea sure of how well al ter nat ing cur rent flows in a con duc tor. It is the re -

cip ro cal of

im ped ance and is ex pressed in siemens. The real part of ad mit tance is

con duc tance; the imag i nary part is susceptance.

AFC (au to matic fre quency con trol) A cir cuit that au to mat i cally keeps an os cil la tor

on fre quency by com par ing the out put of the os cil la tor with a stan dard fre quency
source or sig nal.

air core An in duc tor with no mag netic ma te rial in its core.

al go rithm A pre scribed fi nite set of well-defined rules or pro cesses for the so lu tion of a

prob lem in a fi nite num ber of steps.

align ment The ad just ment of cir cuit com po nents so that an en tire sys tem meets min i -

mum per for mance val ues. For ex am ple, the stages in a ra dio are aligned to en sure
proper re cep tion.

al lo ca tion The planned use of cer tain fa cil i ties and equip ment to meet cur rent, pend -

ing, and/or fore casted cir cuit- and car rier-system re quire ments.

al ter nat ing cur rent (ac) A con tin u ously vari able cur rent, ris ing to a max i mum in one

di rec tion, fall ing to zero, then re vers ing di rec tion and ris ing to a max i mum in the
other di rec tion, then fall ing to zero and re peat ing the cy cle. Al ter nating cur rent usu -
ally fol lows a si nu soi dal growth and de cay curve. Note that the cor rect us age of the
term

ac is lower case.

al ter na tor A gen er a tor that pro duces al ter nat ing cur rent elec tric power.
am bi ent elec tro mag netic en vi ron ment The ra di ated or con ducted elec tro mag netic

sig nals and noise at a spe cific lo ca tion and time.

am bi ent level The mag ni tude of ra di ated or con ducted elec tro mag netic sig nals and

noise at a spe cific test lo ca tion when equip ment-under-test is not pow ered.

am bi ent tem per a ture The tem per a ture of the sur round ing me dium, typ i cally air, that

co mes into con tact with an ap pa ra tus. Am bi ent tem per a ture may also re fer sim ply to
room tem per a ture.

Amer i can Na tional Stan dards In sti tute (ANSI) A non profit or ga ni za tion that co or -

di nates vol un tary stan dards ac tiv i ties in the U.S.

Amer i can Wire Gauge (AWG) The stan dard Amer i can method of clas si fy ing wire

di am e ter.

am me ter An in stru ment that mea sures and re cords the amount of cur rent in am peres

flow ing in a cir cuit.

amp (A) An ab bre vi a tion of the term am pere.
ampacity A mea sure of the cur rent car ry ing ca pac ity of a power ca ble. Ampacity is de -

ter mined by the max i mum con tin u ous-performance tem per a ture of the in su la tion,
by the heat gen er ated in the ca ble (as a re sult of con duc tor and in su la tion losses), and
by the heat-dissipating prop er ties of the ca ble and its en vi ron ment.

am pere (amp) The stan dard unit of elec tric cur rent.

am pere per me ter The stan dard unit of mag netic field strength.

am pere-hour The en ergy that is con sumed when a cur rent of one am pere flows for a

pe riod of one hour.

am pere-turns The prod uct of the num ber of turns of a coil and the cur rent in am peres

flow ing through the coil.

am pli fi ca tion The pro cess that re sults when the out put of a cir cuit is an en larged re pro -

duc tion of the in put sig nal. Am pli fiers may be de signed to pro vide am pli fi ca tion of
volt age, cur rent, or power, or a com bi na tion of these quan ti ties.

am pli fi ca tion fac tor In a vac uum tube, the ra tio of the change in plate volt age to the

change in grid volt age that causes a cor re spond ing change in plate cur rent. Am pli fi -
ca tion fac tor is ex pressed by the Greek let ter

µ

(

mu).

am pli fier (1--gen eral) A de vice that re ceives an in put sig nal and pro vides as an out -

put a mag ni fied rep lica of the in put wave form.

(2--au dio) An am pli fier de signed to

cover the nor mal au dio fre quency range (20 Hz to 20 kHz).

(3--bal anced) A cir cuit

with two iden ti cal con nected sig nal branches that op er ate in phase op po si tion, with
in put and out put con nec tions each bal anced to ground.

(4--bridg ing) An am pli fy -

ing cir cuit fea tur ing high in put im ped ance to pre vent load ing of the source.

(5--broad band) An am pli fier ca pa ble of op er at ing over a spec i fied broad band of
fre quen cies with ac cept ably small am pli tude vari a tions as a func tion of fre quency.
(6--buffer) An am pli fier stage used to iso late a fre quency-sensitive cir cuit from
vari a tions in the load pre sented by fol low ing stages.

(7--lin ear) An am pli fier in

which the in stan ta neous out put sig nal is a lin ear func tion of the cor re spond ing in put
sig nal.

(8--mag netic) An am pli fier in cor po rat ing a con trol de vice de pend ent on

mag netic sat u ra tion. A small dc sig nal ap plied to a con trol cir cuit trig gers a large
change in op er at ing im ped ance and, hence, in the out put of the cir cuit.

(9--mi cro -

phone) A cir cuit that am pli fies the low level out put from a mi cro phone to make it
suf fi cient to be used as an in put sig nal to a power am pli fier or an other stage in a
mod u la tion cir cuit. Such a cir cuit is com monly known as a

preamplifier.

(10--push-pull) A bal anced am pli fier with two sim i lar am pli fy ing units con nected
in phase op po si tion in or der to can cel un de sired har mon ics and min i mize dis tor tion.
(11--tuned ra dio fre quency) An am pli fier tuned to a par tic u lar ra dio fre quency or
band so that only se lected fre quen cies are am pli fied.

am pli fier op er at ing class (1--gen eral) The op er at ing point of an am pli fy ing stage.

The op er at ing point, termed the op er at ing

class, de ter mines the pe riod dur ing which

cur rent flows in the out put.

(2--class A) An am pli fier in which out put cur rent flows

dur ing the whole of the in put cur rent cy cle.

(3--class AB) An am pli fier in which the

out put cur rent flows for more than half but less than the whole of the in put cy cle.
(4--class B) An am pli fier in which out put cur rent is cut off at zero in put sig nal; a
half-wave rec ti fied out put is pro duced.

(5--class C) An am pli fier in which out put

cur rent flows for less than half the in put cy cle.

(6--class D) An am pli fier op er at ing

in a pulse-only mode.

am pli tude The mag ni tude of a sig nal in volt age or cur rent, fre quently ex pressed in

terms of

peak, peak-to-peak, or root-mean-square (RMS). The ac tual am pli tude of a

quan tity at a par tic u lar in stant of ten var ies in a si nu soi dal man ner.

am pli tude dis tor tion A dis tor tion mech a nism oc cur ring in an am pli fier or other de -

vice when the out put am pli tude is not a lin ear func tion of the in put am pli tude un der
spec i fied con di tions.

am pli tude equal izer A cor rec tive net work that is de signed to mod ify the am pli tude

char ac ter is tics of a cir cuit or sys tem over a de sired fre quency range.

am pli tude-versus-frequency dis tor tion The dis tor tion in a trans mis sion sys tem

caused by the non uni form at ten u a tion or gain of the sys tem with re spect to fre -
quency un der spec i fied con di tions.

an a log car rier sys tem A car rier sys tem whose sig nal am pli tude, fre quency, or phase is

var ied con tin u ously as a func tion of a mod u lat ing in put.

an ode (1 -- gen eral) A pos i tive pole or el e ment. (2--vac uum tube) The out er most

pos i tive el e ment in a vac uum tube, also called the

plate.

(3--bat tery) The pos i tive

el e ment of a bat tery or cell.

an od ize The for ma tion of a thin film of ox ide on a me tal lic sur face, usu ally to pro duce

an in su lat ing layer.

an tenna (1--gen eral) A de vice used to trans mit or re ceive a ra dio sig nal. An an tenna

is usu ally de signed for a spec i fied fre quency range and serves to cou ple elec tro mag -
netic en ergy from a trans mis sion line to and/or from the free space through which it
trav els. Di rec tional an ten nas con cen trate the en ergy in a par tic u lar hor i zon tal or
ver ti cal di rec tion.

(2--aperiodic) An an tenna that is not pe ri odic or res o nant at par -

tic u lar fre quen cies, and so can be used over a wide band of fre quen cies.

(3--ar ti fi -

cial) A de vice that be haves, so far as the trans mit ter is con cerned, like a proper an -
tenna, but does not ra di ate any power at ra dio fre quen cies.

(4--broad band) An an -

tenna that op er ates within spec i fied per for mance lim its over a wide band of fre quen -
cies, with out re quir ing re tun ing for each in di vid ual fre quency.

(5--Cassegrain) A

dou ble re flect ing an tenna, of ten used for ground sta tions in sat el lite sys tems.
(6--co ax ial) A di pole an tenna made by fold ing back on it self a quar ter wave length
of the outer con duc tor of a co ax ial line, leav ing a quar ter wave length of the in ner
con duc tor ex posed.

(7--cor ner) An an tenna within the an gle formed by two

plane-reflecting sur faces.

(8--di pole) A cen ter-fed an tenna, one half-wavelength

long.

(9--di rec tional) An an tenna de signed to re ceive or emit ra di a tion more ef fi -

ciently in a par tic u lar di rec tion.

(10--dummy) An ar ti fi cial an tenna, de signed to

ac cept power from the trans mit ter but not to ra di ate it.

(11--fer rite) A com mon AM

broad cast re ceive an tenna that uses a small coil mounted on a short rod of fer rite ma -
te rial.

(12--flat top) An an tenna in which all the hor i zon tal com po nents are in the

same hor i zon tal plane.

(13--folded di pole) A ra di at ing de vice con sist ing of two or -

di nary half-wave di poles joined at their outer ends and fed at the cen ter of one of the
di poles.

(14--horn re flec tor) A ra di a tor in which the feed horn ex tends into a par a -

bolic re flec tor, and the power is ra di ated through a win dow in the horn.

(15--iso tro -

pic) A the o ret i cal an tenna in free space that trans mits or re ceives with the same ef fi -
ciency in all di rec tions.

(16--log-periodic) A broad band di rec tional an tenna in cor -

po rat ing an ar ray of di poles of dif fer ent lengths, the length and spac ing be tween di -
poles in creas ing logarithmically away from the feeder el e ment.

(17--long wire) An

an tenna made up of one or more con duc tors in a straight line point ing in the re quired
di rec tion with a to tal length of sev eral wave lengths at the op er at ing fre quency.
(18--loop) An an tenna con sist ing of one or more turns of wire in the same or par al -
lel planes.

(19--nested rhombic) An as sem bly of two rhombic an ten nas, one

smaller than the other, so that the com plete di a mond-shaped an tenna fits in side the
area oc cu pied by the larger unit.

(20--omnidirectional) An an tenna whose ra di at -

ing or re ceiv ing prop er ties are the same in all hor i zon tal plane di rec tions.

(21--pe -

ri odic) A res o nant an tenna de signed for use at a par tic u lar fre quency. (22--quar -

ter-wave) A di pole an tenna whose length is equal to one quar ter of a wave length at
the op er at ing fre quency.

(23--rhombic) A large di a mond-shaped an tenna, with

sides of the di a mond sev eral wave lengths long. The rhombic an tenna is fed at one of

the cor ners, with di rec tional ef fi ciency in the di rec tion of the di ag o nal.

(24--se ries

fed) A ver ti cal an tenna that is fed at its lower end. (25--shunt fed) A ver ti cal an -
tenna whose base is grounded, and is fed at a spec i fied point above ground. The
point at which the an tenna is fed above ground de ter mines the op er at ing im ped ance.

(26--steer able) An an tenna so con structed that its ma jor lobe may readily be
changed in di rec tion.

(27--top-loaded) A ver ti cal an tenna ca pac i tively loaded at its

up per end, of ten by sim ple en large ment or the at tach ment of a disc or plate.
(28--turn stile) An an tenna with one or more tiers of hor i zon tal di poles, crossed at
right an gles to each other and with ex ci ta tion of the di poles in phase quad ra ture.
(29--whip) An an tenna con structed of a thin semiflexible metal rod or tube, fed at
its base.

(30--Yagi) A di rec tional an tenna con structed of a se ries of di poles cut to

spe cific lengths.

Di rec tor el e ments are placed in front of the ac tive di pole and re -

flec tor el e ments are placed be hind the ac tive el e ment.

an tenna ar ray A group of sev eral an ten nas cou pled to gether to yield a re quired de gree

of directivity.

an tenna beam width The an gle be tween the half-power points (3 dB points) of the

main lobe of the an tenna pat tern when ref er enced to the peak power point of the an -
tenna pat tern.

An tenna beam width is mea sured in de grees and nor mally re fers to the

hor i zon tal ra di a tion pat tern.

an tenna directivity fac tor The ra tio of the power flux den sity in the de sired di rec tion

to the av er age value of power flux den sity at crests in the an tenna directivity pat tern
in the in ter fer ence sec tion.

an tenna fac tor A fac tor that, when ap plied to the volt age ap pear ing at the ter mi nals of

mea sure ment equip ment, yields the elec tri cal field strength at an an tenna. The unit
of an tenna fac tor is volts per me ter per mea sured volt.

an tenna gain The ra tio of the power re quired at the in put of a the o ret i cally per fect

omnidirectional ref er ence an tenna to the power sup plied to the in put of the given an -
tenna to pro duce the same field at the same dis tance. When not spec i fied oth er wise,
the fig ure ex press ing the gain of an an tenna re fers to the gain in the di rec tion of the
ra di a tion main lobe. In ser vices us ing

scat ter ing modes of prop a ga tion, the full gain

of an an tenna may not be re al iz able in prac tice and the ap par ent gain may vary with
time.

an tenna gain-to-noise tem per a ture For a sat el lite earth ter mi nal re ceiv ing sys tem, a

fig ure of merit that equals

G/T, where G is the gain in dB of the earth ter mi nal an -

tenna at the re ceive fre quency, and

T is the equiv a lent noise tem per a ture of the re -

ceiv ing sys tem in Kel vins.

an tenna match ing The pro cess of ad just ing an an tenna match ing cir cuit (or the an -

tenna it self) so that the in put im ped ance of the an tenna is equal to the char ac ter is tic
im ped ance of the trans mis sion line.

an tenna mon i tor A de vice used to mea sure the ra tio and phase be tween the cur rents

flow ing in the tow ers of a di rec tional AM broad cast sta tion.

an tenna noise tem per a ture The tem per a ture of a re sis tor hav ing an avail able noise

power per unit band width equal to that at the an tenna out put at a spec i fied fre -
quency.

an tenna pat tern A di a gram show ing the ef fi ciency of ra di a tion in all di rec tions from

the an tenna.

an tenna power rat ing The max i mum con tin u ous-wave power that can be ap plied to

an an tenna with out de grad ing its per for mance.

an tenna preamplifier A small am pli fier, usu ally mast-mounted, for am pli fy ing weak

sig nals to a level suf fi cient to com pen sate for down-lead losses.

ap par ent power The prod uct of the root-mean-square val ues of the volt age and cur -

rent in an al ter nat ing-current cir cuit with out a cor rec tion for the phase dif fer ence be -
tween the volt age and cur rent.

arc A sus tained lu mi nous dis charge be tween two or more elec trodes.

arith me tic mean The sum of the val ues of sev eral quan ti ties di vided by the num ber of

quan ti ties, also re ferred to as the

av er age.

ar ma ture wind ing The wind ing of an elec tri cal ma chine, ei ther a mo tor or gen er a tor,

in which cur rent is in duced.

ar ray (1--an tenna) An as sem bly of sev eral di rec tional an ten nas so placed and in ter -

con nected that directivity may be en hanced.

(2--broad side) An an tenna ar ray

whose el e ments are all in the same plane, pro duc ing a ma jor lobe per pen dic u lar to
the plane.

(3--colinear) An an tenna ar ray whose el e ments are in the same line, ei -

ther hor i zon tal or ver ti cal.

(4--end-fire) An an tenna ar ray whose el e ments are in

par al lel rows, one be hind the other, pro duc ing a ma jor lobe per pen dic u lar to the
plane in which in di vid ual el e ments are placed.

(5--lin ear) An an tenna ar ray whose

el e ments are ar ranged end-to-end.

(6--stacked) An an tenna ar ray whose el e ments

are stacked, one above the other.

ar ti fi cial line An as sem bly of re sis tors, inductors, and ca pac i tors that sim u lates the

elec tri cal char ac ter is tics of a trans mis sion line.

as sem bly A man u fac tured part made by com bin ing sev eral other parts or sub as sem -

blies.

as sumed val ues A range of val ues, pa ram e ters, lev els, and other el e ments as sumed for

a math e mat i cal model, hy po thet i cal cir cuit, or net work, from which anal y sis, ad di -
tional es ti mates, or cal cu la tions will be made. The range of val ues, while not mea -
sured, rep re sents the best en gi neer ing judg ment and is gen er ally de rived from val -
ues found or mea sured in real cir cuits or net works of the same ge neric type, and in -
cludes pro jected im prove ments.

at mo sphere The gas eous en ve lope sur round ing the earth, com posed largely of ox y -

gen, car bon di ox ide, and wa ter va por. The at mo sphere is di vided into four pri mary
lay ers:

tro po sphere, strato sphere, ion o sphere, and exo sphere.

at mo spheric noise Ra dio noise caused by nat u ral at mo spheric pro cesses, such as

light ning.

at tack time The time in ter val in sec onds re quired for a de vice to re spond to a con trol

stim u lus.

at ten u a tion The de crease in am pli tude of an elec tri cal sig nal trav el ing through a trans -

mis sion me dium caused by di elec tric and con duc tor losses.

at ten u a tion co ef fi cient The rate of de crease in the am pli tude of an elec tri cal sig nal

caused by at ten u a tion. The

at ten u a tion co ef fi cient can be ex pressed in deci bels or

nepers per unit length. It may also be re ferred to as the

at ten u a tion con stant.

at ten u a tion dis tor tion The dis tor tion caused by at ten u a tion that var ies over the fre -

quency range of a sig nal.

at ten u a tion-limited op er a tion The con di tion pre vail ing when the re ceived sig nal am -

pli tude (rather than dis tor tion) lim its over all sys tem per for mance.

at tenu ator A fixed or ad just able com po nent that re duces the am pli tude of an elec tri cal

sig nal with out caus ing dis tor tion.

atto A pre fix mean ing one quin til lionth.

at trac tion The at trac tive force be tween two un like mag netic poles (N/S) or elec tri cally

charged bod ies (+/-).

at trib utes The char ac ter is tics of equip ment that aid plan ning and cir cuit de sign.

au to matic fre quency con trol (AFC) A sys tem de signed to main tain the cor rect op er -

at ing fre quency of a re ceiver. Any drift in tun ing re sults in the pro duc tion of a con -
trol volt age, which is used to ad just the fre quency of a lo cal os cil la tor so as to min i -
mize the tun ing er ror.

au to matic gain con trol (AGC) An elec tronic cir cuit that com pares the level of an in -

com ing sig nal with a pre vi ously de fined stan dard and au to mat i cally am pli fies or at -
tenu ates the sig nal so it ar rives at its des ti na tion at the cor rect level.

autotransformer A trans former in which both the pri mary and sec ond ary cur rents

flow through one com mon part of the coil.

aux il iary power An al ter nate source of elec tric power, serv ing as a back-up for the pri -

mary util ity com pany ac power.

avail abil ity A mea sure of the de gree to which a sys tem, sub sys tem, or equip ment is op -

er a ble and not in a stage of con ges tion or fail ure at any given point in time.

av a lanche ef fect The ef fect ob tained when the elec tric field across a bar rier re gion is

suf fi ciently strong for elec trons to col lide with

va lence elec trons, thereby re leas ing

more elec trons and giv ing a cu mu la tive mul ti pli ca tion ef fect in a semi con duc tor.

av er age life The mean value for a nor mal dis tri bu tion of prod uct or com po nent lives,

gen er ally ap plied to me chan i cal fail ures re sult ing from "wear-out."

B

back emf A volt age in duced in the re verse di rec tion when cur rent flows through an in -

duc tance.

Back emf is also known as coun ter-emf.

back scat ter ing A form of wave scat ter ing in which at least one com po nent of the scat -

tered wave is de flected op po site to the di rec tion of prop a ga tion of the in ci dent wave.

back ground noise The to tal sys tem noise in the ab sence of in for ma tion trans mis sion,

in de pend ent of the pres ence or ab sence of a sig nal.

back scat ter The de flec tion or re flec tion of ra di ant en ergy through an gles greater than

90° with re spect to the orig i nal an gle of travel.

back scat ter range The max i mum dis tance from which back scat tered ra di ant en ergy

can be mea sured.

backup A cir cuit el e ment or fa cil ity used to re place an el e ment that has failed.
backup sup ply A re dun dant power sup ply that takes over if the pri mary power sup ply

fails.

bal ance The pro cess of equal iz ing the volt age, cur rent, or other pa ram e ter be tween two

or more cir cuits or sys tems.

bal anced A cir cuit hav ing two sides (con duc tors) car ry ing volt ages that are sym met ri -

cal about a com mon ref er ence point, typ i cally ground.

bal anced cir cuit A cir cuit whose two sides are elec tri cally equal in all trans mis sion re -

spects.

bal anced line A trans mis sion line con sist ing of two con duc tors in the pres ence of

ground ca pa ble of be ing op er ated in such a way that when the volt ages of the two
con duc tors at all trans verse planes are equal in mag ni tude and op po site in po lar ity
with re spect to ground, the cur rents in the two con duc tors are equal in mag ni tude
and op po site in di rec tion.

bal anced mod u la tor A mod u la tor that com bines the in for ma tion sig nal and the car rier

so that the out put con tains the two side bands with out the car rier.

bal anced three-wire sys tem A power dis tri bu tion sys tem us ing three con duc tors, one

of which is bal anced to have a po ten tial mid way be tween the po ten tials of the other
two.

bal anced-to-ground The con di tion when the im ped ance to ground on one wire of a

two-wire cir cuit is equal to the im ped ance to ground on the other wire.

balun (bal anced/un bal anced) A de vice used to con nect bal anced cir cuits with un bal -

anced cir cuits.

band A range of fre quen cies be tween a spec i fied up per and lower limit.

band elim i na tion fil ter A fil ter hav ing a sin gle con tin u ous at ten u a tion band, with nei -

ther the up per nor lower cut-off fre quen cies be ing zero or in fi nite. A

band elim i na -

tion fil ter may also be re ferred to as a band-stop, notch, or band re ject fil ter.

bandpass fil ter A fil ter hav ing a sin gle con tin u ous trans mis sion band with nei ther the

up per nor the lower cut-off fre quen cies be ing zero or in fi nite. A bandpass fil ter per -
mits only a spe cific band of fre quen cies to pass; fre quen cies above or be low are at -
ten u ated.

band width The range of sig nal fre quen cies that can be trans mit ted by a com mu ni ca -

tions chan nel with a de fined max i mum loss or dis tor tion. Band width in di cates the
in for ma tion-carrying ca pac ity of a chan nel.

band width ex pan sion ra tio The ra tio of the nec es sary band width to the base band

band width.

band width-limited op er a tion The con di tion pre vail ing when the fre quency spec trum

or band width, rather than the am pli tude (or power) of the sig nal, is the lim it ing fac -
tor in com mu ni ca tion ca pa bil ity. This con di tion is reached when the sys tem dis torts
the shape of the wave form be yond tol er a ble lim its.

bank A group of sim i lar items con nected to gether in a spec i fied man ner and used in

con junc tion with one an other.

bare A wire con duc tor that is not enam eled or en closed in an in su lat ing sheath.

base band The band of fre quen cies oc cu pied by a sig nal be fore it mod u lates a car rier

wave to form a trans mit ted ra dio or line sig nal.

base band chan nel A chan nel that car ries a sig nal with out mod u la tion, in con trast to a

passband chan nel.

base band sig nal The orig i nal form of a sig nal, un changed by mod u la tion.
bath tub The shape of a typ i cal graph of com po nent fail ure rates: high dur ing an ini tial

pe riod of op er a tion, fall ing to an ac cept able low level dur ing the nor mal us age pe -
riod, and then ris ing again as the com po nents be come time-expired.

bat tery A group of sev eral cells con nected to gether to fur nish cur rent by con ver sion of

chem i cal, ther mal, so lar, or nu clear en ergy into elec tri cal en ergy. A sin gle cell is it -
self some times also called a bat tery.

bay A row or suite of racks on which trans mis sion, switch ing, and/or pro cess ing equip -

ment is mounted.

Bel A unit of power mea sure ment, named in honor of Al ex an der Gra ham Bell. The

com monly used unit is one tenth of a Bel, or a deci bel (dB). One Bel is de fined as a
ten fold in crease in power. If an am pli fier in creases the power of a sig nal by 10 times,
the power gain of the am pli fier is equal to 1 Bel or 10

deci bels (dB). If power is in -

creased by 100 times, the power gain is 2 Bels or 20 deci bels.

bend A tran si tion com po nent be tween two el e ments of a trans mis sion wave guide.

bend ing ra dius The small est bend that may be put into a ca ble un der a stated pull ing

force. The bend ing ra dius is typ i cally ex pressed in inches.

bias A dc volt age dif fer ence ap plied be tween two el e ments of an ac tive elec tronic de -

vice, such as a vac uum tube, tran sis tor, or in te grated cir cuit. Bias cur rents may or
may not be drawn, de pend ing on the de vice and cir cuit type.

bidirectional An op er a tional qual i fi ca tion which im plies that the trans mis sion of in -

for ma tion oc curs in both di rec tions.

bi fi lar wind ing A type of wind ing in which two in su lated wires are placed side by side.

In some com po nents, bi fi lar wind ing is used to pro duce bal anced cir cuits.

bi po lar A sig nal that con tains both pos i tive-going and neg a tive-going am pli tude com -

po nents. A bi po lar sig nal may also con tain a zero am pli tude state.

bleeder A high re sis tance con nected in par al lel with one or more fil ter ca pac i tors in a

high volt age dc sys tem. If the power sup ply load is dis con nected, the ca pac i tors dis -
charge through the bleeder.

block di a gram An over view di a gram that uses geo met ric fig ures to rep re sent the prin -

ci pal di vi sions or sec tions of a cir cuit, and lines and ar rows to show the path of a sig -
nal, or to show pro gram functionalities. It is not a

sche matic, which pro vides greater

de tail.

block ing ca pac i tor A ca pac i tor in cluded in a cir cuit to stop the pas sage of di rect cur -

rent.

BNC An ab bre vi a tion for bay o net Neill-Concelman, a type of ca ble con nec tor used ex -

ten sively in RF ap pli ca tions (named for its in ven tor).

Boltzmann's con stant 1.38 x 10

23

joules.

bridge A type of net work cir cuit used to match dif fer ent cir cuits to each other, en sur ing

min i mum trans mis sion im pair ment.

bridg ing The shunt ing or par al lel ing of one cir cuit with an other.

broad band The qual ity of a com mu ni ca tions link hav ing es sen tially uni form re sponse

over a given range of fre quen cies. A com mu ni ca tions link is said to be

broad band if

it of fers no per cep ti ble deg ra da tion to the sig nal be ing trans ported.

buffer A cir cuit or com po nent that iso lates one elec tri cal cir cuit from an other.
burn-in The op er a tion of a de vice, some times un der ex treme con di tions, to sta bi lize its

char ac ter is tics and iden tify la tent com po nent fail ures be fore bring ing the de vice
into nor mal ser vice.

bus A cen tral con duc tor for the pri mary sig nal path. The term bus may also re fer to a

sig nal path to which a num ber of in puts may be con nected for feed to one or more
out puts.

busbar A main dc power bus.
by pass ca pac i tor A ca pac i tor that pro vides a sig nal path that ef fec tively shunts or by -

passes other com po nents.

by pass re lay A switch used to by pass the nor mal elec tri cal route of a sig nal or cur rent

in the event of power, sig nal, or equip ment fail ure.

C

ca ble An elec tri cally and/or op ti cally con duc tive in ter con nect ing de vice.

ca ble loss Sig nal loss caused by pass ing a sig nal through a co ax ial ca ble. Losses are the

re sult of re sis tance, ca pac i tance, and in duc tance in the ca ble.

ca ble splice The con nec tion of two pieces of ca ble by join ing them me chan i cally and

clos ing the joint with a weather-tight case or sleeve.

Glossary 621

ca bling The wir ing used to in ter con nect elec tronic equip ment.

cal i brate The pro cess of check ing, and ad just ing if nec es sary, a test in stru ment against

one known to be set cor rectly.

cal i bra tion The pro cess of iden ti fy ing and mea sur ing er rors in in stru ments and/or pro -

ce dures.

ca pac i tance The prop erty of a de vice or com po nent that en ables it to store en ergy in an

elec tro static field and to re lease it later. A ca pac i tor con sists of two con duc tors sep a -
rated by an in su lat ing ma te rial. When the con duc tors have a volt age dif fer ence be -
tween them, a charge will be stored in the elec tro static field be tween the con duc tors.

ca pac i tor A de vice that stores elec tri cal en ergy. A ca pac i tor al lows the ap par ent flow

of al ter nat ing cur rent, while block ing the flow of di rect cur rent. The de gree to which
the de vice per mits ac cur rent flow de pends on the fre quency of the sig nal and the
size of the ca pac i tor. Ca pac i tors are used in fil ters, de lay-line com po nents, cou plers,
fre quency se lec tors, tim ing el e ments, volt age tran sient sup pres sion, and other ap -
pli ca tions.

car rier A sin gle fre quency wave that, prior to trans mis sion, is mod u lated by an other

wave con tain ing in for ma tion. A car rier may be mod u lated by ma nip u lat ing its am -
pli tude and/or fre quency in di rect re la tion to one or more ap plied sig nals.

car rier fre quency The fre quency of an unmodulated os cil la tor or trans mit ter. Also,

the av er age fre quency of a trans mit ter when a sig nal is fre quency mod u lated by a
sym met ri cal sig nal.

cas cade con nec tion A tan dem ar range ment of two or more sim i lar com po nent de vices

or cir cuits, with the out put of one con nected to the in put of the next.

cas caded An ar range ment of two or more cir cuits in which the out put of one cir cuit is

con nected to the in put of the next cir cuit.

cath ode ray tube (CRT) A vac uum tube de vice, usu ally glass, that is nar row at one

end and wid ens at the other to cre ate a sur face onto which im ages can be pro jected.
The nar row end con tains the nec es sary cir cuits to gen er ate and fo cus an elec tron
beam on the lu mi nes cent screen at the other end. CRTs are used to dis play pic tures
in TV re ceiv ers, video mon i tors, os cil lo scopes, com put ers, and other sys tems.

cell An el e men tary unit of com mu ni ca tion, of power sup ply, or of equip ment.

Cel sius A tem per a ture mea sure ment scale, ex pressed in de grees C, in which wa ter

freezes at 0°C and boils at 100°C. To con vert to de grees Fahr en heit, mul ti ply by
0.555 and add 32. To con vert to Kel vins add 273 (ap prox i mately).

cen ter fre quency In fre quency mod u la tion, the rest ing fre quency or ini tial fre quency

of the car rier be fore mod u la tion.

cen ter tap A con nec tion made at the elec tri cal cen ter of a coil.
chan nel The small est sub di vi sion of a cir cuit that pro vides a sin gle type of com mu ni -

ca tion ser vice.

chan nel de coder A de vice that con verts an in com ing mod u lated sig nal on a given

chan nel back into the source-encoded sig nal.

chan nel en coder A de vice that takes a given sig nal and con verts it into a form suit able

for trans mis sion over the com mu ni ca tions chan nel.

chan nel noise level The ra tio of the chan nel noise at any point in a trans mis sion sys tem

to some ar bi trary amount of cir cuit noise cho sen as a ref er ence. This ra tio is usu ally
ex pressed in

deci bels above ref er ence noise, ab bre vi ated dBrn.

chan nel re li abil ity The per cent of time a chan nel is avail able for use in a spe cific di -

rec tion dur ing a spec i fied pe riod.

channelization The al lo ca tion of com mu ni ca tion cir cuits to chan nels and the form ing

of these chan nels into groups for higher or der multiplexing.

char ac ter is tic The prop erty of a cir cuit or com po nent.

char ac ter is tic im ped ance The im ped ance of a trans mis sion line, as mea sured at the

driv ing point, if the line were of in fi nite length. In such a line, there would be no
stand ing waves. The

char ac ter is tic im ped ance may also be re ferred to as the surge

im ped ance.

charge The pro cess of re plen ish ing or re plac ing the elec tri cal charge in a sec ond ary

cell or stor age bat tery.

charger A de vice used to re charge a bat tery. Types of charg ing in clude: (1) con stant

volt age charge, (2) equal iz ing charge, and (3) trickle charge.

chas sis ground A con nec tion to the metal frame of an elec tronic sys tem that holds the

com po nents in a place. The chas sis ground con nec tion serves as the ground re turn or
elec tri cal com mon for the sys tem.

cir cuit Any closed path through which an elec tri cal cur rent can flow. In a par al lel cir -

cuit, com po nents are con nected be tween com mon in puts and out puts such that all
paths are par al lel to each other. The same volt age ap pears across all paths. In a

se ries

cir cuit, the same cur rent flows through all com po nents.

cir cuit noise level The ra tio of the cir cuit noise at some given point in a trans mis sion

sys tem to an es tab lished ref er ence, usu ally ex pressed in deci bels above the ref er -
ence.

cir cuit re li abil ity The per cent age of time a cir cuit is avail able to the user dur ing a spec -

i fied pe riod of sched uled avail abil ity.

cir cu lar mil The mea sure ment unit of the cross-sectional area of a cir cu lar con duc tor.

A

cir cu lar mil is the area of a cir cle whose di am e ter is one mil, or 0.001 inch.

clear chan nel A trans mis sion path wherein the full band width is avail able to the user,

with no por tions of the chan nel used for con trol, fram ing, or sig nal ing. Can also re -
fer to a clas si fi ca tion of AM broad cast sta tion.

clip per A lim it ing cir cuit which en sures that a spec i fied out put level is not ex ceeded by

re strict ing the out put wave form to a max i mum peak am pli tude.

clip ping The dis tor tion of a sig nal caused by re mov ing a por tion of the wave form

through re stric tion of the am pli tude of the sig nal by a cir cuit or de vice.

coax A short-hand ex pres sion for co ax ial ca ble, which is used to trans port

high-frequency sig nals.

co ax ial ca ble A trans mis sion line con sist ing of an in ner con duc tor sur rounded first by

an in su lat ing ma te rial and then by an outer con duc tor, ei ther solid or braided.The
me chan i cal di men sions of the ca ble de ter mine its

char ac ter is tic im ped ance.

co her ence The cor re la tion be tween the phases of two or more waves.

co her ent The con di tion char ac ter ized by a fixed phase re la tion ship among points on an

elec tro mag netic wave.

co her ent pulse The con di tion in which a fixed phase re la tion ship is main tained be -

tween con sec u tive pulses dur ing pulse trans mis sion.

cold joint A sol dered con nec tion that was in ad e quately heated, with the re sult that the

wire is held in place by rosin flux, not sol der. A cold joint is some times re ferred to as
a

dry joint.

comb fil ter An elec tri cal fil ter cir cuit that passes a se ries of fre quen cies and re jects the

fre quen cies in be tween, pro duc ing a fre quency re sponse sim i lar to the teeth of a
comb.

com mon A point that acts as a ref er ence for cir cuits, of ten equal in po ten tial to the lo cal

ground.

com mon mode Sig nals iden ti cal with re spect to am pli tude, fre quency, and phase that

are ap plied to both ter mi nals of a ca ble and/or both the in put and ref er ence of an am -
pli fier.

com mon re turn A re turn path that is com mon to two or more cir cuits, and re turns cur -

rents to their source or to ground.

com mon re turn off set The dc com mon re turn po ten tial dif fer ence of a line.

com mu ni ca tions sys tem A col lec tion of in di vid ual com mu ni ca tions net works, trans -

mis sion sys tems, re lay sta tions, trib u tary sta tions, and ter mi nal equip ment ca pa ble
of in ter con nec tion and interoperation to form an in te gral whole. The in di vid ual
com po nents must serve a com mon pur pose, be tech ni cally com pat i ble, em ploy
com mon pro ce dures, re spond to some form of con trol, and, in gen eral, op er ate in
uni son.

com mu ta tion A suc ces sive switch ing pro cess car ried out by a com mu ta tor.

com mu ta tor A cir cu lar as sem bly of con tacts, in su lated one from an other, each lead ing

to a dif fer ent por tion of the cir cuit or ma chine.

com pat i bil ity The abil ity of di verse sys tems to ex change nec es sary in for ma tion at ap -

pro pri ate lev els of com mand di rectly and in us able form. Com mu ni ca tions equip -
ment items are com pat i ble if sig nals can be ex changed be tween them with out the
ad di tion of buff er ing or trans la tion for the spe cific pur pose of achiev ing work able
in ter face con nec tions, and if the equip ment or sys tems be ing in ter con nected pos sess
com pa ra ble per for mance char ac ter is tics, in clud ing the sup pres sion of un de sired ra -
di a tion.

com plex wave A wave form con sist ing of two or more sinewave com po nents. At any

in stant of time, a com plex wave is the al ge braic sum of all its sinewave com po nents.

com pli ance For me chan i cal sys tems, a prop erty which is the re cip ro cal of stiff ness.
com po nent An as sem bly, or part thereof, that is es sen tial to the op er a tion of some

larger cir cuit or sys tem. A

com po nent is an im me di ate sub di vi sion of the as sem bly

to which it be longs.

COM SAT The Com mu ni ca tions Sat el lite Cor po ra tion, an or ga ni za tion es tab lished by

an act of Con gress in 1962. COM SAT launches and op er ates the in ter na tional sat el -
lites for the INTELSAT con sor tium of coun tries.

con cen tric ity A mea sure of the de vi a tion of the cen ter con duc tor po si tion rel a tive to its

ideal lo ca tion in the ex act cen ter of the di elec tric cross-section of a co ax ial ca ble.

con di tion ing The ad just ment of a chan nel in or der to pro vide the ap pro pri ate trans mis -

sion char ac ter is tics needed for data or other spe cial ser vices.

con di tion ing equip ment The equip ment used to match trans mis sion lev els and im -

ped ances, and to pro vide equal iza tion be tween fa cil i ties.

con duc tance A mea sure of the ca pa bil ity of a ma te rial to con duct elec tric ity. It is the

re cip ro cal of

re sis tance (ohm) and is ex pressed in siemens. (For merly ex pressed as

mho.)

con ducted emis sion An elec tro mag netic en ergy prop a gated along a con duc tor.

con duc tion The trans fer of en ergy through a me dium, such as the con duc tion of elec -

tric ity by a wire, or of heat by a me tal lic frame.

con duc tion band A par tially filled or empty atomic en ergy band in which elec trons are

free to move eas ily, al low ing the ma te rial to carry an elec tric cur rent.

con duc tiv ity The con duc tance per unit length.
con duc tor Any ma te rial that is ca pa ble of car ry ing an elec tric cur rent.

con fig u ra tion A rel a tive ar range ment of parts.

con nec tion A point at which a junc tion of two or more con duc tors is made.
con nec tor A de vice mounted on the end of a wire or fi ber op tic ca ble that mates to a

sim i lar de vice on a spe cific piece of equip ment or an other ca ble.

con stant-current source A source with in fi nitely high out put im ped ance so that out -

put cur rent is in de pend ent of volt age, for a spec i fied range of out put volt ages.

con stant-voltage charge A method of charg ing a sec ond ary cell or stor age bat tery dur -

ing which the ter mi nal volt age is kept at a con stant value.

con stant-voltage source A source with low, ide ally zero, in ter nal im ped ance, so that

volt age will re main con stant, in de pend ent of cur rent sup plied.

con tact The points that are brought to gether or sep a rated to com plete or break an elec -

trical cir cuit.

con tact bounce The re bound of a con tact, which tem po rarily opens the cir cuit af ter its

ini tial

make.

con tact form The con fig u ra tion of a con tact as sem bly on a re lay. Many dif fer ent con -

fig u ra tions are pos si ble from sim ple

sin gle-make con tacts to com plex ar range ments

in volv ing

breaks and makes.

con tact noise A noise re sult ing from cur rent flow through an elec tri cal con tact that has

a rap idly vary ing re sis tance, as when the con tacts are cor roded or dirty.

con tact re sis tance The re sis tance at the sur face when two con duc tors make con tact.
con ti nu ity A con tin u ous path for the flow of cur rent in an elec tri cal cir cuit.

con tin u ous wave An elec tro mag netic sig nal in which suc ces sive os cil la tions of the

waves are iden ti cal.

con trol The su per vi sion that an op er a tor or de vice ex er cises over a cir cuit or sys tem.

con trol grid The grid in an elec tron tube that con trols the flow of cur rent from the cath -

ode to the an ode.

con ven tion A gen er ally ac cept able sym bol, sign, or prac tice in a given in dus try.
Co or di nated Uni ver sal Time (UTC) The time scale, main tained by the BIH (Bu reau

In ter na tional de l'Heure) that forms the ba sis of a co or di nated dis sem i na tion of stan -
dard fre quen cies and time sig nals.

cop per loss The loss re sult ing from the heat ing ef fect of cur rent.

co rona A blu ish lu mi nous dis charge re sult ing from ion iza tion of the air near a con duc -

tor car ry ing a volt age gra di ent above a cer tain

crit i cal level.

cor rec tive main te nance The nec es sary tests, mea sure ments, and ad just ments re -

quired to re move or cor rect a fault.

cos mic noise The ran dom noise orig i nat ing out side the earth's at mo sphere.

cou lomb The stan dard unit of elec tric quan tity or charge. One cou lomb is equal to the

quan tity of elec tric ity trans ported in 1 sec ond by a cur rent of 1 am pere.

Cou lomb's Law The at trac tion and re pul sion of elec tric charges act on a line be tween

them. The charges are in versely pro por tional to the square of the dis tance be tween
them, and pro por tional to the prod uct of their mag ni tudes. (Named for the French
phys i cist Charles-Augustine de Cou lomb, 1736-1806.)

coun ter-electromotive force The ef fec tive elec tro mo tive force within a sys tem that

op poses the pas sage of cur rent in a spec i fied di rec tion.

cou ple The pro cess of link ing two cir cuits by in duc tance, so that en ergy is trans ferred

from one cir cuit to an other.

cou pled mode The se lec tion of ei ther ac or dc cou pling.

cou pling The re la tion ship be tween two com po nents that en ables the trans fer of en ergy

be tween them. In cluded are

di rect cou pling through a di rect elec tri cal con nec tion,

such as a wire;

ca pac i tive cou pling through the ca pac i tance formed by two ad ja cent

con duc tors; and

in duc tive cou pling in which en ergy is trans ferred through a mag -

netic field. Ca pac i tive cou pling is also called

elec tro static cou pling. In duc tive cou -

pling is of ten re ferred to as

elec tro mag netic cou pling.

cou pling co ef fi cient A mea sure of the elec tri cal cou pling that ex ists be tween two cir -

cuits. The

cou pling co ef fi cient is equal to the ra tio of the mu tual im ped ance to the

square root of the prod uct of the self im ped ances of the cou pled cir cuits.

cross cou pling The cou pling of a sig nal from one chan nel, cir cuit, or con duc tor to an -

other, where it be comes an un de sired sig nal.

cross over dis tor tion A dis tor tion that re sults in an am pli fier when an ir reg u lar ity is in -

tro duced into the sig nal as it crosses through a zero ref er ence point. If an am pli fier is
prop erly de signed and bi ased, the up per half cy cle and lower half cy cle of the sig nal
co in cide at the zero cross over ref er ence.

cross over fre quency The fre quency at which out put sig nals pass from one chan nel to

the other in a

cross over net work. At the cross over fre quency it self, the out puts to

each side are equal.

cross over net work A type of fil ter that di vides an in com ing sig nal into two or more

out puts, with higher fre quen cies di rected to one out put, and lower fre quen cies to an -
other.

crosstalk Un de sired trans mis sion of sig nals from one cir cuit into an other cir cuit in the

same sys tem. Crosstalk is usu ally caused by un in ten tional ca pac i tive (ac) cou pling.

crosstalk cou pling The ra tio of the power in a dis turb ing cir cuit to the in duced power

in the dis turbed cir cuit, ob served at a par tic u lar point un der spec i fied con di tions.
Crosstalk cou pling is typ i cally ex pressed in dB.

crow bar A short-circuit or low re sis tance path placed across the in put to a cir cuit, usu -

ally for pro tec tive pur poses..

CRT (cath ode ray tube) A vac uum tube de vice that pro duces light when en er gized by

the elec tron beam gen er ated in side the tube. A CRT in cludes an elec tron gun, de -
flec tion mech a nism, and phos phor-covered face plate.

crys tal A so lid i fied form of a sub stance that has at oms and mol e cules ar ranged in a

sym met ri cal pat tern.

crys tal fil ter A fil ter that uses pi ezo elec tric crys tals to cre ate res o nant or antiresonant

cir cuits.

crys tal os cil la tor An os cil la tor us ing a pi ezo elec tric crys tal as the tuned cir cuit that

con trols the res o nant fre quency.

crys tal-controlled os cil la tor An os cil la tor in which a pi ezo elec tric-effect crys tal is

cou pled to a tuned os cil la tor cir cuit in such a way that the crys tal pulls the os cil la tor
fre quency to its own nat u ral fre quency and does not al low fre quency drift.

cur rent (1--gen eral) A gen eral term for the trans fer of elec tric ity, or the move ment of

elec trons or

holes.

(2--al ter nat ing) An elec tric cur rent that is con stantly vary ing in

am pli tude and pe ri od i cally re vers ing di rec tion.

(3--av er age) The arith me tic mean

of the in stan ta neous val ues of cur rent, av er aged over one com plete half cy cle.
(4--charg ing) The cur rent that flows in to charge a ca pac i tor when it is first con -
nected to a source of elec tric po ten tial.

(5--di rect) Elec tric cur rent that flows in one

di rec tion only.

(6--eddy) A waste ful cur rent that flows in the core of a trans former

and pro duces heat.

Eddy cur rents are largely elim i nated through the use of lam i -

nated cores.

(7--ef fec tive) The ac cur rent that will pro duce the same ef fec tive heat

in a re sis tor as is pro duced by dc. If the ac is si nu soi dal, the

ef fec tive cur rent value is

0.707 times the peak ac value.

(8--fault) The cur rent that flows be tween con duc tors

or to ground dur ing a fault con di tion.

(9--ground fault) A fault cur rent that flows to

ground.

(10--ground re turn) A cur rent that re turns through the earth. (11--lag -

ging) A phe nom e non ob served in an in duc tive cir cuit where al ter nat ing cur rent lags
be hind the volt age that pro duces it.

(12--lead ing) A phe nom e non ob served in a ca -

pac i tive cir cuit where al ter nat ing cur rent leads the volt age that pro duces it.
(13--mag ne tiz ing) The cur rent in a trans former pri mary wind ing that is just suf fi -
cient to mag ne tize the core and off set iron losses.

(14--neu tral) The cur rent that

flows in the neu tral con duc tor of an un bal anced polyphase power cir cuit. If cor -
rectly bal anced, the neu tral would carry no net cur rent.

(15--peak) The max i mum

value reached by a vary ing cur rent dur ing one cy cle.

(16--pick-up) The min i mum

cur rent at which a re lay just be gins to op er ate.

(17--plate) The an ode cur rent of an

elec tron tube.

(18--re sid ual) The vec tor sum of the cur rents in the phase wires of an

un bal anced polyphase power cir cuit.

(19--space) The to tal cur rent flow ing through

an elec tron tube.

cur rent am pli fier A low out put im ped ance am pli fier ca pa ble of pro vid ing high cur -

rent out put.

cur rent probe A sen sor, clamped around an elec tri cal con duc tor, in which an in duced

cur rent is de vel oped from the mag netic field sur round ing the con duc tor. For mea -
sure ments, the cur rent probe is con nected to a suit able test in stru ment.

cur rent trans former A trans former-type of in stru ment in which the pri mary car ries

the cur rent to be mea sured and the sec ond ary is in se ries with a low cur rent am me ter.
A cur rent trans former is used to mea sure high val ues of al ter nat ing cur rent.

cur rent-carrying ca pac ity A mea sure of the max i mum cur rent that can be car ried

con tin u ously with out dam age to com po nents or de vices in a cir cuit.

cut-off fre quency The fre quency above or be low which the out put cur rent in a cir cuit

is re duced to a spec i fied level.

cy cle The in ter val of time or space re quired for a pe ri odic sig nal to com plete one pe -

riod.

cy cles per sec ond The stan dard unit of fre quency, ex pressed in Hertz (one cy cle per

sec ond).

D

damped os cil la tion An os cil la tion ex hib it ing a pro gres sive dim i nu tion of am pli tude

with time.

damp ing The dis si pa tion and re sul tant re duc tion of any type of en ergy, such as elec tro -

mag netic waves.

dB (deci bel) A mea sure of volt age, cur rent, or power gain equal to 0.1 Bel. Deci bels

are given by the equa tions

20 log

V

V

out

in

, 20 log

I

I

out

in

, or 10 log

P

P

out

in

.

dBk A mea sure of power rel a tive to 1 ki lo watt. 0 dBk equals 1 kW.

dBm (deci bels above 1 milliwatt) A log a rith mic mea sure of power with re spect to a

ref er ence power of one milliwatt.

dBmv A mea sure of volt age gain rel a tive to 1 mil li volt at 75 ohms.

dBr The power dif fer ence ex pressed in dB be tween any point and a ref er ence point se -

lected as the

zero rel a tive trans mis sion level point. A power ex pressed in dBr does

not spec ify the ab so lute power; it is a rel a tive mea sure ment only.

dBu A term that re flects com par i son be tween a mea sured value of volt age and a ref er -

ence value of 0.775 V, ex pressed un der con di tions in which the im ped ance at the
point of mea sure ment (and of the ref er ence source) are not con sid ered.

dbV A mea sure of volt age gain rel a tive to 1 V.

dBW A mea sure of power rel a tive to 1 watt. 0 dBW equals 1 W.

dc An ab bre vi a tion for di rect cur rent. Note that the pre ferred us age of the term dc is

lower case.

dc am pli fier A cir cuit ca pa ble of am pli fy ing dc and slowly vary ing al ter nat ing cur rent

sig nals.

dc com po nent The por tion of a sig nal that con sists of di rect cur rent. This term may

also re fer to the av er age value of a sig nal.

dc cou pled A con nec tion con fig ured so that both the sig nal (ac com po nent) and the

con stant volt age on which it is rid ing (dc com po nent) are passed from one stage to
the next.

dc cou pling A method of cou pling one cir cuit to an other so as to trans mit the static (dc)

char ac ter is tics of the sig nal as well as the vary ing (ac) char ac ter is tics. Any dc off set
pres ent on the in put sig nal is main tained and will be pres ent in the out put.

dc off set The amount that the dc com po nent of a given sig nal has shifted from its cor -

rect level.

dc sig nal bounce Over shoot of the proper dc volt age level re sult ing from mul ti ple ac

cou plings in a sig nal path.

de-energized A sys tem from which sources of power have been dis con nected.

deca A pre fix mean ing ten.

de cay The re duc tion in am pli tude of a sig nal on an ex po nen tial ba sis.
de cay time The time re quired for a sig nal to fall to a cer tain frac tion of its orig i nal

value.

deci bel (dB) One tenth of a Bel. The deci bel is a log a rith mic mea sure of the ra tio be -

tween two pow ers.

de code The pro cess of re cov er ing in for ma tion from a sig nal into which the in for ma -

tion has been en coded.

de coder A de vice ca pa ble of de ci pher ing en coded sig nals. A de coder in ter prets in put

in struc tions and ini ti ates the ap pro pri ate con trol op er a tions as a re sult.

de coup ling The re duc tion or re moval of un de sired cou pling be tween two cir cuits or

stages.

deemphasis The re duc tion of the high-frequency com po nents of a re ceived sig nal to

re verse the preemphasis that was placed on them to over come at ten u a tion and noise
in the trans mis sion pro cess.

de fect An er ror made dur ing ini tial plan ning that is nor mally de tected and cor rected

dur ing the de vel op ment phase. Note that a

fault is an er ror that oc curs in an

in-service sys tem.

de flec tion The con trol placed on elec tron di rec tion and mo tion in CRTs and other vac -

uum tube de vices by vary ing the strengths of elec tro static (elec tri cal) or elec tro mag -
netic fields.

deg ra da tion In sus cep ti bil ity test ing, any un de sir able change in the op er a tional per -

for mance of a test spec i men. This term does not nec es sar ily mean mal func tion or
cat a strophic fail ure.

deg ra da tion fail ure A fail ure that re sults from a grad ual change in per for mance char -

ac ter is tics of a sys tem or part with time.

de lay The amount of time by which a sig nal is de layed or an event is re tarded.

de lay cir cuit A cir cuit de signed to de lay a sig nal pass ing through it by a spec i fied

amount.

de lay dis tor tion The dis tor tion re sult ing from the dif fer ence in phase de lays at two fre -

quen cies of in ter est.

de lay equal izer A net work that ad justs the ve loc ity of prop a ga tion of the fre quency

com po nents of a com plex sig nal to coun ter act the de lay dis tor tion char ac ter is tics of
a trans mis sion chan nel.

de lay line A trans mis sion net work that in creases the prop a ga tion time of a sig nal trav -

el ing through it.

delta con nec tion A com mon method of join ing to gether a three-phase power sup ply,

with each phase across a dif fer ent pair of the three wires used.

delta-connected sys tem A 3-phase power dis tri bu tion sys tem where a sin gle-phase

out put can be de rived from each of the ad ja cent pairs of an equi lat eral tri an gle
formed by the ser vice drop trans former sec ond ary wind ings.

de mod u lator Any de vice that re cov ers the orig i nal sig nal af ter it has mod u lated a

high-frequency car rier. The out put from the unit may be in base band com pos ite
form.

demultiplexer (demux) A de vice used to sep a rate two or more sig nals that were pre vi -

ously com bined by a com pat i ble multi plexer and are trans mit ted over a sin gle chan -
nel.

de rat ing fac tor An op er at ing safety mar gin pro vided for a com po nent or sys tem to en -

sure re li able per for mance. A

de rat ing al low ance also is typ i cally pro vided for op er -

a tion un der ex treme en vi ron men tal con di tions, or un der strin gent re li abil ity re -
quire ments.

desiccant A dry ing agent used for dry ing out ca ble splices or sen si tive equip ment.

de sign A lay out of all the nec es sary equip ment and fa cil i ties re quired to make a spe cial

cir cuit, piece of equip ment, or sys tem work.

de sign ob jec tive The de sired elec tri cal or me chan i cal per for mance char ac ter is tic for

elec tronic cir cuits and equip ment.

de tec tion The rec ti fi ca tion pro cess that re sults in the mod u lat ing sig nal be ing sep a -

rated from a mod u lated wave.

detectivity The re cip ro cal of noise equiv a lent power.

de tec tor A de vice that con verts one type of en ergy into an other.

de vice A func tional cir cuit, com po nent, or net work unit, such as a vac uum tube or tran -

sis tor.

dew point The tem per a ture at which mois ture will con dense out.

di ag no sis The pro cess of lo cat ing er rors in soft ware, or equip ment faults in hard ware.

di ag nos tic rou tine A soft ware pro gram de signed to trace er rors in soft ware, lo cate

hard ware faults, or iden tify the cause of a break down.

di elec tric An in su lat ing ma te rial that sep a rates the el e ments of var i ous com po nents,

in clud ing ca pac i tors and trans mis sion lines. Di elec tric ma te ri als in clude air, plas tic,
mica, ce ramic, and Tef lon. A di elec tric ma te rial must be an in su la tor. (

Tef lon is a

reg is tered trade mark of Du Pont.)

di elec tric con stant The ra tio of the ca pac i tance of a ca pac i tor with a cer tain di elec tric

ma te rial to the ca pac i tance with a vac uum as the di elec tric. The

di elec tric con stant

Glossary 629

is con sid ered a mea sure of the ca pa bil ity of a di elec tric ma te rial to store an elec tro -
static charge.

di elec tric strength The po ten tial gra di ent at which elec trical break down oc curs.

dif fer en tial am pli fier An in put cir cuit that re jects volt ages that are the same at both in -

put ter mi nals but am pli fies any volt age dif fer ence be tween the in puts. Use of a dif -
fer en tial am pli fier causes any sig nal pres ent on both ter mi nals, such as com mon
mode hum, to can cel it self.

dif fer en tial dc The max i mum dc volt age that can be ap plied be tween the dif fer en tial

in puts of an am pli fier while main tain ing lin ear op er a tion.

dif fer en tial gain The dif fer ence in out put am pli tude (ex pressed in per cent or dB) of a

small high fre quency sinewave sig nal at two stated lev els of a low fre quency sig nal
on which it is su per im posed.

dif fer en tial phase The dif fer ence in out put phase of a small high fre quency sinewave

sig nal at two stated lev els of a low fre quency sig nal on which it is su per im posed.

dif fer en tial-mode in ter fer ence An in ter fer ence source that causes a change in po ten -

tial of one side of a sig nal trans mis sion path rel a tive to the other side.

dif fuse re flec tion The scat ter ing ef fect that oc curs when light, ra dio, or sound waves

strike a rough sur face.

dif fu sion The spread ing or scat ter ing of a wave, such as a ra dio wave.

di ode A semi con duc tor or vac uum tube with two elec trodes that passes elec tric cur rent

in one di rec tion only. Di odes are used in rec ti fi ers, gates, mod u la tors, and de tec tors.

di rect cou pling A cou pling method be tween stages that per mits dc cur rent to flow be -

tween the stages.

di rect cur rent An elec tri cal sig nal in which the di rec tion of cur rent flow re mains con -

stant.

dis charge The con ver sion of stored en ergy, as in a bat tery or ca pac i tor, into an elec tric

cur rent.

dis con ti nu ity An abrupt non uni form point of change in a trans mis sion cir cuit that

causes a dis rup tion of nor mal op er a tion.

dis crete An in di vid ual cir cuit com po nent.
dis crete com po nent A sep a rately con tained cir cuit el e ment with its own ex ter nal con -

nec tions.

discriminator A de vice or cir cuit whose out put am pli tude and po lar ity vary ac cord ing

to how much the in put sig nal var ies from a stan dard or from an other sig nal. A
discriminator can be used to re cover the mod u lat ing wave form in a fre quency mod -
u lated sig nal.

dish An an tenna sys tem con sist ing of a par a bolic shaped re flec tor with a sig nal feed el -

e ment at the fo cal point. Dish an ten nas com monly are used for trans mis sion and re -
cep tion from mi cro wave sta tions and com mu ni ca tions sat el lites.

dis per sion The wave length de pend ence of a pa ram e ter.

dis play The rep re sen ta tion of text and im ages on a cath ode-ray tube, an ar ray of

light-emitting di odes, a liq uid-crystal read out, or an other sim i lar de vice.

dis play de vice An out put unit that pro vides a vi sual rep re sen ta tion of data.

dis tor tion The dif fer ence be tween the wave shape of an orig i nal sig nal and the sig nal

af ter it has tra versed a trans mis sion cir cuit.

dis tor tion-limited op er a tion The con di tion pre vail ing when the shape of the sig nal,

rather than the am pli tude (or power), is the lim it ing fac tor in com mu ni ca tion ca pa -
bil ity. This con di tion is reached when the sys tem dis torts the shape of the wave form

be yond tol er a ble lim its. For lin ear sys tems,

dis tor tion-limited op er a tion is equiv a -

lent to

band width-limited op er a tion.

dis tur bance The in ter fer ence with nor mal con di tions and com mu ni ca tions by some

ex ter nal en ergy source.

dis tur bance cur rent The un wanted cur rent of any ir reg u lar phe nom e non as so ci ated

with trans mis sion that tends to limit or in ter fere with the in ter change of in for ma tion.

dis tur bance power The un wanted power of any ir reg u lar phe nom e non as so ci ated

with trans mis sion that tends to limit or in ter fere with the in ter change of in for ma tion.

dis tur bance volt age The un wanted volt age of any ir reg u lar phe nom e non as so ci ated

with trans mis sion that tends to limit or in ter fere with the in ter change of in for ma tion.

di ver sity re ceiver A re ceiver us ing two an ten nas con nected through cir cuitry that

senses which an tenna is re ceiv ing the stron ger sig nal. Elec tronic gating per mits the
stron ger source to be routed to the re ceiv ing sys tem.

doc u men ta tion A writ ten de scrip tion of a pro gram. Doc u men ta tion can be con sid ered

as any re cord that has per ma nence and can be read by hu mans or ma chines.

down-lead A lead-in wire from an an tenna to a re ceiver.
downlink The por tion of a com mu ni ca tion link used for trans mis sion of sig nals from a

sat el lite or air borne plat form to a sur face ter mi nal.

down stream A spec i fied sig nal mod i fi ca tion oc cur ring af ter other given de vices in a

sig nal path.

down time The time dur ing which equip ment is not ca pa ble of do ing use ful work be -

cause of mal func tion. This does not in clude pre ven tive main te nance time. In other
words,

down time is mea sured from the oc cur rence of a mal func tion to the cor rec tion

of that mal func tion.

drift A slow change in a nom i nally con stant sig nal char ac ter is tic, such as fre quency.

drift-space The area in a kly stron tube in which elec trons drift at their en ter ing ve loc i -

ties and form elec tron

bunches.

drive The in put sig nal to a cir cuit, par tic u larly to an am pli fier.

driver An elec tronic cir cuit that sup plies an iso lated out put to drive the in put of an other

cir cuit.

drop-out value The value of cur rent or volt age at which a re lay will cease to be op er -

ated.

drop out The mo men tary loss of a sig nal.

drop ping re sis tor A re sis tor de signed to carry cur rent that will make a re quired

voltage avail able.

du plex sep a ra tion The fre quency spac ing re quired in a com mu ni ca tions sys tem be -

tween the

for ward and re turn chan nels to main tain in ter fer ence at an ac cept ably low

level.

du plex sig nal ing A con fig u ra tion per mit ting sig nal ing in both trans mis sion di rec tions

si mul ta neously.

duty cy cle The ra tio of op er at ing time to to tal elapsed time of a de vice that op er ates in -

ter mit tently, ex pressed in per cent.

dy namic A sit u a tion in which the op er at ing pa ram e ters and/or re quire ments of a given

sys tem are con tin u ally chang ing.

dy namic range The max i mum range or ex tremes in am pli tude, from the low est to the

high est (noise floor to sys tem clip ping), that a sys tem is ca pa ble of re pro duc ing. The
dy namic range is ex pressed in dB against a ref er ence level.

dy namo A ro tat ing ma chine, nor mally a dc gen er a tor.
dynamotor A ro tat ing ma chine used to con vert dc into ac.

E

earth A large con duct ing body with no elec tri cal po ten tial, also called ground.

earth ca pac i tance The ca pac i tance be tween a given cir cuit or com po nent and a point

at ground po ten tial.

earth cur rent A cur rent that flows to earth/ground, es pe cially one that fol lows from a

fault in the sys tem.

Earth cur rent may also re fer to a cur rent that flows in the earth,

re sult ing from ion o spheric dis tur bances, light ning, or faults on power lines.

earth fault A fault that oc curs when a con duc tor is ac ci den tally grounded/earthed, or

when the re sis tance to earth of an in su la tor falls be low a spec i fied value.

earth ground A large con duct ing body that rep re sents zero level in the scale of elec tri -

cal po ten tial. An

earth ground is a con nec tion made ei ther ac ci den tally or by de sign

be tween a con duc tor and earth.

earth po ten tial The po ten tial taken to be the ar bi trary zero in a scale of elec tric po ten -

tial.

ef fec tive ground A con nec tion to ground through a me dium of suf fi ciently low im ped -

ance and ad e quate cur rent-carrying ca pac ity to pre vent the buildup of volt ages that
might be haz ard ous to equip ment or per son nel.

ef fec tive re sis tance The in creased re sis tance of a con duc tor to an al ter nat ing cur rent

re sult ing from the

skin ef fect, rel a tive to the di rect-current re sis tance of the con duc -

tor. Higher fre quen cies tend to travel only on the outer skin of the con duc tor,
whereas dc flows uni formly through the en tire area.

ef fi ciency The use ful power out put of an elec tri cal de vice or cir cuit di vided by the to tal

power in put, ex pressed in per cent.

elec tric Any de vice or cir cuit that pro duces, op er ates on, trans mits, or uses elec tric ity.

elec tric charge An ex cess of ei ther elec trons or pro tons within a given space or ma te -

rial.

elec tric field strength The mag ni tude, mea sured in volts per me ter, of the elec tric field

in an elec tro mag netic wave.

elec tric flux The amount of elec tric charge, mea sured in cou lombs, across a di elec tric

of spec i fied area.

Elec tric flux may also re fer sim ply to elec tric lines of force.

elec tric ity An en ergy force de rived from the move ment of neg a tive and pos i tive elec -

tric charges.

elec trode An elec tri cal ter mi nal that emits, col lects, or con trols an elec tric cur rent.

elec trol y sis A chem i cal change in duced in a sub stance re sult ing from the pas sage of

elec tric cur rent through an elec tro lyte.

elec tro lyte A non me tal lic con duc tor of elec tric ity in which cur rent is car ried by the

phys i cal move ment of ions.

elec tro mag net An iron or steel core sur rounded by a wire coil. The core be comes mag -

ne tized when cur rent flows through the coil but loses its mag ne tism when the cur -
rent flow is stopped.

elec tro mag netic com pat i bil ity The ca pa bil ity of elec tronic equip ment or sys tems to

op er ate in a spe cific elec tro mag netic en vi ron ment, at des ig nated lev els of ef fi ciency
and within a de fined mar gin of safety, with out in ter fer ing with it self or other sys -
tems.

elec tro mag netic field The elec tric and mag netic fields as so ci ated with ra dio and light

waves.

elec tro mag netic in duc tion An elec tro mo tive force cre ated with a con duc tor by the

rel a tive mo tion be tween the con duc tor and a nearby mag netic field.

elec tro mag ne tism The study of phe nom ena as so ci ated with vary ing mag netic fields,

elec tro mag netic ra di a tion, and mov ing elec tric charges.

elec tro mo tive force (EMF) An elec tri cal po ten tial, mea sured in volts, that can pro -

duce the move ment of elec tri cal charges.

elec tron A sta ble el e men tary par ti cle with a neg a tive charge that is mainly re spon si ble

for elec tri cal con duc tion. Elec trons move when un der the in flu ence of an elec tric
field. This move ment con sti tutes an

elec tric cur rent.

elec tron beam A stream of emit ted elec trons, usu ally in a vac uum.

elec tron gun A hot cath ode that pro duces a finely fo cused stream of fast elec trons,

which are nec es sary for the op er a tion of a vac uum tube, such as a cath ode ray tube.
The gun is made up of a hot cath ode elec tron source, a con trol grid, ac cel er at ing an -
odes, and (usu ally) fo cus ing elec trodes.

elec tron lens A de vice used for fo cus ing an elec tron beam in a cath ode ray tube. Such

fo cus ing can be ac com plished by ei ther mag netic forces, in which ex ter nal coils are
used to cre ate the proper mag netic field within the tube, or elec tro static forces,
where me tal lic plates within the tube are charged elec tri cally in such a way as to
con trol the move ment of elec trons in the beam.

elec tron volt The en ergy ac quired by an elec tron in pass ing through a po ten tial dif fer -

ence of one volt in a vac uum.

elec tronic A de scrip tion of de vices (or sys tems) that are de pend ent on the flow of elec -

trons in elec tron tubes, semi con duc tors, and other de vices, and not solely on elec -
tron flow in or di nary wires, inductors, ca pac i tors, and sim i lar pas sive com po nents.

Elec tronic In dus tries As so ci a tion (EIA) A trade or ga ni za tion, based in Wash ing ton,

DC, rep re sent ing the man u fac tur ers of elec tronic sys tems and parts, in clud ing com -
mu ni ca tions sys tems. The as so ci a tion de vel ops stan dards for elec tronic com po -
nents and sys tems.

elec tronic switch A tran sis tor, semi con duc tor di ode, or a vac uum tube used as an

on/off switch in an elec tri cal cir cuit. Elec tronic switches can be con trolled man u -
ally, by other cir cuits, or by com put ers.

elec tron ics The field of sci ence and en gi neer ing that deals with elec tron de vices and

their uti li za tion.

elec tro plate The pro cess of coat ing a given ma te rial with a de posit of metal by elec tro -

lytic ac tion.

elec tro static The con di tion per tain ing to elec tric charges that are at rest.
elec tro static field The space in which there is elec tric stress pro duced by static elec tric

charges.

elec tro static in duc tion The pro cess of in duc ing static elec tric charges on a body by

bring ing it near other bod ies that carry high elec tro static charges.

el e ment A sub stance that con sists of at oms of the same atomic num ber. El e ments are

the ba sic units in all chem i cal changes other than those in which

atomic changes,

such as fu sion and fis sion, are in volved.

EMI (elec tro mag netic in ter fer ence) Un de sir able elec tro mag netic waves that are ra -

di ated un in ten tion ally from an elec tronic cir cuit or de vice into other cir cuits or de -
vices, dis rupt ing their op er a tion.

emis sion (1--ra di a tion) The ra di a tion pro duced, or the pro duc tion of ra di a tion by a

ra dio trans mit ting sys tem. The emis sion is con sid ered to be a

sin gle emis sion if the

mod u lat ing sig nal and other char ac ter is tics are the same for ev ery trans mit ter of the
ra dio trans mit ting sys tem and the spac ing be tween an ten nas is not more than a few
wave lengths.

(2--cath ode) The re lease of elec trons from the cath ode of a vac uum

tube.

(3--par a sitic) A spu ri ous ra dio fre quency emis sion un in ten tion ally gen er -

ated at fre quen cies that are in de pend ent of the car rier fre quency be ing am pli fied or
mod u lated.

(4--sec ond ary) In an elec tron tube, emis sion of elec trons by a plate or

grid be cause of bom bard ment by

pri mary emis sion elec trons from the cath ode of the

tube.

(5--spu ri ous) An emis sion out side the ra dio fre quency band au tho rized for a

trans mit ter.

(6--thermonic) An emis sion from a cath ode re sult ing from high tem -

per a ture.

em pha sis The in ten tional al ter ation of the fre quency-amplitude char ac ter is tics of a

sig nal to re duce the ad verse ef fects of noise in a com mu ni ca tion sys tem.

em pir i cal A con clu sion not based on pure the ory, but on prac ti cal and ex per i men tal

work.

em u la tion The use of one sys tem to im i tate the ca pa bil i ties of an other sys tem.

en able To pre pare a cir cuit for op er a tion or to al low an item to func tion.
en abling sig nal A sig nal that per mits the oc cur rence of a spec i fied event.

en code The con ver sion of in for ma tion from one form into an other to ob tain char ac ter -

is tics re quired by a trans mis sion or stor age sys tem.

en coder A de vice that pro cesses one or more in put sig nals into a spec i fied form for

trans mis sion and/or stor age.

en er gized The con di tion when a cir cuit is switched on, or pow ered up.

en ergy spec tral den sity A fre quency-domain de scrip tion of the en ergy in each of the

fre quency com po nents of a pulse.

en ve lope The bound ary of the fam ily of curves ob tained by vary ing a pa ram e ter of a

wave.

en ve lope de lay The dif fer ence in ab so lute de lay be tween the fast est and slow est prop a -

gat ing fre quen cies within a spec i fied band width.

en ve lope de lay dis tor tion The max i mum dif fer ence or de vi a tion of the en ve -

lope-delay char ac ter is tic be tween any two spec i fied fre quen cies.

en ve lope de tec tion A de mod u la tion pro cess that senses the shape of the mod u lated RF

en ve lope. A di ode de tec tor is one type of en velop de tec tion de vice.

en vi ron men tal An equip ment spec i fi ca tion cat e gory re lat ing to tem per a ture and hu -

mid ity.

EQ (equal iza tion) net work A net work con nected to a cir cuit to cor rect or con trol its

trans mis sion fre quency char ac ter is tics.

equal iza tion (EQ) The re duc tion of fre quency dis tor tion and/or phase dis tor tion of a

cir cuit through the in tro duc tion of one or more net works to com pen sate for the dif -
fer ence in at ten u a tion, time de lay, or both, at the var i ous fre quen cies in the trans mis -
sion band.

equal ize The pro cess of in sert ing in a line a net work with com ple men tary trans mis sion

char ac ter is tics to those of the line, so that when the loss or de lay in the line and that
in the equal izer are com bined, the over all loss or de lay is ap prox i mately equal at all
fre quen cies.

equal izer A net work that cor rects the trans mis sion-frequency char ac ter is tics of a cir -

cuit to al low it to trans mit se lected fre quen cies in a uni form man ner.

equa to rial or bit The plane of a sat el lite or bit which co in cides with that of the equa tor

of the pri mary body.

equip ment A gen eral term for elec tri cal ap pa ra tus and hard ware, switch ing sys tems,

and trans mis sion com po nents.

equip ment fail ure The con di tion when a hard ware fault stops the suc cess ful com ple -

tion of a task.

equip ment ground A pro tec tive ground con sist ing of a con duct ing path to ground of

noncurrent car ry ing metal parts.

equiv a lent cir cuit A sim pli fied net work that em u lates the char ac ter is tics of the real

cir cuit it re places. An equiv a lent cir cuit is typ i cally used for math e mat i cal anal y sis.

equiv a lent noise re sis tance A quan ti ta tive rep re sen ta tion in re sis tance units of the

spec tral den sity of a noise volt age gen er a tor at a spec i fied fre quency.

er ror A col lec tive term that in cludes all types of in con sis ten cies, trans mis sion de vi a -

tions, and con trol fail ures.

ex ci ta tion The cur rent that en er gizes field coils in a gen er a tor.
expandor A de vice with a non lin ear gain char ac ter is tic that acts to in crease the gain

more on larger in put sig nals than it does on smaller in put sig nals.

ex tremely high fre quency (EHF) The band of mi cro wave fre quen cies be tween the

lim its of 30 GHz and 300 GHz (wave lengths be tween 1 cm and 1 mm).

ex tremely low fre quency The ra dio sig nals with op er at ing fre quen cies be low 300 Hz

(wave lengths lon ger than 1000 km).

F

fail-safe op er a tion A type of con trol ar chi tec ture for a sys tem that pre vents im proper

func tion ing in the event of cir cuit or op er a tor fail ure.

fail ure A de tected ces sa tion of abil ity to per form a spec i fied func tion or func tions

within pre vi ously es tab lished lim its. A

fail ure is be yond ad just ment by the op er a tor

by means of con trols nor mally ac ces si ble dur ing rou tine op er a tion of the sys tem.
(This re quires that mea sur able lim its be es tab lished to de fine "sat is fac tory per for -
mance".)

fail ure ef fect The re sult of the mal func tion or fail ure of a de vice or com po nent.

fail ure in time (FIT) A unit value that in di cates the re li abil ity of a com po nent or de -

vice. One fail ure in time cor re sponds to a fail ure rate of 10

9

per hour.

fail ure mode and ef fects anal y sis (FMEA) An it er a tive doc u mented pro cess per -

formed to iden tify ba sic faults at the com po nent level and de ter mine their ef fects at
higher lev els of as sem bly.

fail ure rate The ra tio of the num ber of ac tual fail ures to the num ber of times each item

has been sub jected to a set of spec i fied stress con di tions.

fall time The length of time dur ing which a pulse de creases from 90 per cent to 10 per -

cent of its max i mum am pli tude.

farad The stan dard unit of ca pac i tance equal to the value of a ca pac i tor with a po ten tial

of one volt be tween its plates when the charge on one plate is one cou lomb and there
is an equal and op po site charge on the other plate. The farad is a large value and is
more com monly ex pressed in

microfarads or picofarads. The farad is named for the

Eng lish chem ist and phys i cist Mi chael Far a day (1791-1867).

fast fre quency shift key ing (FFSK) A sys tem of dig i tal mod u la tion where the dig its

are rep re sented by dif fer ent fre quen cies that are re lated to the baud rate, and where
tran si tions oc cur at the zero cross ings.

fa tigue The re duc tion in strength of a metal caused by the for ma tion of crys tals re sult -

ing from re peated flex ing of the part in ques tion.

fault A con di tion that causes a de vice, a com po nent, or an el e ment to fail to per form in

a re quired man ner. Ex am ples in clude a short-circuit, bro ken wire, or in ter mit tent
con nec tion.

fault to ground A fault caused by the fail ure of in su la tion and the con se quent es tab -

lish ment of a di rect path to ground from a part of the cir cuit that should not nor mally
be grounded.

fault tree anal y sis (FTA) An it er a tive doc u mented pro cess of a sys tem atic na ture per -

formed to iden tify ba sic faults, de ter mine their causes and ef fects, and es tab lish
their prob a bil i ties of oc cur rence.

fea ture A dis tinc tive char ac ter is tic or part of a sys tem or piece of equip ment, usu ally

vis i ble to end us ers and de signed for their con ve nience.

Fed eral Com mu ni ca tions Com mis sion (FCC) The fed eral agency em pow ered by

law to reg u late all in ter state ra dio and wire line com mu ni ca tions ser vices orig i nat ing
in the United States, in clud ing ra dio, tele vi sion, fac sim ile, tele graph, data trans mis -
sion, and tele phone sys tems. The agency was es tab lished by the Com mu ni ca tions
Act of 1934.

feed back The re turn of a por tion of the out put of a de vice to the in put. Pos i tive feed -

back adds to the in put, neg a tive feed back sub tracts from the in put.

feed back am pli fier An am pli fier with the com po nents re quired to feed a por tion of the

out put back into the in put to al ter the char ac ter is tics of the out put sig nal.

feedline A trans mis sion line, typ i cally co ax ial ca ble, that con nects a high fre quency

en ergy source to its load.

femto A pre fix mean ing one qua dril lionth (10

15

).

fer rite A ce ramic ma te rial made of pow dered and com pressed fer ric ox ide, plus other

ox ides (mainly co balt, nickel, zinc, yt trium-iron, and man ga nese). These ma te ri als
have low eddy cur rent losses at high fre quen cies.

fer ro mag netic ma te rial A ma te rial with low rel a tive per me abil ity and high co er cive

force so that it is dif fi cult to mag ne tize and de mag ne tize. Hard fer ro mag netic ma te -
ri als re tain mag ne tism well, and are com monly used in per ma nent mag nets.

fi del ity The de gree to which a sys tem, or a por tion of a sys tem, ac cu rately re pro duces

at its out put the es sen tial char ac ter is tics of the sig nal im pressed upon its in put.

field strength The strength of an elec tric, mag netic, or elec tro mag netic field.

fil a ment A wire that be comes hot when cur rent is passed through it, used ei ther to emit

light (for a light bulb) or to heat a cath ode to en able it to emit elec trons (for an elec -
tron tube).

film re sis tor A type of re sis tor made by de pos it ing a thin layer of re sis tive ma te rial on

an in su lat ing core.

fil ter A net work that passes de sired fre quen cies but greatly at tenu ates other fre quen -

cies.

fil tered noise White noise that has been passed through a fil ter. The power spec tral

den sity of fil tered white noise has the same shape as the trans fer func tion of the fil -
ter.

fit ting A cou pling or other me chan i cal de vice that joins one com po nent with an other.

fixed A sys tem or de vice that is not change able or mov able.
flashover An arc or spark be tween two con duc tors.

flashover volt age The volt age be tween con duc tors at which flashover just oc curs.

flat face tube The de sign of CRT tube with al most a flat face, giv ing im proved leg i bil -

ity of text and re duced re flec tion of am bi ent light.

flat level A sig nal that has an equal am pli tude re sponse for all fre quen cies within a

stated range.

flat loss A cir cuit, de vice, or chan nel that at tenu ates all fre quen cies of in ter est by the

same amount, also called

flat slope.

flat noise A noise whose power per unit of fre quency is es sen tially in de pend ent of fre -

quency over a spec i fied fre quency range.

flat re sponse The per for mance pa ram e ter of a sys tem in which the out put sig nal am pli -

tude of the sys tem is a faith ful re pro duc tion of the in put am pli tude over some range
of spec i fied in put fre quen cies.

float ing A cir cuit or de vice that is not con nected to any source of po ten tial or to ground.

flu o res cence The char ac ter is tic of a ma te rial to pro duce light when ex cited by an ex ter -

nal en ergy source. Min i mal or no heat re sults from the pro cess.

flux The elec tric or mag netic lines of force re sult ing from an ap plied en ergy source.
fly wheel ef fect The char ac ter is tic of an os cil la tor that en ables it to sus tain os cil la tions

af ter re moval of the con trol stim u lus. This char ac ter is tic may be de sir able, as in the
case of a phase-locked loop em ployed in a syn chro nous sys tem, or un de sir able, as in
the case of a volt age-controlled os cil la tor.

fo cus ing A method of mak ing beams of ra di a tion con verge on a tar get, such as the face

of a CRT.

Fou rier anal y sis A math e mat i cal pro cess for trans form ing val ues be tween the fre -

quency do main and the time do main. This term also re fers to the de com po si tion of a
time-domain sig nal into its fre quency com po nents.

Fou rier trans form An in te gral that per forms an ac tual trans for ma tion be tween the

fre quency do main and the time do main in Fou rier anal y sis.

frame A seg ment of an an a log or dig i tal sig nal that has a re pet i tive char ac ter is tic, in

that cor re spond ing el e ments of suc ces sive

frames rep re sent the same things.

free elec tron An elec tron that is not at tached to an atom and is, thus, mo bile when an

elec tro mo tive force is ap plied.

free run ning An os cil la tor that is not con trolled by an ex ter nal syn chro niz ing sig nal.

free-running os cil la tor An os cil la tor that is not syn chro nized with an ex ter nal tim ing

source.

fre quency The num ber of com plete cy cles of a pe ri odic wave form that oc cur within a

given length of time. Fre quency is usu ally spec i fied in cy cles per sec ond (

Hertz).

Fre quency is the re cip ro cal of wave length. The higher the fre quency, the shorter the
wave length. In gen eral, the higher the fre quency of a sig nal, the more ca pac ity it has
to carry in for ma tion, the smaller an an tenna is re quired, and the more sus cep ti ble the
sig nal is to ab sorp tion by the at mo sphere and by phys i cal struc tures. At mi cro wave
fre quen cies, ra dio sig nals take on a

line-of-sight char ac ter is tic and re quire highly di -

rec tional and fo cused an ten nas to be used suc cess fully.

fre quency ac cu racy The de gree of con for mity of a given sig nal to the spec i fied value

of a fre quency.

fre quency al lo ca tion The des ig na tion of ra dio-frequency bands for use by spe cific ra -

dio ser vices.

fre quency con tent The band of fre quen cies or spe cific fre quency com po nents con -

tained in a sig nal.

fre quency con verter A cir cuit or de vice used to change a sig nal of one fre quency into

an other of a dif fer ent fre quency.

fre quency co or di na tion The pro cess of an a lyz ing fre quen cies in use in var i ous bands

of the spec trum to achieve re li able per for mance for cur rent and new ser vices.

fre quency coun ter An in stru ment or test set used to mea sure the fre quency of a ra dio

sig nal or any other al ter nat ing wave form.

fre quency de par ture An un in ten tional de vi a tion from the nom i nal fre quency value.

fre quency dif fer ence The al ge braic dif fer ence be tween two fre quen cies. The two fre -

quen cies can be of iden ti cal or dif fer ent nom i nal val ues.

fre quency dis place ment The end-to-end shift in fre quency that may re sult from in de -

pend ent fre quency trans la tion er rors in a cir cuit.

fre quency dis tor tion The dis tor tion of a multifrequency sig nal caused by un equal at -

ten u a tion or am pli fi ca tion at the dif fer ent fre quen cies of the sig nal. This term may
also be re ferred to as

am pli tude dis tor tion.

fre quency do main A rep re sen ta tion of sig nals as a func tion of fre quency, rather than

of time.

fre quency mod u la tion (FM) The mod u la tion of a car rier sig nal so that its in stan ta -

neous fre quency is pro por tional to the in stan ta neous value of the mod u lat ing wave.

fre quency mul ti plier A cir cuit that pro vides as an out put an ex act mul ti ple of the in put

fre quency.

fre quency off set A fre quency shift that oc curs when a sig nal is sent over an an a log

trans mis sion fa cil ity in which the mod u lat ing and de mod u lat ing fre quen cies are not
iden ti cal. A chan nel with fre quency off set does not pre serve the wave form of a
trans mit ted sig nal.

fre quency re sponse The mea sure of sys tem lin ear ity in re pro duc ing sig nals across a

spec i fied band width. Fre quency re sponse is ex pressed as a fre quency range with a
spec i fied am pli tude tol er ance in dB.

fre quency re sponse char ac ter is tic The vari a tion in the trans mis sion per for mance

(gain or loss) of a sys tem with re spect to vari a tions in fre quency.

fre quency re use A tech nique used to ex pand the ca pac ity of a given set of fre quen cies

or chan nels by sep a rat ing the sig nals ei ther geo graph i cally or through the use of dif -
fer ent po lar iza tion tech niques. Fre quency re use is a com mon el e ment of the

fre -

quency co or di na tion pro cess.

fre quency se lec tiv ity The abil ity of equip ment to sep a rate or dif fer en ti ate be tween

sig nals at dif fer ent fre quen cies.

fre quency shift The dif fer ence be tween the fre quency of a sig nal ap plied at the in put

of a cir cuit and the fre quency of that sig nal at the out put.

fre quency shift key ing (FSK) A com monly-used method of dig i tal mod u la tion in

which a one and a zero (the two pos si ble states) are each trans mit ted as sep a rate fre -
quen cies.

fre quency sta bil ity A mea sure of the vari a tions of the fre quency of an os cil la tor from

its mean fre quency over a spec i fied pe riod of time.

fre quency stan dard An os cil la tor with an out put fre quency suf fi ciently sta ble and ac -

cu rate that it is used as a ref er ence.

fre quency-division mul ti ple ac cess (FDMA) The pro vi sion of mul ti ple ac cess to a

trans mis sion fa cil ity, such as an earth sat el lite, by as sign ing each trans mit ter its own
fre quency band.

fre quency-division multiplexing (FDM) The pro cess of trans mit ting mul ti ple an a log

sig nals by an or derly as sign ment of fre quency slots, that is, by di vid ing trans mis sion
band width into sev eral nar row bands, each of which car ries a sin gle com mu ni ca tion
and is sent si mul ta neously with oth ers over a com mon trans mis sion path.

full du plex A com mu ni ca tions sys tem ca pa ble of trans mis sion si mul ta neously in two

di rec tions.

full-wave rec ti fier A cir cuit con fig u ra tion in which both pos i tive and neg a tive

half-cycles of the in com ing ac sig nal are rec ti fied to pro duce a uni di rec tional (dc)
cur rent through the load.

func tional block di a gram A di a gram il lus trat ing the def i ni tion of a de vice, sys tem, or

prob lem on a log i cal and func tional ba sis.

func tional unit An en tity of hard ware and/or soft ware ca pa ble of ac com plish ing a

given pur pose.

fun da men tal fre quency The low est fre quency com po nent of a com plex sig nal.
fuse A pro tec tive de vice used to limit cur rent flow in a cir cuit to a spec i fied level. The

fuse con sists of a me tal lic link that melts and opens the cir cuit at a spec i fied cur rent
level.

fuse wire A fine-gauge wire made of an al loy that over heats and melts at the rel a tively

low tem per a tures pro duced when the wire car ries over load cur rents. When used in a
fuse, the wire is called a fuse (or fus ible) link.

G

gain An in crease or de crease in the level of an elec tri cal sig nal. Gain is mea sured in

terms of deci bels or num ber-of-times of mag ni fi ca tion. Strictly speak ing,

gain re -

fers to an in crease in level. Neg a tive num bers, how ever, are com monly used to de -
note a de crease in level.

gain-bandwidth The gain times the fre quency of mea sure ment when a de vice is bi -

ased for max i mum ob tain able gain.

gain/fre quency char ac ter is tic The gain-versus-frequency char ac ter is tic of a chan nel

over the band width pro vided, also re ferred to as

fre quency re sponse.

gain/fre quency dis tor tion A cir cuit de fect in which a change in fre quency causes a

change in sig nal am pli tude.

gal vanic A de vice that pro duces di rect cur rent by chem i cal ac tion.

gang The me chan i cal con nec tion of two or more cir cuit de vices so that they can all be

ad justed si mul ta neously.

gang ca pac i tor A vari able ca pac i tor with more than one set of mov ing plates linked to -

gether.

gang tun ing The si mul ta neous tun ing of sev eral dif fer ent cir cuits by turn ing a sin gle

shaft on which ganged ca pac i tors are mounted.

ganged One or more de vices that are me chan i cally cou pled, nor mally through the use

of a shared shaft.

gas break down The ion iza tion of a gas be tween two elec trodes caused by the ap pli ca -

tion of a volt age that ex ceeds a thresh old value. The ion ized path has a low im ped -
ance. Cer tain types of cir cuit and line pro tec tors rely on gas break down to di vert
haz ard ous cur rents away from pro tected equip ment.

gas tube A pro tec tion de vice in which a suf fi cient volt age across two elec trodes causes

a gas to ion ize, cre at ing a low im ped ance path for the dis charge of dan ger ous volt -
ages.

gas-discharge tube A gas-filled tube de signed to carry cur rent dur ing gas break down.

The gas-discharge tube is com monly used as a pro tec tive de vice, pre vent ing high
volt ages from dam ag ing sen si tive equip ment.

gauge A mea sure of wire di am e ter. In mea sur ing wire gauge, the lower the num ber, the

thicker the wire.

Gaussi an dis tri bu tion A sta tis ti cal dis tri bu tion, also called the nor mal dis tri bu tion.

The graph of a Gaussi an dis tri bu tion is a bell-shaped curve.

Gaussi an noise Noise in which the dis tri bu tion of am pli tude fol lows a Gaussi an

model, that is, the noise is ran dom but dis trib uted about a ref er ence volt age of zero.

Gaussi an pulse A pulse that has the same form as its own Fou rier trans form.

gen er a tor A ma chine that con verts me chan i cal en ergy into elec tri cal en ergy, or one

form of elec tri cal en ergy into an other form.

geosynchronous The at trib ute of a sat el lite in which the rel a tive po si tion of the sat el -

lite as viewed from the sur face of a given planet is sta tion ary. For earth, the
geosynchronous po si tion is 22,300 miles above the planet.

get ter A metal used in va por ized form to re move re sid ual gas from in side an elec tron

tube dur ing man u fac ture.

giga A pre fix mean ing one bil lion.
giga hertz (GHz) A mea sure of fre quency equal to one bil lion cy cles per sec ond. Sig -

nals op er at ing above 1 giga hertz are com monly known as

mi cro waves, and be gin to

take on the char ac ter is tics of vis i ble light.

glitch A gen eral term used to de scribe a wide va ri ety of mo men tary sig nal dis con ti nu -

ities.

grace ful deg ra da tion An equip ment fail ure mode in which the sys tem suf fers re duced

ca pa bil ity, but does not fail al to gether.

graticule A fixed pat tern of ref er ence mark ings used with os cil lo scope CRTs to sim -

plify mea sure ments. The graticule may be etched on a trans par ent plate cov er ing the
front of the CRT or, for greater ac cu racy in read ings, may be elec tri cally gen er ated
within the CRT it self.

grid (1--gen eral) A mesh elec trode within an elec tron tube that con trols the flow of

elec trons be tween the cath ode and plate of the tube.

(2--bias) The po ten tial ap plied

to a grid in an elec tron tube to con trol its cen ter op er at ing point.

(3--con trol) The

grid in an elec tron tube to which the in put sig nal is usu ally ap plied.

(4--screen) The

grid in an elec tron tube, typ i cally held at a steady po ten tial, that screens the con trol
grid from changes in an ode po ten tial.

(5--sup pres sor) The grid in an elec tron tube

near the an ode (plate) that sup presses the emis sion of sec ond ary elec trons from the
plate.

ground An elec tri cal con nec tion to earth or to a com mon con duc tor usu ally con nected

to earth.

ground clamp A clamp used to con nect a ground wire to a ground rod or sys tem.

ground loop An un de sir able cir cu lat ing ground cur rent in a cir cuit grounded via mul ti -

ple con nec tions or at mul ti ple points.

ground plane A con duct ing ma te rial at ground po ten tial, phys i cally close to other

equip ment, so that con nec tions may be made readily to ground the equip ment at the
re quired points.

ground po ten tial The point at zero elec tric po ten tial.
ground re turn A con duc tor used as a path for one or more cir cuits back to the ground

plane or cen tral fa cil ity ground point.

ground rod A metal rod driven into the earth and con nected into a mesh of in ter con -

nected rods so as to pro vide a low re sis tance link to ground.

ground win dow A sin gle-point in ter face be tween the in te grated ground plane of a

build ing and an iso lated ground plane.

ground wire A cop per con duc tor used to ex tend a good low-resistance earth ground to

pro tec tive de vices in a fa cil ity.

grounded The con nec tion of a piece of equip ment to earth via a low re sis tance path.

ground ing The act of con nect ing a de vice or cir cuit to ground or to a con duc tor that is

grounded.

group de lay A con di tion where the dif fer ent fre quency el e ments of a given sig nal suf -

fer dif fer ing prop a ga tion de lays through a cir cuit or a sys tem. The de lay at a lower
fre quency is dif fer ent from the de lay at a higher fre quency, re sult ing in a
time-related dis tor tion of the sig nal at the re ceiv ing point.

group de lay time The rate of change of the to tal phase shift of a wave form with an gu lar

fre quency through a de vice or trans mis sion fa cil ity.

group ve loc ity The speed of a pulse on a trans mis sion line.

guard band A nar row band width be tween ad ja cent chan nels in tended to re duce in ter -

fer ence or crosstalk.

H

half-wave rec ti fier A cir cuit or de vice that changes only pos i tive or neg a tive

half-cycle in puts of al ter nat ing cur rent into di rect cur rent.

Hall ef fect The phe nom e non by which a volt age de vel ops be tween the edges of a cur -

rent-carrying metal strip whose faces are per pen dic u lar to an ex ter nal mag netic
field.

hard-wired Elec tri cal de vices con nected through phys i cal wir ing.

harden The pro cess of con struct ing mil i tary tele com mu ni ca tions fa cil i ties so as to

pro tect them from dam age by en emy ac tion, es pe cially

elec tro mag netic pulse

(EMP) ra di a tion.

hard ware Phys i cal equip ment, such as me chan i cal, mag netic, elec tri cal, or elec tronic

de vices or com po nents.

har monic A pe ri odic wave hav ing a fre quency that is an in te gral mul ti ple of the fun da -

men tal fre quency. For ex am ple, a wave with twice the fre quency of the fun da men tal
is called the

sec ond har monic.

har monic an a lyzer A test set ca pa ble of iden ti fy ing the fre quen cies of the in di vid ual

sig nals that make up a com plex wave.

har monic dis tor tion The pro duc tion of har mon ics at the out put of a cir cuit when a pe -

ri odic wave is ap plied to its in put. The level of the dis tor tion is usu ally ex pressed as a
per cent age of the level of the in put.

haz ard A con di tion that could lead to dan ger for op er at ing per son nel.
head room The dif fer ence, in deci bels, be tween the typ i cal op er at ing sig nal level and a

peak over load level.

heat loss The loss of use ful elec tri cal en ergy re sult ing from con ver sion into un wanted

heat.

heat sink A de vice that con ducts heat away from a heat-producing com po nent so that it

stays within a safe work ing tem per a ture range.

heater In an elec tron tube, the fil a ment that heats the cath ode to en able it to emit elec -

trons.

hecto A pre fix mean ing 100.

henry The stan dard unit of elec tri cal in duc tance, equal to the self-inductance of a cir -

cuit or the mu tual in duc tance of two cir cuits when there is an in duced elec tro mo tive
force of one volt and a cur rent change of one am pere per sec ond. The sym bol for in -
duc tance is

H, named for the Amer i can phys i cist Jo seph Henry (1797-1878).

hertz (Hz) The unit of fre quency that is equal to one cy cle per sec ond. Hertz is the re -

cip ro cal of the

pe riod, the in ter val af ter which the same por tion of a pe ri odic wave -

form re curs. Hertz was named for the Ger man phys i cist Hein rich R. Hertz
(1857-1894).

het ero dyne The mix ing of two sig nals in a non lin ear de vice in or der to pro duce two ad -

di tional sig nals at fre quen cies that are the sum and dif fer ence of the orig i nal fre -
quen cies.

het ero dyne fre quency The sum of, or the dif fer ence be tween, two fre quen cies, pro -

duced by com bin ing the two sig nals to gether in a mod u la tor or sim i lar de vice.

het ero dyne wavemeter A test set that uses the het ero dyne prin ci ple to mea sure the fre -

quen cies of in com ing sig nals.

high-frequency loss Loss of sig nal am pli tude at higher fre quen cies through a given

cir cuit or me dium. For ex am ple, high fre quency loss could be caused by pass ing a
sig nal through a co ax ial ca ble.

high Q An in duc tance or ca pac i tance whose ra tio of reactance to re sis tance is high.

high ten sion A high volt age cir cuit.

high-pass fil ter A net work that passes sig nals of higher than a spec i fied fre quency but

at tenu ates sig nals of all lower fre quen cies.

homochronous Sig nals whose cor re spond ing sig nif i cant in stants have a con stant but

un con trolled phase re la tion ship with each other.

horn gap A light ning ar rester uti liz ing a gap be tween two horns. When light ning

causes a dis charge be tween the horns, the heat pro duced length ens the arc and
breaks it.

horse power The ba sic unit of me chan i cal power. One horse power (hp) equals 550

foot-pounds per sec ond or 746 watts.

hot A charged elec tri cal cir cuit or de vice.

hot dip gal va nized The pro cess of gal va niz ing steel by dip ping it into a bath of mol ten

zinc.

hot standby Sys tem equip ment that is fully pow ered but not in ser vice. A hot standby

can rap idly re place a pri mary sys tem in the event of a fail ure.

hum Un de sir able cou pling of the 60 Hz power sine wave into other elec tri cal sig nals

and/or cir cuits.

HVAC An ab bre vi a tion for heat ing, ven ti la tion, and air con di tion ing sys tem.

hy brid sys tem A com mu ni ca tion sys tem that ac com mo dates both dig i tal and an a log

sig nals.

hy drom e ter A test ing de vice used to mea sure spe cific grav ity, par tic u larly the spe cific

grav ity of the di lute sulphuric acid in a lead-acid stor age bat tery, to learn the state of
charge of the bat tery.

hy grom e ter An in stru ment that mea sures the rel a tive hu mid ity of the at mo sphere.

hy gro scopic The abil ity of a sub stance to ab sorb mois ture from the air.

hys ter esis The prop erty of an el e ment ev i denced by the de pend ence of the value of the

out put, for a given ex cur sion of the in put, upon the his tory of prior ex cur sions and
di rec tion of the in put. Or i ginally,

hys ter esis was the name for mag netic phe nom ena

only--the lag ging of flux den sity be hind the change in value of the mag ne tiz ing
flux--but now, the term is also used to de scribe other in elas tic be hav ior.

hys ter esis loop The plot of mag ne tiz ing cur rent against mag netic flux den sity (or of

other sim i larly re lated pairs of pa ram e ters), which ap pears as a loop. The area within
the loop is pro por tional to the power loss re sult ing from hys ter esis.

hys ter esis loss The loss in a mag netic core re sult ing from hys ter esis.

I

I

2

R

loss The power lost as a re sult of the heat ing ef fect of cur rent pass ing through re sis -

tance.

idling cur rent The cur rent drawn by a cir cuit, such as an am pli fier, when no sig nal is

pres ent at its in put.

im age fre quency A fre quency on which a car rier sig nal, when heterodyned with the

lo cal os cil la tor in a super het ero dyne re ceiver, will cause a sum or dif fer ence fre -
quency that is the same as the in ter me di ate fre quency of the re ceiver. Thus, a sig nal
on an

im age fre quency will be de mod u lated along with the de sired sig nal and will

in ter fere with it.

im pact ion iza tion The ion iza tion of an atom or mol e cule as a re sult of a high en ergy

col li sion.

im ped ance The to tal pas sive op po si tion of fered to the flow of an al ter nat ing cur rent.

Im ped ance con sists of a com bi na tion of re sis tance, in duc tive reactance, and ca pac i -
tive reactance. It is the vec tor sum of re sis tance and reactance (

R + jX) or the vec tor

of mag ni tude

Z at an an gle

.

im ped ance char ac ter is tic A graph of the im ped ance of a cir cuit show ing how it var ies

with fre quency.

im ped ance ir reg u lar ity A dis con ti nu ity in an im ped ance char ac ter is tic caused, for

ex am ple, by the use of dif fer ent co ax ial ca ble types.

im ped ance match ing The ad just ment of the im ped ances of ad join ing cir cuit com po -

nents to a com mon value so as to min i mize re flected en ergy from the junc tion and to
max i mize en ergy trans fer across it. In cor rect ad just ment re sults in an

im ped ance

mis match.

im ped ance match ing trans former A trans former used be tween two cir cuits of dif fer -

ent im ped ances with a turns ra tio that pro vides for max i mum power trans fer and
min i mum loss by re flec tion.

im pulse A short high en ergy surge of elec tri cal cur rent in a cir cuit or on a line.

im pulse cur rent A cur rent that rises rap idly to a peak then de cays to zero with out os -

cil lat ing.

im pulse ex ci ta tion The pro duc tion of an os cil la tory cur rent in a cir cuit by im press ing a

volt age for a rel a tively short pe riod com pared with the du ra tion of the cur rent pro -
duced.

im pulse noise A noise sig nal con sist ing of ran dom oc cur rences of en ergy spikes, hav -

ing ran dom am pli tude and band width.

im pulse re sponse The am pli tude-versus-time out put of a trans mis sion fa cil ity or de -

vice in re sponse to an im pulse.

im pulse volt age A uni di rec tional volt age that rises rap idly to a peak and then falls to

zero, with out any ap pre cia ble os cil la tion.

in-phase The prop erty of al ter nat ing cur rent sig nals of the same fre quency that achieve

their peak pos i tive, peak neg a tive, and zero am pli tude val ues si mul ta neously.

in ci dence an gle The an gle be tween the per pen dic u lar to a sur face and the di rec tion of

ar rival of a sig nal.

in cre ment A small change in the value of a quan tity.

in duce To pro duce an elec tri cal or mag netic ef fect in one con duc tor by chang ing the

con di tion or po si tion of an other con duc tor.

in duced cur rent The cur rent that flows in a con duc tor be cause a volt age has been in -

duced across two points in, or con nected to, the con duc tor.

in duced volt age A volt age de vel oped in a con duc tor when the con duc tor passes

through mag netic lines of force.

in duc tance The prop erty of an in duc tor that op poses any change in a cur rent that flows

through it. The stan dard unit of in duc tance is the

Henry.

in duc tion The elec tri cal and mag netic in ter ac tion pro cess by which a chang ing cur rent

in one cir cuit pro duces a volt age change not only in its own cir cuit (

self in duc tance)

but also in other cir cuits to which it is linked mag net i cally.

in duc tive A cir cuit el e ment ex hib it ing in duc tive reactance.

in duc tive kick A volt age surge pro duced when a cur rent flow ing through an in duc -

tance is in ter rupted.

in duc tive load A load that pos sesses a net in duc tive reactance.

in duc tive reactance The reactance of a cir cuit re sult ing from the pres ence of in duc -

tance and the phe nom e non of in duc tion.

in duc tor A coil of wire, usu ally wound on a core of high per me abil ity, that pro vides

high in duc tance with out nec es sar ily ex hib it ing high re sis tance.

in ert An in ac tive unit, or a unit that has no power re quire ments.
in fi nite line A trans mis sion line that ap pears to be of in fi nite length. There are no re -

flec tions back from the far end be cause it is ter mi nated in its char ac ter is tic im ped -
ance.

in fra low fre quency (ILF) The fre quency band from 300 Hz to 3000 Hz.
in hibit A con trol sig nal that pre vents a de vice or cir cuit from op er at ing.

in jec tion The ap pli ca tion of a sig nal to an elec tronic de vice.

in put The wave form fed into a cir cuit, or the ter mi nals that re ceive the in put wave form.
in ser tion gain The gain re sult ing from the in ser tion of a trans ducer in a trans mis sion

sys tem, ex pressed as the ra tio of the power de liv ered to that part of the sys tem fol -
low ing the trans ducer to the power de liv ered to that same part be fore in ser tion. If
more than one com po nent is in volved in the in put or out put, the par tic u lar com po -
nent used must be spec i fied. This ra tio is usu ally ex pressed in deci bels. If the re sult -
ing num ber is neg a tive, an

in ser tion loss is in di cated.

in ser tion loss The sig nal loss within a cir cuit, usu ally ex pressed in deci bels as the ra tio

of in put power to out put power.

in ser tion loss-vs.-fre quency char ac ter is tic The am pli tude trans fer char ac ter is tic of a

sys tem or com po nent as a func tion of fre quency. The am pli tude re sponse may be
stated as ac tual gain, loss, am pli fi ca tion, or at ten u a tion, or as a ra tio of any one of
these quan ti ties at a par tic u lar fre quency, with re spect to that at a spec i fied ref er ence
fre quency.

in spec tion lot A col lec tion of units of prod uct from which a sam ple is drawn and in -

spected to de ter mine con for mance with ac cept abil ity cri te ria.

in stan ta neous value The value of a vary ing wave form at a given in stant of time. The

value can be in volts, am peres, or phase an gle.

In sti tute of Elec tri cal and Elec tronics En gi neers (IEEE) The or ga ni za tion of elec -

tri cal and elec tron ics sci en tists and en gi neers formed in 1963 by the merger of the
In sti tute of Ra dio En gi neers (IRE) and the Amer i can In sti tute of Elec tri cal En gi -
neers (AIEE).

in stru ment mul ti plier A mea sur ing de vice that en ables a high volt age to be mea sured

us ing a me ter with only a low volt age range.

in stru ment rat ing The range within which an in stru ment has been de signed to op er ate

with out dam age.

in su late The pro cess of sep a rat ing one con duct ing body from an other con duc tor.

in su la tion The ma te rial that sur rounds and in su lates an elec tri cal wire from other wires

or cir cuits.

In su la tion may also re fer to any ma te rial that does not ion ize eas ily and

thus pres ents a large im ped ance to the flow of elec tri cal cur rent.

in su la tor A ma te rial or de vice used to sep a rate one con duct ing body from an other.

in tel li gence sig nal A sig nal con tain ing in for ma tion.
in ten sity The strength of a given sig nal un der spec i fied con di tions.

in ter con nect ca ble A short dis tance ca ble in tended for use be tween equip ment (gen er -

ally less than 3 m in length)

in ter face A de vice or cir cuit used to in ter con nect two pieces of elec tronic equip ment.
in ter face de vice A unit that joins two in ter con nect ing sys tems.

in ter fer ence emis sion An emis sion that re sults in an elec tri cal sig nal be ing prop a gated

into and in ter fer ing with the proper op er a tion of elec tri cal or elec tronic equip ment.

in ter lock A pro tec tion de vice or sys tem de signed to re move all dan ger ous volt ages

from a ma chine or piece of equip ment when ac cess doors or pan els are opened or re -
moved.

in ter me di ate fre quency A fre quency that re sults from com bin ing a sig nal of in ter est

with a sig nal gen er ated within a ra dio re ceiver. In super het ero dyne re ceiv ers, all in -
com ing sig nals are con verted to a sin gle in ter me di ate fre quency for which the am -
pli fi ers and fil ters of the re ceiver have been op ti mized.

in ter mit tent A non con tin u ous re cur ring event, of ten used to de note a prob lem that is

dif fi cult to find be cause of its un pre dict able na ture.

intermodulation The pro duc tion, in a non lin ear trans ducer el e ment, of fre quen cies

cor re spond ing to the sums and dif fer ences of the fun da men tals and har mon ics of
two or more fre quen cies that are trans mit ted through the trans ducer.

intermodulation dis tor tion (IMD) The dis tor tion that re sults from the mix ing of two

in put sig nals in a non lin ear sys tem. The re sult ing out put con tains new fre quen cies
that rep re sent the sum and dif fer ence of the in put sig nals and the sums and dif fer -
ences of their har mon ics. IMD is also called

intermodulation noise.

intermodulation noise In a trans mis sion path or de vice, the noise sig nal that is con tin -

gent upon mod u la tion and de mod u la tion, re sult ing from non lin ear char ac ter is tics in
the path or de vice.

in ter nal re sis tance The ac tual re sis tance of a source of elec tric power. The to tal elec -

tro mo tive force pro duced by a power source is not avail able for ex ter nal use; some
of the en ergy is used in driv ing cur rent through the source it self.

In ter na tional Stan dards Or ga ni za tion (ISO) An in ter na tional body con cerned with

world wide stan dard iza tion for a broad range of in dus trial prod ucts, in clud ing tele -
com mu ni ca tions equip ment. Mem bers are rep re sented by na tional stan dards or ga -
ni za tions, such as ANSI (Amer i can Na tional Stan dards In sti tute) in the United
States. ISO was es tab lished in 1947 as a spe cial ized agency of the United Na tions.

In ter na tional Tele com mu ni ca tions Un ion (ITU) A spe cial ized agency of the United

Na tions es tab lished to main tain and ex tend in ter na tional co op er a tion for the main -
te nance, de vel op ment, and ef fi cient use of tele com mu ni ca tions. The un ion does this
through stan dards and rec om mended reg u la tions, and through tech ni cal and tele -
com mu ni ca tions stud ies.

In ter na tional Tele com mu ni ca tions Sat el lite Con sor tium (Intelsat) A non profit co -

op er a tive of mem ber na tions that owns and op er ates a sat el lite sys tem for in ter na -
tional and, in many in stances, do mes tic com mu ni ca tions.

interoperability The con di tion achieved among com mu ni ca tions and elec tron ics sys -

tems or equip ment when in for ma tion or ser vices can be ex changed di rectly be tween
them or their us ers, or both.

in ter po late The pro cess of es ti mat ing un known val ues based on a knowl edge of com -

pa ra ble data that falls on both sides of the point in ques tion.

in ter rupt ing ca pac ity The rat ing of a cir cuit breaker or fuse that spec i fies the max i -

mum cur rent the de vice is de signed to in ter rupt at its rated volt age.

in ter val The points or num bers ly ing be tween two spec i fied end points.

in verse volt age The ef fec tive value of volt age across a rec ti fy ing de vice, which con -

ducts a cur rent in one di rec tion dur ing one half cy cle of the al ter nat ing in put, dur ing
the half cy cle when cur rent is not flow ing.

in ver sion The change in the po lar ity of a pulse, such as from pos i tive to neg a tive.

in verter A cir cuit or de vice that con verts a di rect cur rent into an al ter nat ing cur rent.
ion iz ing ra di a tion The form of elec tro mag netic ra di a tion that can turn an atom into an

ion by knock ing one or more of its elec trons loose. Ex am ples of ion iz ing ra di a tion
in clude X rays, gamma rays, and cos mic rays

IR

drop A drop in volt age be cause of the flow of cur rent (I) through a re sis tance (R),

also called

re sis tance drop.

IR

loss The con ver sion of elec tri cal power to heat caused by the flow of elec tri cal cur -

rent through a re sis tance.

isochronous A sig nal in which the time in ter val sep a rat ing any two sig nif i cant in stants

is the o ret i cally equal to a spec i fied unit in ter val or to an in te gral mul ti ple of the unit
in ter val.

iso lated ground A ground cir cuit that is iso lated from all equip ment frame work and

any other grounds, ex cept for a sin gle-point ex ter nal con nec tion.

iso lated ground plane A set of con nected frames that are grounded through a sin gle

con nec tion to a ground ref er ence point. That point and all parts of the frames are in -
su lated from any other ground sys tem in a build ing.

iso lated pulse A pulse un in flu enced by other pulses in the same sig nal.

isophasing am pli fier A tim ing de vice that cor rects for small tim ing er rors.
iso tro pic A quan tity ex hib it ing the same prop er ties in all planes and di rec tions.

J

jack A re cep ta cle or con nec tor that makes elec tri cal con tact with the mat ing con tacts of

a plug. In com bi na tion, the plug and jack pro vide a ready means for mak ing con nec -
tions in elec tri cal cir cuits.

jacket An in su lat ing layer of ma te rial sur round ing a wire in a ca ble.
jit ter Small, rapid vari a tions in a wave form re sult ing from fluc tu a tions in a sup ply

volt age or other causes.

joule The stan dard unit of work that is equal to the work done by one new ton of force

when the point at which the force is ap plied is dis placed a dis tance of one me ter in
the di rec tion of the force. The

joule is named for the Eng lish phys i cist James

Prescott Joule (1818-1889).

Julian date A chro no log i cal date in which days of the year are num bered in se quence.

For ex am ple, the first day is 001, the sec ond is 002, and the last is 365 (or 366 in a
leap year).

K

Kel vin (K) The stan dard unit of ther mo dy namic tem per a ture. Zero de grees Kel vin rep -

re sents

ab so lute zero. Wa ter freezes at 273 K and wa ter boils at 373 K un der stan -

dard pres sure con di tions.

kilo A pre fix mean ing one thou sand.

ki lo hertz (kHz) A unit of mea sure of fre quency equal to 1,000 Hz.

kilovar A unit equal to one thou sand volt-amperes.
kilo volt (kV) A unit of mea sure of elec tri cal volt age equal to 1,000 V.

ki lo watt A unit equal to one thou sand watts.

Kirchoff's Law At any point in a cir cuit, there is as much cur rent flow ing into the point

as there is flow ing away from it.

kly stron (1--gen eral) A fam ily of elec tron tubes that func tion as mi cro wave am pli fi -

ers and os cil la tors. Sim plest in form are two-cavity kly strons in which an elec tron
beam passes through a cav ity that is ex cited by a mi cro wave in put, pro duc ing a ve -
loc ity-modulated beam which passes through a sec ond cav ity a pre cise dis tance
away that is cou pled to a tuned cir cuit, thereby pro duc ing an am pli fied out put of the
orig i nal in put sig nal fre quency. If part of the out put is fed back to the in put, an os cil -
la tor can be the re sult.

(2--multi-cavity) An am pli fier de vice for UHF and mi cro -

wave sig nals based on ve loc ity mod u la tion of an elec tron beam. The beam is di -
rected through an in put cav ity, where the in put RF sig nal po lar ity initializes a
bunch ing ef fect on elec trons in the beam. The bunch ing ef fect ex cites sub se quent
cav i ties, which in crease the bunch ing through an en ergy fly wheel con cept. Finally,
the beam passes to an out put cav ity that cou ples the am pli fied sig nal to the load (an -
tenna sys tem). The beam falls onto a col lec tor el e ment that forms the re turn path for
the cur rent and dis si pates the heat re sult ing from elec tron beam bom bard ment.
(3--re flex) A kly stron with only one cav ity. The ac tion is the same as in a
two-cavity kly stron but the beam is re flected back into the cav ity in which it was
first ex cited, af ter be ing sent out to a re flec tor. The one cav ity, there fore, acts both as
the orig i nal ex citer (or buncher) and as the col lec tor from which the out put is taken.

knee In a re sponse curve, the re gion of max i mum cur va ture.

ku band Ra dio fre quen cies in the range of 15.35 GHz to 17.25 GHz, typ i cally used for

sat el lite tele com mu ni ca tions.

L

lad der net work A type of fil ter with com po nents al ter nately across the line and in the

line.

lag The dif fer ence in phase be tween a cur rent and the volt age that pro duced it, ex -

pressed in elec tri cal de grees.

lag ging cur rent A cur rent that lags be hind the al ter nat ing elec tro mo tive force that pro -

duced it. A cir cuit that pro duces a

lag ging cur rent is one con tain ing in duc tance

alone, or whose ef fec tive im ped ance is in duc tive.

lag ging load A load whose com bined in duc tive reactance ex ceeds its ca pac i tive

reactance. When an al ter nat ing volt age is ap plied, the cur rent lags be hind the volt -
age.

lam i nate A ma te rial con sist ing of lay ers of the same or dif fer ent ma te ri als bonded to -

gether and built up to the re quired thick ness.

lat i tude An an gu lar mea sure ment of a point on the earth above or be low the equa tor.

The equa tor rep re sents 0°, the north pole +90°, and the south pole 90°.

lay out A pro posed or ac tual ar range ment or al lo ca tion of equip ment.

LC cir cuit An elec tri cal cir cuit with both in duc tance (L) and ca pac i tance (C) that is

res o nant at a par tic u lar fre quency.

LC ra tio The ra tio of in duc tance to ca pac i tance in a given cir cuit.

lead An elec tri cal wire, usu ally in su lated.
lead ing edge The ini tial por tion of a pulse or wave in which volt age or cur rent rise rap -

idly from zero to a fi nal value.

lead ing load A re ac tive load in which the reactance of ca pac i tance is greater than that

of in duc tance. Cur rent through such a load

leads the ap plied volt age caus ing the cur -

rent.

leak age The loss of en ergy re sult ing from the flow of elec tric ity past an in su lat ing ma -

te rial, the es cape of elec tro mag netic ra di a tion be yond its shield ing, or the ex ten sion
of mag netic lines of force be yond their in tended work ing area.

leak age re sis tance The re sis tance of a path through which leak age cur rent flows.

level The strength or in ten sity of a given sig nal.

level align ment The ad just ment of trans mis sion lev els of sin gle links and links in tan -

dem to pre vent over load ing of trans mis sion sub sys tems.

life cy cle The pre dicted use ful life of a class of equip ment, op er at ing un der nor mal

(spec i fied) work ing con di tions.

life safety sys tem A sys tem de signed to pro tect life and prop erty, such as emer gency

light ing, fire alarms, smoke ex haust and ven ti lat ing fans, and site se cu rity.

life test A test in which ran dom sam ples of a prod uct are checked to see how long they

can con tinue to per form their func tions sat is fac to rily. A form of

stress test ing is

used, in clud ing tem per a ture, cur rent, volt age, and/or vi bra tion ef fects, cy cled at
many times the rate that would ap ply in nor mal us age.

lim iter An elec tronic de vice in which some char ac ter is tic of the out put is au to mat i cally

pre vented from ex ceed ing a pre de ter mined value.

lim iter cir cuit A cir cuit of non lin ear el e ments that re stricts the elec tri cal ex cur sion of a

vari able in ac cor dance with some spec i fied cri te ria.

lim it ing A pro cess by which some char ac ter is tic at the out put of a de vice is pre vented

from ex ceed ing a pre de ter mined value.

line loss The to tal end-to-end loss in deci bels in a trans mis sion line.

line-up The pro cess of ad just ing trans mis sion pa ram e ters to bring a cir cuit to its spec i -

fied val ues.

lin ear A cir cuit, de vice, or chan nel whose out put is di rectly pro por tional to its in put.

lin ear dis tor tion A dis tor tion mech a nism that is in de pend ent of sig nal am pli tude.

lin ear ity A con stant re la tion ship, over a des ig nated range, be tween the in put and out -

put char ac ter is tics of a cir cuit or de vice.

lines of force A group of imag i nary lines in di cat ing the di rec tion of the elec tric or mag -

netic field at all points along it.

lissa jous pat tern The loop ing pat terns gen er ated by a CRT spot when the hor i zon tal

(X) and ver ti cal (Y) de flec tion sig nals are si nu soids. The lissa jous pat tern is use ful
for eval u at ing the de lay or phase of two si nu soids of the same fre quency.

live A de vice or sys tem con nected to a source of elec tric po ten tial.
load The work re quired of an elec tri cal or me chan i cal sys tem.
load fac tor The ra tio of the av er age load over a des ig nated pe riod of time to the peak

load oc cur ring dur ing the same pe riod.

load line A straight line drawn across a group ing of plate cur rent/plate volt age char ac -

ter is tic curves show ing the re la tion ship be tween grid volt age and plate cur rent for a
par tic u lar plate load re sis tance of an elec tron tube.

log a rithm The power to which a base must be raised to pro duce a given num ber. Com -

mon log a rithms are to base 10.

log a rith mic scale A me ter scale with dis place ment pro por tional to the log a rithm of the

quan tity rep re sented.

long per sis tence The qual ity of a cath ode ray tube that has phos pho res cent com pounds

on its screen (in ad di tion to flu o res cent com pounds) so that the im age con tin ues to
glow af ter the orig i nal elec tron beam has ceased to cre ate it by pro duc ing the usual
flu o res cence ef fect. Long per sis tence is of ten used in ra dar screens or where pho to -
graphic ev i dence is needed of a dis play. Most such ap pli ca tions, how ever, have
been su per seded through the use of dig i tal stor age tech niques.

lon gi tude The an gu lar mea sure ment of a point on the sur face of the earth in re la tion to

the me rid ian of Green wich (Lon don). The earth is di vided into 360° of lon gi tude,
be gin ning at the Green wich mean. As one trav els west around the globe, the lon gi -
tude in creases.

lon gi tu di nal cur rent A cur rent that trav els in the same di rec tion on both wires of a

pair. The re turn cur rent ei ther flows in an other pair or via a ground re turn path.

loss The power dis si pated in a cir cuit, usu ally ex pressed in deci bels, that per forms no

use ful work.

loss de vi a tion The change of ac tual loss in a cir cuit or sys tem from a de signed value.

loss vari a tion The change in ac tual mea sured loss over time.
lossy The con di tion when the line loss per unit length is sig nif i cantly greater than some

de fined nor mal pa ram e ter.

lossy ca ble A co ax ial ca ble con structed to have high trans mis sion loss so it can be used

as an ar ti fi cial load or as an at tenu ator.

lot size A spe cific quan tity of sim i lar ma te rial or a col lec tion of sim i lar units from a

com mon source; in in spec tion work, the quan tity of fered for in spec tion and ac cep -
tance at any one time. The

lot size may be a col lec tion of raw ma te rial, parts, sub as -

sem blies in spected dur ing pro duc tion, or a con sign ment of fin ished prod ucts to be
sent out for ser vice.

low ten sion A low volt age cir cuit.

low-pass fil ter A fil ter net work that passes all fre quen cies be low a spec i fied fre quency

with lit tle or no loss, but that sig nif i cantly at tenu ates higher fre quen cies.

lug A tag or pro ject ing ter mi nal onto which a wire may be con nected by wrap ping, sol -

der ing, or crimp ing.

lumped con stant A re sis tance, in duc tance, or ca pac i tance con nected at a point, and

not dis trib uted uni formly through out the length of a route or cir cuit.

M

mA An ab bre vi a tion for mil li am peres (0.001 A).

mag net A de vice that pro duces a mag netic field and can at tract iron, and at tract or re pel

other mag nets.

mag netic field An en ergy field that ex ists around mag netic ma te ri als and cur -

rent-carrying con duc tors. Mag netic fields com bine with elec tric fields in light and
ra dio waves.

mag netic flux The field pro duced in the area sur round ing a mag net or elec tric cur rent.

The stan dard unit of flux is the

Weber.

mag netic flux den sity A vec tor quan tity mea sured by a stan dard unit called the Tesla.

The

mag netic flux den sity is the num ber of mag netic lines of force per unit area, at

right an gles to the lines.

mag netic leak age The mag netic flux that does not fol low a use ful path.

mag netic pole A point that ap pears from the out side to be the cen ter of mag netic at trac -

tion or re pul sion at or near one end of a mag net.

mag netic storm A vi o lent lo cal vari a tion in the earth's mag netic field, usu ally the re -

sult of sun spot ac tiv ity.

mag ne tism A prop erty of iron and some other ma te ri als by which ex ter nal mag netic

fields are main tained, other mag nets be ing thereby at tracted or re pelled.

mag ne ti za tion The ex po sure of a mag netic ma te rial to a mag ne tiz ing cur rent, field, or

force.

mag ne tiz ing force The force pro duc ing mag ne ti za tion.

magnetomotive force The force that tends to pro duce lines of force in a mag netic cir -

cuit. The

magnetomotive force bears the same re la tion ship to a mag netic cir cuit that

volt age does to an elec tri cal cir cuit.

mag ne tron A high-power, ul tra high fre quency elec tron tube os cil la tor that em ploys

the in ter ac tion of a strong elec tric field be tween an an ode and cath ode with the field
of a strong per ma nent mag net to cause os cil la tory elec tron flow through mul ti ple in -
ter nal cav ity res o na tors. The mag ne tron may op er ate in a con tin u ous or pulsed
mode.

main tain abil ity The prob a bil ity that a fail ure will be re paired within a spec i fied time

af ter the fail ure oc curs.

main te nance Any ac tiv ity in tended to keep a func tional unit in sat is fac tory work ing

con di tion. The term in cludes the tests, mea sure ments, re place ments, ad just ments,
and re pairs nec es sary to keep a de vice or sys tem op er at ing prop erly.

mal func tion An equip ment fail ure or a fault.

ma nom e ter A test de vice for mea sur ing gas pres sure.
mar gin The dif fer ence be tween the value of an op er at ing pa ram e ter and the value that

would re sult in un sat is fac tory op er a tion. Typ i cal

mar gin pa ram e ters in clude sig nal

level, sig nal-to-noise ra tio, dis tor tion, crosstalk cou pling, and/or un de sired emis -
sion level.

Markov model A sta tis ti cal model of the be hav ior of a com plex sys tem over time in

which the prob a bil i ties of the oc cur rence of var i ous fu ture states de pend only on the
pres ent state of the sys tem, and not on the path by which the pres ent state was
achieved. This term was named for the Rus sian math e ma ti cian Andrei Andreevich
Markov (1856-1922).

mas ter clock An ac cu rate tim ing de vice that gen er ates a syn chro nous sig nal to con trol

other clocks or equip ment.

mas ter os cil la tor A sta ble os cil la tor that pro vides a stan dard fre quency sig nal for other

hard ware and/or sys tems.

matched ter mi na tion A ter mi na tion that ab sorbs all the in ci dent power and so pro -

duces no re flected waves or mis match loss.

match ing The con nec tion of chan nels, cir cuits, or de vices in a man ner that re sults in

min i mal re flected en ergy.

ma trix A log i cal net work con fig ured in a rect an gu lar ar ray of in ter sec tions of in -

put/out put sig nals.

Maxwell's equa tions Four dif fer en tial equa tions that re late elec tric and mag netic

fields to elec tro mag netic waves. The equa tions are a ba sis of elec tri cal and elec -
tronic en gi neer ing.

mean An arith me tic av er age in which val ues are added and di vided by the num ber of

such val ues.

mean time be tween fail ures (MTBF) For a par tic u lar in ter val, the to tal func tion ing

life of a pop u la tion of an item di vided by the to tal num ber of fail ures within the pop -
u la tion dur ing the mea sure ment in ter val.

mean time to fail ure (MTTF) The mea sured op er at ing time of a sin gle piece of equip -

ment di vided by the to tal num ber of fail ures dur ing the mea sured pe riod of time.
This mea sure ment is nor mally made dur ing that pe riod be tween early life and
wear-out fail ures.

mean time to re pair (MTTR) The to tal cor rec tive main te nance time on a com po nent

or sys tem di vided by the to tal num ber of cor rec tive main te nance ac tions dur ing a
given pe riod of time.

mea sure ment A pro ce dure for de ter min ing the amount of a quan tity.
me dian A value in a se ries that has as many read ings or val ues above it as be low.

me dium An elec tronic path way or mech a nism for pass ing in for ma tion from one point

to an other.

mega A pre fix mean ing one mil lion.
mega hertz (MHz) A quan tity equal to one mil lion Hertz (cy cles per sec ond).

meg ohm A quan tity equal to one mil lion ohms.

met ric sys tem A dec i mal sys tem of mea sure ment based on the me ter, the ki lo gram,

and the sec ond.

mi cro A pre fix mean ing one mil lionth.

mi cron A unit of length equal to one mil lionth of a me ter (1/25,000 of an inch).

microphonic(s) Un in tended noise in tro duced into an elec tronic sys tem by me chan i cal

vi bra tion of elec tri cal com po nents.

mi cro sec ond One mil lionth of a sec ond (0.000001 s).

microvolt A quan tity equal to one-millionth of a volt.
milli A pre fix mean ing one thou sandth.

milliammeter A test in stru ment for mea sur ing elec tri cal cur rent, of ten part of a

multimeter.

millihenry A quan tity equal to one-thousandth of a henry.
milliwatt A quan tity equal to one thou sandth of a watt.

min i mum dis cern ible sig nal The small est in put that will pro duce a dis cern ible change

in the out put of a cir cuit or de vice.

mixer A cir cuit used to com bine two or more sig nals to pro duce a third sig nal that is a

func tion of the in put wave forms.

mix ing ra tio The ra tio of the mass of wa ter va por to the mass of dry air in a given vol -

ume of air. The

mix ing ra tio af fects ra dio prop a ga tion.

mode An elec tro mag netic field dis tri bu tion that sat is fies the o ret i cal re quire ments for

prop a ga tion in a wave guide or os cil la tion in a cav ity.

mod i fied re frac tive in dex The sum of the re frac tive in dex of the air at a given height

above sea level, and the ra tio of this height to the ra dius of the earth.

mod u lar An equip ment de sign in which ma jor el e ments are readily sep a ra ble, and

which the user may re place, re duc ing the mean-time-to-repair.

mod u la tion The pro cess whereby the am pli tude, fre quency, or phase of a sin -

gle-frequency wave (the

car rier) is var ied in step with the in stan ta neous value of, or

sam ples of, a com plex wave (the

mod u lat ing wave).

mod u la tor A de vice that en ables the in tel li gence in an in for ma tion-carrying mod u lat -

ing wave to be con veyed by a sig nal at a higher fre quency. A

mod u la tor mod i fies a

car rier wave by am pli tude, phase, and/or fre quency as a func tion of a con trol sig nal
that car ries in tel li gence. Sig nals are

mod u lated in this way to per mit more ef fi cient

and/or re li able trans mis sion over any of sev eral me dia.

mod ule An as sem bly re place able as an en tity, of ten as an in ter change able plug-in item.

A

mod ule is not nor mally ca pa ble of be ing dis as sem bled.

mono stable A de vice that is sta ble in one state only. An in put pulse causes the de vice to

change state, but it re verts im me di ately to its sta ble state.

mo tor A ma chine that con verts elec tri cal en ergy into me chan i cal en ergy.
mo tor ef fect The re pul sion force ex erted be tween ad ja cent con duc tors car ry ing cur -

rents in op po site di rec tions.

mov ing coil Any de vice that uti lizes a coil of wire in a mag netic field in such a way that

the coil is made to move by vary ing the ap plied cur rent, or it self pro duces a vary ing
volt age be cause of its move ment.

ms An ab bre vi a tion for mil li sec ond (0.001 s).

multimeter A test in stru ment fit ted with sev eral ranges for mea sur ing volt age, re sis -

tance, and cur rent, and equipped with an an a log me ter or dig i tal dis play read out.
The

multimeter is also known as a volt-ohm-milliammeter, or VOM.

mul ti plex (MUX) The use of a com mon chan nel to con vey two or more chan nels. This

is done ei ther by split ting of the com mon chan nel fre quency band into nar rower
bands, each of which is used to con sti tute a dis tinct chan nel (

fre quency di vi sion mul -

ti plex), or by al lot ting this com mon chan nel to mul ti ple us ers in turn to con sti tute
dif fer ent in ter mit tent chan nels (

time di vi sion mul ti plex).

multi plexer A de vice or cir cuit that com bines sev eral sig nals onto a sin gle sig nal.

multiplexing A tech nique that uses a sin gle trans mis sion path to carry mul ti ple chan -

nels. In

time di vi sion multiplexing (TDM), path time is shared. For fre quency di vi -

sion multiplexing (FDM) or wave length di vi sion multiplexing (WDM), sig nals are
di vided into in di vid ual chan nels sent along the same path but at dif fer ent fre quen -
cies.

mul ti pli ca tion Sig nal mix ing that oc curs within a mul ti plier cir cuit.
mul ti plier A cir cuit in which one or more in put sig nals are mixed un der the di rec tion of

one or more con trol sig nals. The re sult ing out put is a com pos ite of the in put sig nals,
the char ac ter is tics of which are de ter mined by the scal ing spec i fied for the cir cuit.

mu tual in duc tion The prop erty of the mag netic flux around a con duc tor that in duces a

volt age in a nearby con duc tor. The volt age gen er ated in the sec ond ary con duc tor in
turn in duces a volt age in the pri mary con duc tor. The in duc tance of two con duc tors
so cou pled is re ferred to as

mu tual in duc tance.

mV An ab bre vi a tion for mil li volt (0.001 V).

mW An ab bre vi a tion for milliwatt (0.001 W).

N

nano A pre fix mean ing one bil lionth.

nanometer 1 x 10

9

me ter.

nano sec ond (ns) One bil lionth of a sec ond (1 x 10

9

s).

652 Chapter Thirteen

narrowband A com mu ni ca tions chan nel of re stricted band width, of ten re sult ing in

deg ra da tion of the trans mit ted sig nal.

narrowband emis sion An emis sion hav ing a spec trum ex hib it ing one or more sharp

peaks that are nar row in width com pared to the nom i nal band width of the mea sur ing
in stru ment, and are far enough apart in fre quency to be re solv able by the in stru ment.

Na tional Elec tri cal Code (NEC) A doc u ment pro vid ing rules for the in stal la tion of

elec tric wir ing and equip ment in pub lic and pri vate build ings, pub lished by the Na -
tional Fire Pro tec tion As so ci a tion. The NEC has been adopted as law by many states
and mu nic i pal i ties in the U.S.

Na tional In sti tute of Stan dards and Tech nol ogy (NIST) A nonregulatory agency of

the De part ment of Com merce that serves as a na tional ref er ence and mea sure ment
lab o ra tory for the phys i cal and en gi neer ing sci ences. For merly called the

Na tional

Bu reau of Stan dards, the agency was re named in 1988 and given the ad di tional re -
spon si bil ity of aid ing U.S. com pa nies in adopt ing new tech nol o gies to in crease their
in ter na tional com pet i tive ness.

neg a tive In a con duc tor or semi con duc tor ma te rial, an ex cess of elec trons or a de fi -

ciency of pos i tive charge.

neg a tive feed back The re turn of a por tion of the out put sig nal from a cir cuit to the in -

put but 180° out of phase. This type of feed back de creases sig nal am pli tude but sta -
bi lizes the am pli fier and re duces dis tor tion and noise.

neg a tive im ped ance An im ped ance char ac ter ized by a de crease in volt age drop across

a de vice as the cur rent through the de vice is in creased, or a de crease in cur rent
through the de vice as the volt age across it is in creased.

neu tral A de vice or ob ject hav ing no elec tri cal charge.

neu tral con duc tor A con duc tor in a power dis tri bu tion sys tem con nected to a point in

the sys tem that is de signed to be at neu tral po ten tial. In a bal anced sys tem, the neu -
tral con duc tor car ries no cur rent.

neu tral ground An in ten tional ground ap plied to the neu tral con duc tor or neu tral point

of a cir cuit, trans former, ma chine, ap pa ra tus, or sys tem.

new ton The stan dard unit of force. One new ton is the force that, when ap plied to a body

hav ing a mass of 1 kg, gives it an ac cel er a tion of 1 m/s

2

.

ni tro gen A gas widely used to pres sur ize ra dio fre quency trans mis sion lines. If a small

punc ture oc curs in the ca ble sheath, the ni tro gen keeps mois ture out so that ser vice is
not ad versely af fected.

node The points at which the cur rent is at min i mum in a trans mis sion sys tem in which

stand ing waves are pres ent.

noise Any ran dom dis tur bance or un wanted sig nal in a com mu ni ca tion sys tem that

tends to ob scure the clar ity or use ful ness of a sig nal in re la tion to its in tended use.

noise fac tor (NF) The ra tio of the noise power mea sured at the out put of a re ceiver to

the noise power that would be pres ent at the out put if the ther mal noise re sult ing
from the re sis tive com po nent of the source im ped ance were the only source of noise
in the sys tem.

noise fig ure A mea sure of the noise in dB gen er ated at the in put of an am pli fier, com -

pared with the noise gen er ated by an im ped ance-method re sis tor at a spec i fied tem -
per a ture.

noise fil ter A net work that at tenu ates noise fre quen cies.

noise gen er a tor A gen er a tor of wideband ran dom noise.

noise power ra tio (NPR) The ra tio, ex pressed in deci bels, of sig nal power to

intermodulation prod uct power plus re sid ual noise power, mea sured at the base band
level.

noise sup pres sor A fil ter or dig i tal sig nal pro cess ing cir cuit in a re ceiver or trans mit ter

that au to mat i cally re duces or elim i nates noise.

noise tem per a ture The tem per a ture, ex pressed in Kel vin, at which a re sis tor will de -

velop a par tic u lar noise volt age. The noise tem per a ture of a ra dio re ceiver is the
value by which the tem per a ture of the re sis tive com po nent of the source im ped ance
should be in creased--if it were the only source of noise in the sys tem--to cause the
noise power at the out put of the re ceiver to be the same as in the real sys tem.

nom i nal The most com mon value for a com po nent or pa ram e ter that falls be tween the

max i mum and min i mum lim its of a tol er ance range.

nom i nal value A spec i fied or in tended value in de pend ent of any un cer tainty in its re al -

iza tion.

nomogram A chart show ing three or more scales across which a straight edge may be

held in or der to read off a graph i cal so lu tion to a three-variable equa tion.

nonionizing ra di a tion Elec tro mag netic ra di a tion that does not turn an atom into an

ion. Ex am ples of nonionizing ra di a tion in clude vis i ble light and ra dio waves.

non con duc tor A ma te rial that does not con duct en ergy, such as elec tric ity, heat, or

sound.

non crit i cal tech ni cal load That part of the tech ni cal power load for a fa cil ity not re -

quired for min i mum ac cept able op er a tion.

noninductive A de vice or cir cuit with out sig nif i cant in duc tance.
nonlinearity A dis tor tion in which the out put of a cir cuit or sys tem does not rise or fall

in di rect pro por tion to the in put.

non tech ni cal load The part of the to tal op er a tional load of a fa cil ity used for such pur -

poses as gen eral light ing, air con di tion ing, and ven ti lat ing equip ment dur ing nor mal
op er a tion.

nor mal A line per pen dic u lar to an other line or to a sur face.

nor mal-mode noise Un wanted sig nals in the form of volt ages ap pear ing in line-to-line

and line-to-neutral sig nals.

nor mal ized fre quency The ra tio be tween the ac tual fre quency and its nom i nal value.

nor mal ized fre quency de par ture The fre quency de par ture di vided by the nom i nal

fre quency value.

nor mal ized fre quency dif fer ence The al ge braic dif fer ence be tween two nor mal ized

fre quen cies.

nor mal ized fre quency drift The fre quency drift di vided by the nom i nal fre quency

value.

nor mally closed Switch con tacts that are closed in their nonoperated state, or re lay

con tacts that are closed when the re lay is de-energized.

nor mally open Switch con tacts that are open in their nonoperated state, or re lay con -

tacts that are open when the re lay is de-energized.

north pole The pole of a mag net that seeks the north mag netic pole of the earth.

notch fil ter A cir cuit de signed to at ten u ate a spe cific fre quency band; also known as a

band stop fil ter.

notched noise A noise sig nal in which a nar row band of fre quen cies has been re moved.

ns An ab bre vi a tion for nano sec ond.

null A zero or min i mum amount or po si tion.

O

oc tave Any fre quency band in which the high est fre quency is twice the low est fre -

quency.

off-line A con di tion wherein de vices or sub sys tems are not con nected into, do not form

a part of, and are not sub ject to the same con trols as an op er a tional sys tem.

off set An in ten tional dif fer ence be tween the re al ized value and the nom i nal value.

ohm The unit of elec tric re sis tance through which one am pere of cur rent will flow

when there is a dif fer ence of one volt. The quan tity is named for the Ger man phys i -
cist Georg Si mon Ohm (1787-1854).

Ohm's law A law that sets forth the re la tion ship be tween volt age (E), cur rent (I), and

re sis tance (

R). The law states that E

I

R

= ×

,

I

E

R

=

, and

R

E

I

=

.

Ohm's Law is

named for the Ger man phys i cist Georg Si mon Ohm (1787-1854).

ohmic loss The power dis si pa tion in a line or cir cuit caused by elec tri cal re sis tance.

ohm me ter A test in stru ment used for mea sur ing re sis tance, of ten part of a multimeter.
ohms-per-volt A mea sure of the sen si tiv ity of a volt me ter.
on-line A de vice or sys tem that is en er gized and op er a tional, and ready to per form use -

ful work.

open An in ter rup tion in the flow of elec tri cal cur rent, as caused by a bro ken wire or

con nec tion.

open-circuit A de fined loop or path that closes on it self and con tains an in fi nite im ped -

ance.

open-circuit im ped ance The in put im ped ance of a cir cuit when its out put ter mi nals

are open, that is, not ter mi nated.

open-circuit volt age The volt age mea sured at the ter mi nals of a cir cuit when there is

no load and, hence, no cur rent flow ing.

op er at ing life time The pe riod of time dur ing which the prin ci pal pa ram e ters of a com -

po nent or sys tem re main within a pre scribed range.

op ti mize The pro cess of ad just ing for the best out put or max i mum re sponse from a cir -

cuit or sys tem.

or bit The path, rel a tive to a spec i fied frame of ref er ence, de scribed by the cen ter of

mass of a sat el lite or other ob ject in space, sub jected solely to nat u ral forces (mainly
grav i ta tional at trac tion).

or der of di ver sity The num ber of in de pend ently fad ing prop a ga tion paths or fre quen -

cies, or both, used in a di ver sity re cep tion sys tem.

orig i nal equip ment man u fac turer (OEM) A man u fac turer of equip ment that is used

in sys tems as sem bled and sold by oth ers.

os cil la tion A vari a tion with time of the mag ni tude of a quan tity with re spect to a spec i -

fied ref er ence when the mag ni tude is al ter nately greater than and smaller than the
ref er ence.

os cil la tor A nonrotating de vice for pro duc ing al ter nat ing cur rent, the out put fre quency

of which is de ter mined by the char ac ter is tics of the cir cuit.

os cil lo scope A test in stru ment that uses a dis play, usu ally a cath ode-ray tube, to show

the in stan ta neous val ues and wave forms of a sig nal that var ies with time or some
other pa ram e ter.

out-of-band en ergy En ergy emit ted by a trans mis sion sys tem that falls out side the fre -

quency spec trum of the in tended trans mis sion.

out age du ra tion The av er age elapsed time be tween the start and the end of an out age

pe riod.

out age prob a bil ity The prob a bil ity that an out age state will oc cur within a spec i fied

time pe riod. In the ab sence of spe cific known causes of out ages, the

out age prob a -

bil ity is the sum of all out age du ra tions di vided by the time pe riod of mea sure ment.

out age thresh old A de fined value for a sup ported per for mance pa ram e ter that es tab -

lishes the min i mum op er a tional ser vice per for mance level for that pa ram e ter.

out put im ped ance The im ped ance pre sented at the out put ter mi nals of a cir cuit, de -

vice, or chan nel.

out put stage The fi nal driv ing cir cuit in a piece of elec tronic equip ment.

ovenized crys tal os cil la tor (OXO) A crys tal os cil la tor en closed within a tem per a ture

reg u lated heater (oven) to main tain a sta ble fre quency de spite ex ter nal tem per a ture
vari a tions.

overcoupling A de gree of cou pling greater than the crit i cal cou pling be tween two res -

o nant cir cuits.

Overcoupling re sults in a wide band width cir cuit with two peaks in

the re sponse curve.

over load In a trans mis sion sys tem, a power greater than the amount the sys tem was de -

signed to carry. In a power sys tem, an over load could cause ex ces sive heat ing. In a
com mu ni ca tions sys tem, dis tor tion of a sig nal could re sult.

over shoot The first max i mum ex cur sion of a pulse be yond the 100% level. Over shoot

is the por tion of the pulse that ex ceeds its de fined level tem po rarily be fore set tling to
the cor rect level. Over shoot am pli tude is ex pressed as a per cent age of the de fined
level.

P

pentode An elec tron tube with five elec trodes, the cath ode, con trol grid, screen grid,

sup pres sor grid, and plate.

photocathode An elec trode in an elec tron tube that will emit elec trons when bom -

barded by pho tons of light.

pic ture tube A cath ode-ray tube used to pro duce an im age by vari a tion of the in ten sity

of a scan ning beam on a phos phor screen.

pin A ter mi nal on the base of a com po nent, such as an elec tron tube.

plasma (1--arc) An ion ized gas in an arc-discharge tube that pro vides a con duct ing

path for the dis charge.

(2--so lar) The ion ized gas at ex tremely high tem per a ture

found in the sun.

plate (1--elec tron tube) The an ode of an elec tron tube. (2--bat tery) An elec trode in

a stor age bat tery.

(3--ca pac i tor) One of the sur faces in a ca pac i tor. (4--chas sis) A

mount ing sur face to which equip ment may be fas tened.

prop a ga tion time de lay The time re quired for a sig nal to travel from one point to an -

other.

pro tec tor A de vice or cir cuit that pre vents dam age to lines or equip ment by con duct ing

dan ger ously high volt ages or cur rents to ground. Pro tec tor types in clude spark gaps,
semi con duc tors, varistors, and gas tubes.

prox im ity ef fect A non uni form cur rent dis tri bu tion in a con duc tor, caused by cur rent

flow in a nearby con duc tor.

pseudonoise In a spread-spectrum sys tem, a seem ingly ran dom se ries of pulses whose

fre quency spec trum re sem bles that of con tin u ous noise.

pseudorandom A se quence of sig nals that ap pears to be com pletely ran dom but have,

in fact, been care fully drawn up and re peat af ter a sig nif i cant time in ter val.

pseudorandom noise A noise sig nal that sat is fies one or more of the stan dard tests for

sta tis ti cal ran dom ness. Al though it seems to lack any def i nite pat tern, there is a se -
quence of pulses that re peats af ter a long time in ter val.

pseudorandom num ber se quence A se quence of num bers that sat is fies one or more

of the stan dard tests for sta tis ti cal ran dom ness. Al though it seems to lack any def i -
nite pat tern, there is a se quence that re peats af ter a long time in ter val.

pul sat ing di rect cur rent A cur rent chang ing in value at reg u lar or ir reg u lar in ter vals

but which has the same di rec tion at all times.

pulse One of the el e ments of a re pet i tive sig nal char ac ter ized by the rise and de cay in

time of its mag ni tude. A

pulse is usu ally short in re la tion to the time span of in ter est.

pulse de cay time The time re quired for the trail ing edge of a pulse to de crease from 90

per cent to 10 per cent of its peak am pli tude.

pulse du ra tion The time in ter val be tween the points on the lead ing and trail ing edges

of a pulse at which the in stan ta neous value bears a spec i fied re la tion to the peak
pulse am pli tude.

pulse du ra tion mod u la tion (PDM) The mod u la tion of a pulse car rier by vary ing the

width of the pulses ac cord ing to the in stan ta neous val ues of the volt age sam ples of
the mod u lat ing sig nal (also called

pulse width mod u la tion).

pulse edge The lead ing or trail ing edge of a pulse, de fined as the 50 per cent point of the

pulse rise or fall time.

pulse fall time The in ter val of time re quired for the edge of a pulse to fall from 90 per -

cent to 10 per cent of its peak am pli tude.

pulse in ter val The time be tween the start of one pulse and the start of the next.

pulse length The du ra tion of a pulse (also called pulse width).

pulse level The volt age am pli tude of a pulse.
pulse pe riod The time be tween the start of one pulse and the start of the next.

pulse ra tio The ra tio of the length of any pulse to the to tal pulse pe riod.

pulse rep e ti tion pe riod The time in ter val from the be gin ning of one pulse to the be gin -

ning of the next pulse.

pulse rep e ti tion rate The num ber of times each sec ond that pulses are trans mit ted.

pulse rise time The time re quired for the lead ing edge of a pulse to rise from 10 per cent

to 90 per cent of its peak am pli tude.

pulse train A se ries of pulses hav ing sim i lar char ac ter is tics.

pulse width The mea sured in ter val be tween the 50 per cent am pli tude points of the

lead ing and trail ing edges of a pulse.

punc ture A break down of in su la tion or of a di elec tric, such as in a ca ble sheath or in the

insulant around a con duc tor.

pW An ab bre vi a tion for picowatt, a unit of power equal to 10

12

W (-90 dBm).

Q

Q (qual ity fac tor) A fig ure of merit that de fines how close a coil co mes to func tion ing

as a pure in duc tor.

High Q de scribes an in duc tor with lit tle en ergy loss re sult ing

from re sis tance.

Q is found by di vid ing the in duc tive reactance of a de vice by its re -

sis tance.

quad ra ture A state of al ter nat ing cur rent sig nals sep a rated by one quar ter of a cy cle

(90°).

quad ra ture am pli tude mod u la tion (QAM) A pro cess that al lows two dif fer ent sig -

nals to mod u late a sin gle car rier fre quency. The two sig nals of in ter est am pli tude
mod u late two sam ples of the car rier that are of the same fre quency, but dif fer in
phase by 90°. The two re sul tant sig nals can be added and trans mit ted. Both sig nals
may be re cov ered at a de coder when they are de mod u lated 90° apart.

quad ra ture com po nent The com po nent of a volt age or cur rent at an an gle of 90° to a

ref er ence sig nal, re sult ing from in duc tive or ca pac i tive reactance.

quad ra ture phase shift key ing (QPSK) A type of phase shift key ing us ing four phase

states.

qual ity The ab sence of ob jec tion able dis tor tion.

qual ity as sur ance (QA) All those ac tiv i ties, in clud ing sur veil lance, in spec tion, con -

trol, and doc u men ta tion, aimed at en sur ing that a given prod uct will meet its per for -
mance spec i fi ca tions.

qual ity con trol (QC) A func tion whereby man age ment ex er cises con trol over the

qual ity of raw ma te rial or in ter me di ate prod ucts in or der to pre vent the pro duc tion
of de fec tive de vices or sys tems.

quan tum noise Any noise at trib ut able to the dis crete na ture of elec tro mag netic ra di a -

tion. Ex am ples in clude shot noise, pho ton noise, and re com bi na tion noise.

quan tum-limited op er a tion An op er a tion wherein the min i mum de tect able sig nal is

lim ited by quan tum noise.

quartz A crys tal line min eral that when elec tri cally ex cited vi brates with a sta ble pe -

riod. Quartz is typ i cally used as the fre quency-determining el e ment in os cil la tors
and fil ters.

quasi-peak de tec tor A de tec tor that de liv ers an out put volt age that is some frac tion of

the peak value of the reg u larly re peated pulses ap plied to it. The frac tion in creases
to ward unity as the pulse rep e ti tion rate in creases.

quick-break fuse A fuse in which the fus ible link is un der ten sion, pro vid ing for rapid

op er a tion.

qui es cent An in ac tive de vice, sig nal, or sys tem.

qui es cent cur rent The cur rent that flows in a de vice in the ab sence of an ap plied sig -

nal.

R

rack An equip ment rack, usu ally mea sur ing 19 in (48.26 cm) wide at the front mount -

ing rails.

rack unit (RU) A unit of mea sure of ver ti cal space in an equip ment en clo sure. One

rack unit is equal to 1.75 in (4.45 cm).

ra di ate The pro cess of emit ting elec tro mag netic en ergy.
ra di a tion The emis sion and prop a ga tion of elec tro mag netic en ergy in the form of

waves.

Ra di a tion is also called ra di ant en ergy.

ra di a tion scat ter ing The di ver sion of ther mal, elec tro mag netic, or nu clear ra di a tion

from its orig i nal path as a re sult of in ter ac tions or col li sions with at oms, mol e cules,
or large par ti cles in the at mo sphere or other me dia be tween the source of ra di a tion
and a point some dis tance away. As a re sult of scat ter ing, ra di a tion (es pe cially
gamma rays and neu trons) will be re ceived at such a point from many di rec tions,
rather than only from the di rec tion of the source.

ra dio The trans mis sion of sig nals over a dis tance by means of elec tro mag netic waves

in the ap prox i mate fre quency range of 150 kHz to 300 GHz. The term may also be
used to de scribe the equip ment used to trans mit or re ceive elec tro mag netic waves.

ra dio de tec tion The de tec tion of the pres ence of an ob ject by ra dio lo ca tion with out

pre cise de ter mi na tion of its po si tion.

ra dio fre quency in ter fer ence (RFI) The in tru sion of un wanted sig nals or elec tro mag -

netic noise into var i ous types of equip ment re sult ing from ra dio fre quency trans mis -
sion equip ment or other de vices us ing ra dio fre quen cies.

ra dio fre quency spec trum Those fre quency bands in the elec tro mag netic spec trum

that range from sev eral hun dred thou sand cy cles per sec ond (

very low fre quency) to

sev eral bil lion cy cles per sec ond (

mi cro wave fre quen cies).

ra dio rec og ni tion In mil i tary com mu ni ca tions, the de ter mi na tion by ra dio means of

the "friendly" or "un friendly" char ac ter of an air craft or ship.

ran dom noise Elec tro mag netic sig nals that orig i nate in tran sient elec tri cal dis tur -

bances and have ran dom time and am pli tude pat terns. Ran dom noise is gen er ally
un de sir able; how ever, it may also be gen er ated for test ing pur poses.

rated out put power The power avail able from an am pli fier or other de vice un der spec -

i fied con di tions of op er a tion.

RC con stant The time con stant of a re sis tor-capacitor cir cuit. The RC con stant is the

time in sec onds re quired for cur rent in an RC cir cuit to rise to 63 per cent of its fi nal
steady value or fall to 37 per cent of its orig i nal steady value, ob tained by mul ti ply -
ing the re sis tance value in ohms by the ca pac i tance value in far ads.

RC net work A cir cuit that con tains re sis tors and ca pac i tors, nor mally con nected in se -

ries.

reactance The part of the im ped ance of a net work re sult ing from in duc tance or ca pac i -

tance. The

reactance of a com po nent var ies with the fre quency of the ap plied sig nal.

re ac tive power The power cir cu lat ing in an ac cir cuit. It is de liv ered to the cir cuit dur -

ing part of the cy cle and is re turned dur ing the other half of the cy cle. The

re ac tive

power is ob tained by mul ti ply ing the volt age, cur rent, and the sine of the phase an -
gle be tween them.

re ac tor A com po nent with in duc tive reactance.

re ceived sig nal level (RSL) The value of a spec i fied band width of sig nals at the re -

ceiver in put ter mi nals rel a tive to an es tab lished ref er ence.

re ceiver Any de vice for re ceiv ing elec tri cal sig nals and con vert ing them to au di ble

sound, vis i ble light, data, or some com bi na tion of these elements.

re cep ta cle An elec tri cal socket de signed to re ceive a mat ing plug.

re cep tion The act of re ceiv ing, lis ten ing to, or watch ing in for ma tion-carrying sig nals.
rec ti fi ca tion The con ver sion of al ter nat ing cur rent into di rect cur rent.

rec ti fier A de vice for con vert ing al ter nat ing cur rent into di rect cur rent. A rec ti fier nor -

mally in cludes fil ters so that the out put is, within spec i fied lim its, smooth and free of
ac com po nents.

rec tify The pro cess of con vert ing al ter nat ing cur rent into di rect cur rent.

re dun dancy A sys tem de sign that pro vides a back-up for key cir cuits or com po nents in

the event of a fail ure. Re dun dancy im proves the over all re li abil ity of a sys tem.

re dun dant A con fig u ra tion when two com plete sys tems are avail able at one time. If

the on line sys tem fails, the backup will take over with no loss of ser vice.

ref er ence volt age A volt age used for con trol or com par i son pur poses.

reflectance The ra tio of re flected power to in ci dent power.

re flec tion An abrupt change, re sult ing from an im ped ance mis match, in the di rec tion

of prop a ga tion of an elec tro mag netic wave. For light, at the in ter face of two dis sim i -
lar ma te ri als, the in ci dent wave is re turned to its me dium of or i gin.

re flec tion co ef fi cient The ra tio be tween the am pli tude of a re flected wave and the am -

pli tude of the in ci dent wave. For large smooth sur faces, the re flec tion co ef fi cient
may be near unity.

re flec tion gain The in crease in sig nal strength that re sults when a re flected wave com -

bines, in phase, with an in ci dent wave.

re flec tion loss The ap par ent loss of sig nal strength caused by an im ped ance mis match

in a trans mis sion line or cir cuit. The loss re sults from the re flec tion of part of the sig -
nal back to ward the source from the point of the im ped ance dis con ti nu ity. The
greater the mis match, the greater the loss.

reflectometer A de vice that mea sures en ergy trav el ing in each di rec tion in a wave -

guide, used in de ter min ing the stand ing wave ra tio.

re frac tion The bend ing of a sound, ra dio, or light wave as it passes obliquely from a

me dium of one den sity to a me dium of an other den sity that var ies its speed.

reg u la tion The pro cess of ad just ing the level of some quan tity, such as cir cuit gain, by

means of an elec tronic sys tem that mon i tors an out put and feeds back a con trol ling
sig nal to con stantly main tain a de sired level.

reg u la tor A de vice that main tains its out put volt age at a con stant level.
rel a tive en ve lope de lay The dif fer ence in en ve lope de lay at var i ous fre quen cies when

com pared with a ref er ence fre quency that is cho sen as hav ing zero de lay.

rel a tive hu mid ity The ra tio of the quan tity of wa ter va por in the at mo sphere to the

quan tity that would cause sat u ra tion at the am bi ent tem per a ture.

rel a tive trans mis sion level The ra tio of the sig nal power in a trans mis sion sys tem to

the sig nal power at some point cho sen as a ref er ence. The ra tio is usu ally de ter mined
by ap ply ing a stan dard test sig nal at the in put to the sys tem and mea sur ing the gain
or loss at the lo ca tion of in ter est.

re lay A de vice by which cur rent flow ing in one cir cuit causes con tacts to op er ate that

con trol the flow of cur rent in an other cir cuit.

re lay ar ma ture The mov able part of an elec tro me chani cal re lay, usu ally cou pled to

spring sets on which con tacts are mounted.

re lay by pass A de vice that, in the event of a loss of power or other fail ure, routes a crit i -

cal sig nal around the equip ment that has failed.

re lease time The time re quired for a pulse to drop from steady-state level to zero, also

re ferred to as the

de cay time.

re li abil ity The abil ity of a sys tem or sub sys tem to per form within the pre scribed pa -

ram e ters of qual ity of ser vice.

Re li abil ity is of ten ex pressed as the prob a bil ity that a

sys tem or sub sys tem will per form its in tended func tion for a spec i fied in ter val un der
stated con di tions.

re li abil ity growth The ac tion taken to move a hard ware item to ward its re li abil ity po -

ten tial, dur ing de vel op ment or sub se quent man u fac tur ing or op er a tion.

re li abil ity pre dic tions The com piled fail ure rates for parts, com po nents, sub as sem -

blies, as sem blies, and sys tems. These ge neric fail ure rates are used as ba sic data to
pre dict the re li abil ity of a given de vice or sys tem.

re mote con trol A sys tem used to con trol a de vice from a dis tance.

re mote sta tion A sta tion or ter mi nal that is phys i cally re mote from a main sta tion or

cen ter but can gain ac cess through a com mu ni ca tion chan nel.

re peater The equip ment be tween two cir cuits that re ceives a sig nal de graded by nor -

mal fac tors dur ing trans mis sion and am pli fies the sig nal to its orig i nal level for re -
trans mis sion.

rep e ti tion rate The rate at which reg u larly re cur ring pulses are re peated.

re ply A trans mit ted mes sage that is a di rect re sponse to an orig i nal mes sage.
re pul sion The me chan i cal force that tends to sep a rate like mag netic poles, like elec tric

charges, or con duc tors car ry ing cur rents in op po site di rec tions.

re set The act of re stor ing a de vice to its de fault or orig i nal state.

re sid ual flux The mag netic flux that re mains af ter a magnetomotive force has been re -

moved.

re sid ual mag ne tism The mag ne tism or flux that re mains in a core af ter cur rent ceases

to flow in the coil pro duc ing the magnetomotive force.

re sid ual volt age The vec tor sum of the volt ages in all the phase wires of an un bal anced

polyphase power sys tem.

re sis tance The op po si tion of a ma te rial to the flow of elec tri cal cur rent. Re sis tance is

equal to the volt age drop through a given ma te rial di vided by the cur rent flow
through it. The stan dard unit of re sis tance is the

ohm, named for the Ger man phys i -

cist Georg Si mon Ohm (1787-1854).

re sis tance drop The fall in po ten tial (volts) be tween two points, the prod uct of the cur -

rent and re sis tance.

re sis tance-grounded A cir cuit or sys tem grounded for safety through a re sis tance,

which lim its the value of the cur rent flow ing through the cir cuit in the event of a
fault.

re sis tive load A load in which the volt age is in phase with the cur rent.

re sis tiv ity The re sis tance per unit vol ume or per unit area.

re sis tor A de vice the pri mary func tion of which is to in tro duce a spec i fied re sis tance

into an elec tri cal cir cuit.

res o nance A tuned con di tion con du cive to os cil la tion, when the reactance re sult ing

from ca pac i tance in a cir cuit is equal in value to the reactance re sult ing from in duc -
tance.

res o nant fre quency The fre quency at which the in duc tive reactance and ca pac i tive

reactance of a cir cuit are equal.

res o na tor A res o nant cav ity.

re turn A re turn path for cur rent, some times through ground.
re ver sal A change in mag netic po lar ity, in the di rec tion of cur rent flow.

re verse cur rent A small cur rent that flows through a di ode when the volt age across it

is such that nor mal for ward cur rent does not flow.

re verse volt age A volt age in the re verse di rec tion from that nor mally ap plied.

rheo stat A two-terminal vari able re sis tor, usu ally con structed with a slid ing or ro tat ing

shaft that can be used to vary the re sis tance value of the de vice.

rip ple An ac volt age su per im posed on the out put of a dc power sup ply, usu ally re sult -

ing from im per fect fil ter ing.

rise time The time re quired for a pulse to rise from 10 per cent to 90 per cent of its peak

value.

roll-off A grad ual at ten u a tion of gain-frequency re sponse at ei ther or both ends of a

trans mis sion pass band.

root-mean-square (RMS) The square root of the av er age value of the squares of all

the in stan ta neous val ues of cur rent or volt age dur ing one half-cycle of an al ter nat ing
cur rent. For an al ter nat ing cur rent, the RMS volt age or cur rent is equal to the

amount of di rect cur rent or volt age that would pro duce the same heat ing ef fect in a
purely re sis tive cir cuit. For a sinewave, the root-mean-square value is equal to 0.707
times the peak value. RMS is also called the

ef fec tive value.

ro tor The ro tat ing part of an elec tric gen er a tor or mo tor.

RU An ab bre vi a tion for rack unit.

S

scan One sweep of the tar get area in a cam era tube, or of the screen in a pic ture tube.

screen grid A grid in an elec tron tube that im proves per for mance of the de vice by

shield ing the con trol grid from the plate.

self-bias The pro vi sion of bias in an elec tron tube through a volt age drop in the cath ode

cir cuit.

shot noise The noise de vel oped in a vac uum tube or photoconductor re sult ing from the

ran dom num ber and ve loc ity of emit ted charge car ri ers.

slope The rate of change, with re spect to fre quency, of trans mis sion line at ten u a tion

over a given fre quency spec trum.

slope equal izer A de vice or cir cuit used to achieve a spec i fied slope in a trans mis sion

line.

smooth ing cir cuit A fil ter de signed to re duce the amount of rip ple in a cir cuit, usu ally

a dc power sup ply.

snub ber An elec tronic cir cuit used to sup press high fre quency noise.

so lar wind Charged par ti cles from the sun that con tin u ously bom bard the sur face of

the earth.

solid A sin gle wire con duc tor, as con trasted with a stranded, braided, or rope-type wire.

solid-state The use of semi con duc tors rather than elec tron tubes in a cir cuit or sys tem.

source The part of a sys tem from which sig nals or mes sages are con sid ered to orig i -

nate.

source ter mi nated A cir cuit whose out put is ter mi nated for cor rect im ped ance match -

ing with stan dard ca ble.

spare A sys tem that is avail able but not pres ently in use.
spark gap A gap be tween two elec trodes de signed to pro duce a spark un der given con -

di tions.

spe cific grav ity The ra tio of the weight of a vol ume, liq uid, or solid to the weight of the

same vol ume of wa ter at a spec i fied tem per a ture.

spec trum A con tin u ous band of fre quen cies within which waves have some com mon

char ac ter is tics.

spec trum an a lyzer A test in stru ment that pres ents a graphic dis play of sig nals over a

se lected fre quency band width. A cath ode-ray tube is of ten used for the dis play.

spec trum des ig na tion of fre quency A method of re fer ring to a range of com mu ni ca -

tion fre quen cies. In Amer i can prac tice, the des ig na tion is a two or three let ter ac ro -
nym for the name. The ranges are: be low 300 Hz, ELF (ex tremely low fre quency);
300 Hz3000 Hz, ILF (in fra low fre quency); 3 kHz30 kHz, VLF (very low fre -
quency); 30 kHz300 kHz, LF (low fre quency); 300 kHz3000 kHz, MF (me dium
fre quency); 3 MHz30 MHz, HF (high fre quency); 30 MHz300 MHz, VHF (very
high fre quency); 300 MHz3000 MHz, UHF (ul tra high fre quency); 3 GHz30
GHz, SHF (super high fre quency); 30 GHz300 GHz, EHF (ex tremely high fre -
quency); 300 GHz3000 GHz, THF (tre men dously high fre quency).

spher i cal an tenna A type of sat el lite re ceiv ing an tenna that per mits more than one sat -

el lite to be ac cessed at any given time. A spher i cal an tenna has a broader an gle of ac -
cep tance than a par a bolic an tenna.

spike A high am pli tude, short du ra tion pulse su per im posed on an oth er wise reg u lar

wave form.

split-phase A de vice that de rives a sec ond phase from a sin gle phase power sup ply by

pass ing it through a ca pac i tive or in duc tive re ac tor.

split ter A cir cuit or de vice that ac cepts one in put sig nal and dis trib utes it to sev eral out -

puts.

split ting ra tio The ra tio of the power emerg ing from the out put ports of a cou pler.

spo radic An event oc cur ring at ran dom and in fre quent in ter vals.

spread spec trum A com mu ni ca tions tech nique in which the fre quency com po nents of

a narrowband sig nal are spread over a wide band. The re sult ing sig nal re sem bles
white noise. The tech nique is used to achieve sig nal se cu rity and pri vacy, and to en -
able the use of a com mon band by many us ers.

spu ri ous sig nal Any por tion of a given sig nal that is not part of the fun da men tal wave -

form. Spu ri ous sig nals in clude tran sients, noise, and hum.

square wave A square or rect an gu lar-shaped pe ri odic wave that al ter nately as sumes

two fixed val ues for equal lengths of time, the tran si tion be ing neg li gi ble in com par -
i son with the du ra tion of each fixed value.

square wave test ing The use of a square wave con tain ing many odd har mon ics of the

fun da men tal fre quency as an in put sig nal to a de vice. Vi sual ex am i na tion of the out -
put sig nal on an os cil lo scope in di cates the amount of dis tor tion in tro duced.

sta bil ity The abil ity of a de vice or cir cuit to re main sta ble in fre quency, power level,

and/or other spec i fied pa ram e ters.

stan dard The spe cific sig nal con fig u ra tion, ref er ence pulses, volt age lev els, and other

pa ram e ters that de scribe the in put/out put re quire ments for a par tic u lar type of
equip ment.

stan dard time and fre quency sig nal A time-controlled ra dio sig nal broad cast at

sched uled in ter vals on a num ber of dif fer ent fre quen cies by gov ern ment-operated
ra dio sta tions to pro vide a method for cal i brat ing in stru ments.

stand ing wave ra tio (SWR) The ra tio of the max i mum to the min i mum value of a

com po nent of a wave in a trans mis sion line or wave guide, such as the max i mum
volt age to the min i mum volt age.

static charge An elec tric charge on the sur face of an ob ject, par tic u larly a di elec tric.

sta tion One of the in put or out put points in a com mu ni ca tions sys tem.

stator The sta tion ary part of a ro tat ing elec tric ma chine.
sta tus The pres ent con di tion of a de vice.

stat ute mile A unit of dis tance equal to 1,609 km or 5,280 ft.

steady-state A con di tion in which cir cuit val ues re main es sen tially con stant, oc cur ring

af ter all ini tial tran sients or fluc tu at ing con di tions have passed.

steady-state con di tion A con di tion oc cur ring af ter all ini tial tran sient or fluc tu at ing

con di tions have damped out in which cur rents, volt ages, or fields re main es sen tially
con stant or os cil late uni formly with out changes in char ac ter is tics such as am pli -
tude, fre quency, or wave shape.

steep wave front A rapid rise in volt age of a given sig nal, in di cat ing the pres ence of

high fre quency odd har mon ics of a fun da men tal wave fre quency.

step up (or down) The pro cess of in creas ing (or de creas ing) the volt age of an elec tri cal

sig nal, as in a step-up (or step-down) trans former.

straight-line ca pac i tance A ca pac i tance em ploy ing a vari able ca pac i tor with plates so

shaped that ca pac i tance var ies di rectly with the an gle of ro ta tion.

stray ca pac i tance An un in tended--and usu ally un de sired--ca pac i tance be tween

wires and com po nents in a cir cuit or sys tem.

stray cur rent A cur rent through a path other than the in tended one.
stress The force per unit of cross-sectional area on a given ob ject or struc ture

sub as sem bly A func tional unit of a sys tem.

subcarrier (SC) A car rier ap plied as mod u la tion on an other car rier, or on an in ter me -

di ate subcarrier.

subharmonic A fre quency equal to the fun da men tal fre quency of a given sig nal di -

vided by a whole num ber.

submodule A small cir cuit board or de vice that mounts on a larger mod ule or device.

subrefraction A re frac tion for which the refractivity gra di ent is greater than stan dard.
sub sys tem A func tional unit of a sys tem.

super het ero dyne re ceiver A ra dio re ceiver in which all sig nals are first con verted to a

com mon fre quency for which the in ter me di ate stages of the re ceiver have been op ti -
mized, both for tun ing and fil ter ing. Sig nals are con verted by mix ing them with the
out put of a lo cal os cil la tor whose out put is var ied in ac cor dance with the fre quency
of the re ceived sig nals so as to main tain the de sired

in ter me di ate fre quency.

sup pres sor grid The fifth grid of a pentode elec tron tube, which pro vides screen ing

be tween plate and screen grid.

sur face leak age A leak age cur rent from line to ground over the face of an in su la tor

sup port ing an open wire route.

sur face refractivity The re frac tive in dex, cal cu lated from ob ser va tions of pres sure,

tem per a ture, and hu mid ity at the sur face of the earth.

surge A rapid rise in cur rent or volt age, usu ally fol lowed by a fall back to the nor mal

value.

susceptance The re cip ro cal of reactance, and the imag i nary com po nent of ad mit tance,

ex pressed in siemens.

sweep The pro cess of vary ing the fre quency of a sig nal over a spec i fied band width.
sweep gen er a tor A test os cil la tor, the fre quency of which is con stantly var ied over a

spec i fied band width.

switch ing The pro cess of mak ing and break ing (con nect ing and dis con nect ing) two or

more elec tri cal cir cuits.

syn chro ni za tion The pro cess of ad just ing the cor re spond ing sig nif i cant in stants of

sig nals--for ex am ple, the zero-crossings--to make them syn chro nous. The term
syn chro ni za tion is of ten ab bre vi ated as sync.

syn chro nize The pro cess of caus ing two sys tems to op er ate at the same speed.

syn chro nous In step or in phase, as ap plied to two or more de vices; a sys tem in which

all events oc cur in a pre de ter mined timed se quence.

syn chro nous de tec tion A de mod u la tion pro cess in which the orig i nal sig nal is re cov -

ered by mul ti ply ing the mod u lated sig nal by the out put of a syn chro nous os cil la tor
locked to the car rier.

syn chro nous sys tem A sys tem in which the trans mit ter and re ceiver are op er at ing in a

fixed time re la tion ship.

sys tem stan dards The min i mum re quired elec tri cal per for mance char ac ter is tics of a

spe cific col lec tion of hard ware and/or soft ware.

sys tems anal y sis An anal y sis of a given ac tiv ity to de ter mine pre cisely what must be

ac com plished and how it is to be done.

T

tetrode A four el e ment elec tron tube con sist ing of a cath ode, con trol grid, screen grid,

and plate.

thyratron A gas-filled elec tron tube in which plate cur rent flows when the grid volt age

reaches a pre de ter mined level. At that point, the grid has no fur ther con trol over the
cur rent, which con tin ues to flow un til it is in ter rupted or re versed.

tol er ance The per mis si ble vari a tion from a stan dard.
torque A mo ment of force act ing on a body and tend ing to pro duce ro ta tion about an

axis.

to tal har monic dis tor tion (THD) The ra tio of the sum of the am pli tudes of all sig nals

har mon i cally re lated to the fun da men tal ver sus the am pli tude of the fun da men tal
sig nal. THD is ex pressed in per cent.

trace The pat tern on an os cil lo scope screen when dis play ing a sig nal.

track ing The lock ing of tuned stages in a ra dio re ceiver so that all stages are changed

ap pro pri ately as the re ceiver tun ing is changed.

trade-off The pro cess of weigh ing con flict ing re quire ments and reach ing a com pro -

mise de ci sion in the de sign of a com po nent or a sub sys tem.

trans ceiver Any cir cuit or de vice that re ceives and trans mits sig nals.
transconductance The mu tual con duc tance of an elec tron tube ex pressed as the

change in plate cur rent di vided by the change in con trol grid volt age that pro duced
it.

trans ducer A de vice that con verts en ergy from one form to an other.
trans fer char ac ter is tics The in trin sic pa ram e ters of a sys tem, sub sys tem, or unit of

equip ment which, when ap plied to the in put of the sys tem, sub sys tem, or unit of
equip ment, will fully de scribe its out put.

trans former A de vice con sist ing of two or more wind ings wrapped around a sin gle

core or linked by a com mon mag netic cir cuit.

trans former ra tio The ra tio of the num ber of turns in the sec ond ary wind ing of a trans -

former to the num ber of turns in the pri mary wind ing, also known as the

turns ra tio.

tran sient A sud den vari ance of cur rent or volt age from a steady-state value. A tran -

sient nor mally re sults from changes in load or ef fects re lated to switch ing ac tion.

tran sient dis tur bance A volt age pulse of high en ergy and short du ra tion im pressed

upon the ac wave form. The overvoltage pulse can be one to 100 times the nor mal ac
po ten tial (or more) and can last up to 15 ms. Rise times mea sure in the nano sec ond
range.

tran sient re sponse The time re sponse of a sys tem un der test to a stated in put stim u lus.
tran si tion A se quence of ac tions that oc curs when a pro cess changes from one state to

an other in re sponse to an in put.

trans mis sion The trans fer of elec tri cal power, sig nals, or an in tel li gence from one lo -

ca tion to an other by wire, fi ber op tic, or ra dio means.

trans mis sion fa cil ity A trans mis sion me dium and all the as so ci ated equip ment re -

quired to trans mit in for ma tion.

trans mis sion loss The ra tio, in deci bels, of the power of a sig nal at a point along a

trans mis sion path to the power of the same sig nal at a more dis tant point along the
same path. This value is of ten used as a mea sure of the qual ity of the trans mis sion
me dium for con vey ing sig nals. Changes in power level are nor mally ex pressed in
deci bels by cal cu lat ing ten times the log a rithm (base 10) of the ra tio of the two pow -
ers.

trans mis sion mode One of the field pat terns in a wave guide in a plane trans verse to the

di rec tion of prop a ga tion.

trans mis sion sys tem The set of equip ment that pro vides sin gle or mul ti chan nel com -

mu ni ca tions fa cil i ties ca pa ble of car ry ing au dio, video, or data sig nals.

trans mit ter The de vice or cir cuit that launches a sig nal into a pas sive me dium, such as

the at mo sphere.

trans par ency The prop erty of a com mu ni ca tions sys tem that en ables it to carry a sig -

nal with out al ter ing or oth er wise af fect ing the elec tri cal char ac ter is tics of the sig nal.

tray The metal cab i net that holds cir cuit boards.
tre men dously high fre quency (THF) The fre quency band from 300 GHz to 3000

GHz.

tri an gu lar wave An os cil la tion, the val ues of which rise and fall lin early, and im me di -

ately change upon reach ing their peak max i mum and min i mum. A graph i cal rep re -
sen ta tion of a tri an gu lar wave re sem bles a tri an gle.

trim The pro cess of mak ing fine ad just ments to a cir cuit or a cir cuit el e ment.

trim mer A small me chan i cally-adjustable com po nent con nected in par al lel or se ries

with a ma jor com po nent so that the net value of the two can be finely ad justed for
tun ing pur poses.

tri ode A three-element elec tron tube, con sist ing of a cath ode, con trol grid, and plate.

tri ple beat A third-order beat whose three beat ing car ri ers all have dif fer ent fre quen -

cies, but are spaced at equal fre quency sep a ra tions.

tro po sphere The layer of the earth's at mo sphere, be tween the sur face and the strato -

sphere, in which about 80 per cent of the to tal mass of at mo spheric air is con cen -
trated and in which tem per a ture nor mally de creases with al ti tude.

trou ble A fail ure or fault af fect ing the ser vice pro vided by a sys tem.

trou ble shoot The pro cess of in ves ti gat ing, lo cal iz ing and (if pos si ble) cor rect ing a

fault.

tube (1--elec tron) An evac u ated or gas-filled tube en closed in a glass or metal case in

which the elec trodes are main tained at dif fer ent volt ages, giv ing rise to a con trolled
flow of elec trons from the cath ode to the an ode.

(2--cath ode ray, CRT) An elec -

tron beam tube used for the dis play of chang ing elec tri cal phe nom ena, gen er ally
sim i lar to a tele vi sion pic ture tube.

(3--cold-cathode) An elec tron tube whose cath -

ode emits elec trons with out the need of a heat ing fil a ment.

(4--gas) A gas-filled

elec tron tube in which the gas plays an es sen tial role in op er a tion of the de vice.
(5--mer cury-vapor) A tube filled with mer cury va por at low pres sure, used as a
rec ti fy ing de vice.

(6--metal) An elec tron tube en closed in a metal case. (7--trav -

el ing wave, TWT) A wide band mi cro wave am pli fier in which a stream of elec trons
in ter acts with a guided elec tro mag netic wave mov ing sub stan tially in syn chro nism
with the elec tron stream, re sult ing in a net trans fer of en ergy from the elec tron
stream to the wave.

(8--ve loc ity-modulated) An elec tron tube in which the ve loc -

ity of the elec tron stream is con tin u ally chang ing, as in a kly stron.

tune The pro cess of ad just ing the fre quency of a de vice or cir cuit, such as for res o nance

or for max i mum re sponse to an in put sig nal.

tuned trap A se ries res o nant net work bridged across a cir cuit that elim i nates ("traps")

the fre quency of the res o nant net work.

tuner The ra dio fre quency and in ter me di ate fre quency parts of a ra dio re ceiver that

pro duce a low level out put sig nal.

tun ing The pro cess of ad just ing a given fre quency; in par tic u lar, to ad just for res o -

nance or for max i mum re sponse to a par tic u lar in com ing sig nal.

turns ra tio In a trans former, the ra tio of the num ber of turns on the sec ond ary to the

num ber of turns on the pri mary.

tweak ing The pro cess of ad just ing an elec tronic cir cuit to op ti mize its per for mance.

twin-line A feeder ca ble with two par al lel, in su lated con duc tors.

two-phase A source of al ter nat ing cur rent cir cuit with two si nu soi dal volt ages that are

90° apart.

U

ul tra high fre quency (UHF) The fre quency range from 300 MHz to 3000 MHz.
ul tra vi o let ra di a tion Elec tro mag netic ra di a tion in a fre quency range be tween vis i ble

light and high-frequency X-rays.

un at tended A de vice or sys tem de signed to op er ate with out a hu man at ten dant.

un at tended op er a tion A sys tem that per mits a sta tion to re ceive and trans mit mes -

sages with out the pres ence of an at ten dant or op er a tor.

un avail abil ity A mea sure of the de gree to which a sys tem, sub sys tem, or piece of

equip ment is not op er a ble and not in a com mit ta ble state at the start of a mis sion,
when the mis sion is called for at a ran dom point in time.

un bal anced cir cuit A two-wire cir cuit with legs that dif fer from one an other in re sis -

tance, ca pac ity to earth or to other con duc tors, leak age, or in duc tance.

un bal anced line A trans mis sion line in which the mag ni tudes of the volt ages on the

two con duc tors are not equal with re spect to ground. A co ax ial ca ble is an ex am ple
of an un bal anced line.

un bal anced mod u la tor A mod u la tor whose out put in cludes the car rier sig nal.
un bal anced out put An out put with one leg at ground po ten tial.

un bal anced wire cir cuit A cir cuit whose two sides are in her ently elec tri cally un like.

un cer tainty An ex pres sion of the mag ni tude of a pos si ble de vi a tion of a mea sured

value from the true value. Fre quently, it is pos si ble to dis tin guish two com po nents:
the

sys tem atic un cer tainty and the ran dom un cer tainty. The ran dom un cer tainty is

ex pressed by the stan dard de vi a tion or by a mul ti ple of the stan dard de vi a tion. The
sys tem atic un cer tainty is gen er ally es ti mated on the ba sis of the pa ram e ter char ac -
ter is tics.

un damped wave A sig nal with con stant am pli tude.

underbunching A con di tion in a trav el ing wave tube wherein the tube is not op er at ing

at its op ti mum bunch ing rate.

Un der writers Lab o ra tories, Inc. A lab o ra tory es tab lished by the Na tional Board of

Fire Un der writers which tests equip ment, ma te ri als, and sys tems that may af fect in -
sur ance risks, with spe cial at ten tion to fire dan gers and other haz ards to life.

un grounded A cir cuit or line not con nected to ground.
unicoupler A de vice used to cou ple a bal anced cir cuit to an un bal anced cir cuit.

uni di rec tional A sig nal or cur rent flow ing in one di rec tion only.

uni form trans mis sion line A trans mis sion line with elec tri cal char ac ter is tics that are

iden ti cal, per unit length, over its en tire length.

unit An as sem bly of equip ment and as so ci ated wir ing that to gether forms a com plete

sys tem or in de pend ent sub sys tem.

unity cou pling In a the o ret i cally per fect trans former, com plete elec tro mag netic cou -

pling be tween the pri mary and sec ond ary wind ings with no loss of power.

unity gain An am pli fier or ac tive cir cuit in which the out put am pli tude is the same as

the in put am pli tude.

unity power fac tor A power fac tor of 1.0, which means that the load is--in ef fect--a

pure re sis tance, with ac volt age and cur rent com pletely in phase.

unterminated A de vice or sys tem that is not ter mi nated.

up-converter A fre quency trans la tion de vice in which the fre quency of the out put sig -

nal is greater than that of the in put sig nal. Such de vices are com monly found in mi -
cro wave ra dio and sat el lite sys tems.

uplink A trans mis sion sys tem for send ing ra dio sig nals from the ground to a sat el lite or

air craft.

up stream A de vice or sys tem placed ahead of other de vices or sys tems in a sig nal path.
use ful life The pe riod dur ing which a low, con stant fail ure rate can be ex pected for a

given de vice or sys tem. The

use ful life is the por tion of a prod uct life cy cle be tween

break-in and wear out.

user A per son, or ga ni za tion, or group that em ploys the ser vices of a sys tem for the

trans fer of in for ma tion or other pur poses.

V

VA An ab bre vi a tion for volt-amperes, volts times am peres.

vac uum re lay A re lay whose con tacts are en closed in an evac u ated space, usu ally to

pro vide re li able long-term op er a tion.

vac uum switch A switch whose con tacts are en closed in an evac u ated con tainer so that

spark for ma tion is dis cour aged.

vac uum tube An elec tron tube. The most com mon vac uum tubes in clude the di ode, tri -

ode, tetrode, and pentode.

va lid ity check A test de signed to en sure that the qual ity of trans mis sion is main tained

over a given sys tem.

varactor A semi con duc tor that be haves like a ca pac i tor un der the in flu ence of an ex -

ter nal con trol volt age.

varactor di ode A semi con duc tor de vice whose ca pac i tance is a func tion of the ap plied

volt age. A varactor di ode, also called a

vari able reactance di ode or sim ply a

varactor, is of ten used to tune the op er at ing fre quency of a ra dio cir cuit.

vari able fre quency os cil la tor (VFO) An os cil la tor whose fre quency can be set to any

re quired value in a given range of fre quen cies.

vari able im ped ance A ca pac i tor, in duc tor, or re sis tor that is ad just able in value.

vari able-gain am pli fier An am pli fier whose gain can be con trolled by an ex ter nal sig -

nal source.

vari able-reluctance A trans ducer in which the in put (usu ally a me chan i cal move -

ment) var ies the mag netic re luc tance of a de vice.

vari a tion mon i tor A de vice used for sens ing a de vi a tion in volt age, cur rent, or fre -

quency, which is ca pa ble of pro vid ing an alarm and/or ini ti at ing trans fer to an other
power source when pro grammed lim its of volt age, fre quency, cur rent, or time are
ex ceeded.

varicap A di ode used as a vari able ca pac i tor.

VCXO (volt age con trolled crys tal os cil la tor) A de vice whose out put fre quency is

de ter mined by an in put con trol volt age.

vec tor A quan tity hav ing both mag ni tude and di rec tion.
vec tor di a gram A di a gram us ing vec tors to in di cate the re la tion ship be tween volt age

and cur rent in a cir cuit.

vec tor sum The sum of two vec tors which, when they are at right an gles to each other,

equal the length of the hy pot e nuse of the right tri an gle so formed. In the gen eral
case, the vec tor sum of the two vec tors equals the di ag o nal of the par al lel o gram
formed on the two vec tors.

ve loc ity of light The speed of prop a ga tion of elec tro mag netic waves in a vac uum,

equal to 299,792,458 m/s, or ap prox i mately 186,000 mi/s. For rough cal cu la tions,
the fig ure of 300,000 km/s is used.

ve loc ity of prop a ga tion The ve loc ity of sig nal trans mis sion. In free space, elec tro -

mag netic waves travel at the speed of light. In a ca ble, the ve loc ity is sub stan tially
lower.

ver nier A de vice that en ables pre ci sion read ing of a mea sur ing set or gauge, or the set -

ting of a dial with pre ci sion.

very low fre quency (VLF) A ra dio fre quency in the band 3 kHz to 30 kHz.
ves ti gial side band A form of trans mis sion in which one side band is sig nif i cantly at -

ten u ated. The car rier and the other side band are trans mit ted with out at ten u a tion.

vi bra tion test ing A test ing pro ce dure whereby sub sys tems are mounted on a test base

that vi brates, thereby re veal ing any faults re sult ing from badly sol dered joints or
other poor me chan i cal de sign fea tures.

volt The stan dard unit of elec tro mo tive force, equal to the po ten tial dif fer ence be tween

two points on a con duc tor that is car ry ing a con stant cur rent of one am pere when the
power dis si pated be tween the two points is equal to one watt. One

volt is equiv a lent

to the po ten tial dif fer ence across a re sis tance of one ohm when one am pere is flow -
ing through it. The volt is named for the Ital ian phys i cist Alessandro Volta
(1745-1827).

volt-ampere (VA) The ap par ent power in an ac cir cuit (volts times am peres).

volt-ohm-milliammeter (VOM) A gen eral pur pose multirange test me ter used to

mea sure volt age, re sis tance, and cur rent.

volt age The po ten tial dif fer ence be tween two points.

volt age drop A de crease in elec tri cal po ten tial re sult ing from cur rent flow through a

re sis tance.

volt age gra di ent The con tin u ous drop in elec tri cal po ten tial, per unit length, along a

uni form con duc tor or thick ness of a uni form di elec tric.

volt age level The ra tio of the volt age at a given point to the volt age at an ar bi trary ref er -

ence point.

volt age ref er ence cir cuit A sta ble volt age ref er ence source.

volt age reg u la tion The de vi a tion from a nom i nal volt age, ex pressed as a per cent age of

the nom i nal volt age.

volt age reg u la tor A cir cuit used for con trol ling and main tain ing a volt age at a con stant

level.

volt age sta bi lizer A de vice that pro duces a con stant or sub stan tially con stant out put

volt age de spite vari a tions in in put volt age or out put load cur rent.

volt age to ground The volt age be tween any given por tion of a piece of equip ment and

the ground po ten tial.

volt me ter An in stru ment used to mea sure dif fer ences in elec tri cal po ten tial.
vox A voice-operated re lay cir cuit that per mits the equiv a lent of push-to-talk op er a tion

of a trans mit ter by the op er a tor.

VSAT (very small ap er ture ter mi nal) A sat el lite Ku-band earth sta tion in tended for

fixed or por ta ble use. The an tenna di am e ter of a VSAT is on the or der of 1.5 m or
less.

W

watt The unit of power equal to the work done at one joule per sec ond, or the rate of

work mea sured as a cur rent of one am pere un der an elec tric po ten tial of one volt.
Des ig nated by the sym bol

W, the watt is named af ter the Scot tish in ven tor James

Watt (1736-1819).

watt me ter A me ter in di cat ing in watts the rate of con sump tion of elec tri cal en ergy.

watt-hour The work per formed by one watt over a one hour pe riod.
wave A dis tur bance that is a func tion of time or space, or both, and is prop a gated in a

me dium or through space.

wave num ber The re cip ro cal of wave length; the num ber of wave lengths per unit dis -

tance in the di rec tion of prop a ga tion of a wave.

waveband A band of wave lengths de fined for some given pur pose.
wave form The char ac ter is tic shape of a pe ri odic wave, de ter mined by the fre quen cies

pres ent and their am pli tudes and rel a tive phases.

wave front A con tin u ous sur face that is a lo cus of points hav ing the same phase at a

given in stant. A

wave front is a sur face at right an gles to rays that pro ceed from the

wave source. The sur face passes through those parts of the wave that are in the same
phase and travel in the same di rec tion. For par al lel rays the wave front is a plane; for
rays that ra di ate from a point, the wave front is spher i cal.

wave guide Gen erally, a rect an gu lar or cir cu lar pipe that con strains the prop a ga tion of

an acous tic or elec tro mag netic wave along a path be tween two lo ca tions. The di -
men sions of a wave guide de ter mine the fre quen cies for op ti mum trans mis sion.

wave length For a si nu soi dal wave, the dis tance be tween points of cor re spond ing phase

of two con sec u tive cy cles.

weber The unit of mag netic flux equal to the flux that, when linked to a cir cuit of one

turn, pro duces an elec tro mo tive force of one volt as the flux is re duced at a uni form
rate to zero in one sec ond. The

weber is named for the Ger man phys i cist Wil helm

Eduard Weber (1804-1891).

weighted The con di tion when a cor rec tion fac tor is ap plied to a measurement.
weight ing The ad just ment of a mea sured value to ac count for con di tions that would

oth er wise be dif fer ent or ap pro pri ate dur ing a mea sure ment.

weight ing net work A cir cuit, used with a test in stru ment, that has a spec i fied am pli -

tude-versus-frequency char ac ter is tic.

wideband The pass ing or pro cess ing of a wide range of fre quen cies. The mean ing var -

ies with the con text.

Wien bridge An ac bridge used to mea sure ca pac i tance or in duc tance.
wind ing A coil of wire used to form an in duc tor.

wire A sin gle me tal lic con duc tor, usu ally solid-drawn and cir cu lar in cross sec tion.

work ing range The per mit ted range of val ues of an an a log sig nal over which trans mit -

ting or other pro cess ing equip ment can op er ate.

work ing volt age The rated volt age that may safely be ap plied con tin u ously to a given

cir cuit or de vice.

X

x-band A mi cro wave fre quency band from 5.2 GHz to 10.9 GHz.

x-cut A method of cut ting a quartz plate for an os cil la tor, with the x-axis of the crys tal

per pen dic u lar to the faces of the plate.

X ray An elec tro mag netic ra di a tion of ap prox i mately 100 nm to 0.1 nm, ca pa ble of

pen e trat ing non me tal lic ma te ri als.

Y

y-cut A method of cut ting a quartz plate for an os cil la tor, with the y-axis of the crys tal

per pen dic u lar to the faces of the plate.

yield strength The mag ni tude of me chan i cal stress at which a ma te rial will be gin to de -

form. Be yond the

yield strength point, ex ten sion is no lon ger pro por tional to stress

and rup ture is pos si ble.

yoke A ma te rial that in ter con nects mag netic cores. Yoke can also re fer to the de flec tion

wind ings of a CRT.

yt trium-iron gar net (YIG) A crys tal line ma te rial used in mi cro wave de vices.

Whitaker, Jerry C. "

Index of Figures"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

Index of Figures

Power Vac uum Tube Ap pli ca tions

Fig ure 1.1 Pat ent draw ing of a mul ti stage de pressed-collector kly stron us ing six col -

lec tor el e ments (num bered 1621). 7

Fig ure 1.2 Vac uum tube op er at ing pa ram e ters based on power and fre quency. 13
Fig ure 1.3 Op er ating fre quency bands for shortwave broad cast ing. 14

Fig ure 1.4 Com pos ite base band ste reo FM sig nal. A full left-only or right-only sig nal

will mod u late the main (L+R) chan nel to a max i mum of 45 per cent. The ste reo -
phonic subchannel is com posed of up per side band (USB) and lower side band (LSB)
com po nents. 16

Fig ure 1.5 Prin ci pal el e ments of a sat el lite com mu ni ca tions link. 19

Fig ure 1.6 The power lev els in trans mis sion of a video sig nal via sat el lite. 20
Fig ure 1.7 Sim plified block di a gram of a pulsed ra dar sys tem. 21

Fig ure 1.8 Di rec tion-finding er ror re sult ing from bea con re flec tions. 25

Fig ure 1.9 The con cept of two-way dis tance rang ing. 26

Fig ure 1.10 The con cept of dif fer en tial dis tance rang ing (hy per bolic). 27
Fig ure 1.11 The con cept of one-way dis tance rang ing. 27

Fig ure 1.12 Ba sic sche matic of a 20 kW in duc tion heater cir cuit. 30

Fig ure 1.13 The elec tro mag netic spec trum. 31

Mod u la tion Sys tems and Char ac ter is tics

Fig ure 2.1 Char ac ter is tics of a se ries res o nant cir cuit as a func tion of fre quency for a

con stant ap plied volt age and dif fer ent cir cuit

Qs: (a) mag ni tude, (b) phase an gle. 40

Fig ure 2.2 Char ac ter is tics of a par al lel res o nant cir cuit as a func tion of fre quency for

dif fer ent cir cuit

Qs: (a) mag ni tude, (b) phase an gle. 41

Fig ure 2.3 Cy lin dri cal res o na tor in cor po rat ing a cou pling loop: (a) ori en ta tion of loop

with re spect to cav ity, (

b) equiv a lent cou pled cir cuit. 44

Fig ure 2.4 Cross sec tion of a quartz crys tal taken in the plane per pen dic u lar to the op ti -

cal axis: (

a) Y-cut plate, (b) X-cut plate. 44

Fig ure 2.5 The ef fects of tem per a ture on two types of AT-cut crys tals. 45

Fig ure 2.6 Equiv a lent net work of a pi ezo elec tric res o na tor. 46

Fig ure 2.7 Fre quency vs. tem per a ture char ac ter is tics for a typ i cal tem per a -

ture-compensated crys tal os cil la tor. 48

Fig ure 2.8 Plate ef fi ciency as a func tion of con duc tion an gle for an am pli fier with a

tuned load. 51

Fig ure 2.9 A plot of sig nal power vs. fre quency for white noise and fre quency lim ited

noise. 54

Fig ure 2.10 Fre quency-domain rep re sen ta tion of an am pli tude-modulated sig nal at

100 per cent mod u la tion.

E

c

= car rier power,

F

c

= fre quency of the car rier, and

F

m

=

fre quency of the mod u lat ing sig nal. 56

Fig ure 2.11 Time-domain rep re sen ta tion of an am pli tude-modulated sig nal. Mod u la -

tion at 100 per cent is de fined as the point at which the peak of the wave form reaches
twice the car rier level, and the min i mum point of the wave form is zero. 57

Fig ure 2.12 Sim plified di a gram of a high-level am pli tude-modulated am pli fier. 59

Fig ure 2.13 Ide al ized am pli tude char ac ter is tics of the FCC stan dard wave form for

mono chrome and color TV trans mis sion. 60

Fig ure 2.14 Types of sup pressed car rier am pli tude mod u la tion: (a) the mod u lat ing

sig nal, (

b) dou ble-sideband AM sig nal, (c) dou ble-sideband sup pressed car rier AM,

(

d) sin gle-sideband sup pressed car rier AM. 61

Fig ure 2.15 Sim plified QAM mod u la tor. 62
Fig ure 2.16 Sim plified QAM de mod u lator. 63

Fig ure 2.17 Plot of Bessel func tions of the first kind as a func tion of mod u la tion in dex.

65

Fig ure 2.18 RF spec trum of a fre quency-modulated sig nal with a mod u la tion in dex of

5 and other op er at ing pa ram e ters as shown. 67

Fig ure 2.19 RF spec trum of a fre quency-modulated sig nal with a mod u la tion in dex of

15 and op er at ing pa ram e ters as shown. 67

Fig ure 2.20 Preemphasis curves for time con stants of 50, 75, and 100

µ

s. 69

Fig ure 2.21 Block di a gram of an FM ex citer. 70

Fig ure 2.22 Sim plified reactance mod u la tor. 71

Fig ure 2.23 Equiv a lent cir cuit of the reactance mod u la tor. 72
Fig ure 2.24 Vec tor di a gram of a reactance mod u la tor pro duc ing FM. Note: V

g

=

V

C2

. 72

Fig ure 2.25 Volt age-controlled di rect-FM mod u la tor. 72

Fig ure 2.26 Me chan i cal lay out of a com mon type of

1

4

-wave PA cav ity for FM ser -

vice. 73

Fig ure 2.27 Pulse am pli tude mod u la tion wave forms: (a) mod u lat ing sig nal; (b)

square-topped sam pling, bi po lar pulse train; (

c) topped sam pling, bi po lar pulse

train; (

d) square-topped sam pling, uni po lar pulse train; (e) top sam pling, uni po lar

pulse train. 76

Fig ure 2.28 Pulse time mod u la tion wave forms: (a) mod u lat ing sig nal and sam -

ple-and-hold (S/H) wave forms, (

b) sawtooth wave form added to S/H, (c) lead -

ing-edge PTM, (

d) trail ing-edge PTM. 77

Fig ure 2.29 Pulse fre quency mod u la tion. 78

Fig ure 2.30 The quantization pro cess. 78

Fig ure 2.31 Delta mod u la tion wave forms: (a) mod u lat ing sig nal, (b) quantized mod u -

lat ing sig nal, (

c) pulse train, (d) re sult ing delta mod u la tion wave form. 80

Fig ure 2.32 Bi nary FSK wave form. 81

Fig ure 2.33 Bi nary PSK wave form. 81
Fig ure 2.34 Var i ous base band mod u la tion for mats: (a) non-return-to zero, (b) uni po -

lar-return-to-zero, (

c) dif fer en tial en coded (NRZ-mark), (d) split phase. 83

Vac uum Tube Prin ci ples

Fig ure 3.1 Vari a tion of elec tron emis sion as a func tion of ab so lute tem per a ture for a

thoriated-tungsten emit ter. 91

Fig ure 3.2 Com mon types of heater and cath ode struc tures. 91
Fig ure 3.3 Ra tio of sec ond ary emis sion cur rent to pri mary cur rent as a func tion of pri -

mary elec tron ve loc ity. 92

Fig ure 3.4 Vac uum di ode: (a) di rectly heated cath ode, (b) in di rectly heated cath ode.

93

Fig ure 3.5 An ode cur rent as a func tion of an ode volt age in a two-electrode tube for

three cath ode tem per a tures. 93

Fig ure 3.6 Me chan i cal con fig u ra tion of a power tri ode. 95

Fig ure 3.7 Con stant-current char ac ter is tics for a tri ode with a

µ

of 12. 97

Fig ure 3.8 Con stant-current char ac ter is tics for a tri ode with a

µ

of 20. 98

Fig ure 3.9 Grounded-grid con stant-current char ac ter is tics for a zero-bias tri ode with a

µ

of 200. 98

Fig ure 3.10 In ter nal con fig u ra tion of a pla nar tri ode. 99
Fig ure 3.11 Tetrode plate cur rent char ac ter is tics. Plate cur rent is plot ted as a func tion

of plate volt age, with grid volt ages as shown. 100

Fig ure 3.12 Ra dial beam power tetrode (4CX15000A). 101
Fig ure 3.13 Prin ci pal di men sions of 4CX15000A tetrode. 103
Fig ure 3.14 5CX1500B power pentode de vice. 105

Fig ure 3.15 Prin ci pal di men sions of 5CX1500B pentode. 108

Fig ure 3.16 Con tin u ous wave out put power ca pa bil ity of a gridded vac uum tube. 109
Fig ure 3.17 Per for mance of a class C am pli fier as the op er at ing fre quency is in creased

be yond the de sign lim its of the vac uum tube. 109

Fig ure 3.18 Tran sit-time ef fects in a class C am pli fier: (a) con trol grid volt age, (b)

elec tron po si tion as a func tion of time (tri ode case), (

c) elec tron po si tion as a func -

tion of time (tetrode case), (

d) plate cur rent (tri ode case). 111

Fig ure 3.19 The re la tion ship be tween an ode dis si pa tion and cool ing method. 113

Fig ure 3.20 A typ i cal trans mit ter PA stage cool ing sys tem. 113
Fig ure 3.21 Wa ter-cooled an ode with grooves for con trolled wa ter flow. 114

Fig ure 3.22 Typ i cal va por-phase cool ing sys tem. 115

Fig ure 3.23 Va por-phase cool ing sys tem for a 4-tube trans mit ter us ing a com mon wa -

ter sup ply. 116

Fig ure 3.24 Photo of a typ i cal ox ide cath ode. 118

Fig ure 3.25 Spi ral-type tung sten fil a ment. 120

Fig ure 3.26 Bar-type tung sten fil a ment. 120
Fig ure 3.27 Mesh tung sten fil a ment. 121

Fig ure 3.28 Mesh grid struc ture. 122

Fig ure 3.29 Pyrolytic graph ite grid: (a) be fore la ser pro cess ing, (b) com pleted as sem -

bly. 124

Fig ure 3.30 The screen grid as sem bly of a typ i cal tetrode PA tube. 125

Fig ure 3.31 The in ter nal ar range ment of the an ode, screen, con trol grid, and cath ode

as sem blies of a tetrode power tube. 125

Fig ure 3.32 Cir cuit for pro vid ing a low-impedance screen sup ply for a tetrode. 126

Fig ure 3.33 Se ries power sup ply con fig u ra tion for swamp ing the screen cir cuit of a

tetrode. 127

Fig ure 3.34 Typ i cal curve of grid cur rent as a func tion of con trol grid volt age for a

high-power thoriated-tungsten fil a ment tetrode. 128

Fig ure 3.35 Sec ond ary emis sion char ac ter is tics of var i ous met als un der or di nary con -

di tions. 128

Fig ure 3.36 A cutaway view of the an ode struc ture of an RF power am pli fier tube. 130

Fig ure 3.37 Ther mal con duc tiv ity of com mon ce ram ics. 131

Fig ure 3.38 Cross sec tion of the base of a tetrode socket show ing the con nec tion

points. 134

Fig ure 3.39 The el e ments in volved in the neu tral iza tion of a tetrode PA stage. 135
Fig ure 3.40 A graph i cal rep re sen ta tion of the el e ments in volved in the

self-neutralization of a tetrode RF stage. 136

Fig ure 3.41 Push-pull grid neu tral iza tion. 137
Fig ure 3.42 Sym met ri cal stage neu tral iza tion for a grounded-grid cir cuit. 138
Fig ure 3.43 Push-pull grid neu tral iza tion in a sin gle-ended tetrode stage. 138

Fig ure 3.44 Sin gle-ended grid neu tral iza tion for a tetrode. 139

Fig ure 3.45 The pre vi ous fig ure re drawn to show the el e ments in volved in neu tral iza -

tion. 139

Fig ure 3.46 Sin gle-ended plate neu tral iza tion. 140

Fig ure 3.47 Sin gle-ended plate neu tral iza tion show ing the ca pac i tance bridge pres ent.

140

Fig ure 3.48 A grounded-screen PA stage neu tral ized through the use of stray in duc -

tance be tween the screen con nec tor and the cav ity deck: (

a) sche matic di a gram, (b)

me chan i cal lay out of cav ity. 142

Fig ure 3.49 Cir cuit of grounded-grid am pli fier hav ing grid im ped ance and neu tral ized

for re ac tive cur rents. 143

Fig ure 3.50 Neu tral iza tion by cross-connected ca pac i tors of a sym met ri cal cath -

ode-excited am pli fier wth com pen sa tion of lead in duc tance. 144

Fig ure 3.51 Graph i cal rep re sen ta tion of the el e ments of a tetrode when

self-neutralized. 145

Fig ure 3.52 Com po nents of the out put volt age of a tetrode when neu tral ized by add ing

se ries screen-lead ca pac i tance. 146

Fig ure 3.53 Com po nents of the out put volt age of a tetrode when neu tral ized by add ing

ex ter nal grid-to-plate ca pac i tance. 146

Fig ure 3.54 Com po nents of the out put volt age of a tetrode neu tral ized by add ing in -

duc tance com mon to the screen and cath ode re turn. 147

De signing Vac uum Tube Cir cuits

Fig ure 4.1 Plate ef fi ciency as a func tion of con duc tion an gle for an am pli fier with a

tuned load. 152

Fig ure 4.2 Ba sic grounded-cathode RF am pli fier cir cuit. 153
Fig ure 4.3 Typ i cal am pli fier cir cuit us ing a zero-bias tri ode. Grid cur rent is mea sured

in the re turn lead from ground to the fil a ment. 154

Fig ure 4.4 Vari a tion of plate volt age as a func tion of grid volt age. 155
Fig ure 4.5 Re la tion ship be tween grid volt age and plate volt age plot ted on a con -

stant-current curve. 155

Fig ure 4.6 In stan ta neous val ues of plate and grid cur rent as a func tion of time. 156

Fig ure 4.7 Sche matic of a tetrode RF power am pli fier show ing tube el e ment by pass

ca pac i tors. 162

Fig ure 4.8 Sche matic of a tri ode RF power am pli fier show ing tube el e ment by pass ca -

pac i tors: (

a) con trol grid at dc ground po ten tial, (b) con trol grid by passed to ground.

163

Fig ure 4.9 Interelectrode ca pac i tances sup port ing VHF par a sitic os cil la tion in a

high-frequency RF amplfier. 164

Fig ure 4.10 Place ment of par a sitic suppressors to elim i nate VHF par a sitic os cil la tions

in a high-frequency RF am pli fier: (

a) re sis tor-coil com bi na tion, (b) par a sitic choke.

165

Fig ure 4.11 A sec tion of hon ey comb shield ing ma te rial used in an RF am pli fier. 168

Fig ure 4.12 A se lec tion of com mon fingerstock. 169

Fig ure 4.13 Shorted

1

4

-wave length line: (

a) phys i cal cir cuit, (b) elec tri cal equiv a lent.

172

Fig ure 4.14 Shorted line less than

1

4

wave length: (

a) phys i cal cir cuit, (b) elec tri cal

equiv a lent. 172

Fig ure 4.15 A sim pli fied rep re sen ta tion of the grid in put cir cuit of a PA tube. 174
Fig ure 4.16 Three com mon meth ods for RF by pass ing of the cath ode of a tetrode PA

tube: (

a) ground ing the cath ode be low 30 MHz, (b) ground ing the cath ode above 30

MHz, (

c) ground ing the cath ode via a

1

2

-wave trans mis sion line. 175

Fig ure 4.17 RF cir cu lat ing cur rent path be tween the plate and screen in a tetrode PA

tube. 176

Fig ure 4.18 The phys i cal lay out of a com mon type of

1

4

-wave length PA cav ity de -

signed for VHF op er a tion. 177

Fig ure 4.19 The RF cir cu lat ing cur rent paths for the

1

4

-wave length cav ity shown in

Fig ure 4.18. 178

Fig ure 4.20 Graph of the RF cur rent, RF volt age, and RF im ped ance for a

1

4

-wave -

length shorted trans mis sion line. At the feed point, RF cur rent is zero, RF volt age is
max i mum, and RF im ped ance is in fi nite. 178

Fig ure 4.21 Graph of the RF cur rent, RF volt age, and RF im ped ance pro duced by the

phys i cally fore short ened co ax ial trans mis sion line cav ity. 179

Fig ure 4.22 Ba sic load ing mech a nism for a VHF

1

4

-wave cav ity. 180

Fig ure 4.23 The

1

2

-wave length PA cav ity in its ba sic form. 181

Fig ure 4.24 The dis tri bu tion of RF cur rent, volt age, and im ped ance along the in ner

con duc tor of a

1

2

-wave length cav ity. 182

Fig ure 4.25 The con fig u ra tion of a prac ti cal

1

2

-wave length PA cav ity. 183

Fig ure 4.26 Coarse-tuning mech a nisms for a

1

2

-wave cav ity. 184

Fig ure 4.27 Using the cav ity's mov able sec tion to ad just for res o nance: (a) the ro tary

sec tion at max i mum height, (

b) the ro tary sec tion at min i mum height. 185

Fig ure 4.28 The ba sic de sign of a folded

1

2

-wave length cav ity. 186

Fig ure 4.29 Dou ble-tuned overcoupled broad band cav ity am pli fier. 188

Fig ure 4.30 The use of in duc tive cou pling in a

1

4

-wave length PA stage. 189

Fig ure 4.31 Top view of a cav ity show ing the in duc tive cou pling loop ro tated away

from the axis of the mag netic field of the cav ity. 189

Fig ure 4.32 Cutaway view of the cav ity show ing the cou pling loop ro tated away from

the axis of the mag netic field. 190

Fig ure 4.33 A

1

4

-wave length cav ity with ca pac i tive cou pling to the out put load. 191

Fig ure 4.34 The equiv a lent cir cuit of a

1

4

-wave length cav ity am pli fier with ca pac i tive

cou pling. 192

Fig ure 4.35 The equiv a lent cir cuit of a

1

4

-wave length cav ity am pli fier show ing how

se ries ca pac i tive cou pling ap pears elec tri cally as a par al lel cir cuit to the PA tube.
192

Fig ure 4.36 Typ i cal VHF cav ity am pli fier. 194

Fig ure 4.37 Elec tri cal equiv a lent of the cav ity am pli fier shown in Fig ure 4.36. 195

Fig ure 4.38 VHF cav ity am pli fi ers: (a, above) cross-sectional view of a broad band de -

sign, (

b, be low) cav ity de signed for tele vi sion ser vice, (c, next page) FM broad cast

cav ity am pli fier. 196

Fig ure 4.39 Con ven tional ca pac i tor in put fil ter full-wave bridge. 198

Fig ure 4.40 A por tion of a high-voltage se ries-connected rec ti fier stack. 199

Fig ure 4.41 A 3-phase delta-connected high-voltage rec ti fier. 199
Fig ure 4.42 High-voltage rec ti fier as sem bly for a 3-phase delta-connected cir cuit. 200
Fig ure 4.43 Using buildout resistances to force cur rent shar ing in a par al lel rec ti fier

as sem bly. 201

Fig ure 4.44 In verse-parallel thy ris tor ac power con trol: (a) cir cuit di a gram, (b) volt -

age and cur rent wave forms for full and re duced thy ris tor con trol an gles. The wave -
forms ap ply for a purely re sis tive load. 202

Fig ure 4.45 Wave forms in an ac cir cuit us ing thy ris tor power con trol. 203
Fig ure 4.46 Volt age and cur rent wave forms for in verse-parallel thy ris tor power con -

trol with an in duc tive load: (

a) full con duc tion, (b) small-angle phase re duc tion, (c)

large-angle phase re duc tion. 204

Fig ure 4.47 Mod ified full-thyristor 3-phase ac con trol of an in duc tive delta load. 205
Fig ure 4.48 A 6-phase boost rec ti fier cir cuit. 206
Fig ure 4.49 Thy ris tor-controlled high-voltage servo power sup ply. 207

Fig ure 4.50 Phase-controlled power sup ply with pri mary reg u la tion. 208

Fig ure 4.51 Sim plified block di a gram of the gating cir cuit for a phase-control sys tem

us ing back-to-back SCRs. 209

Fig ure 4.52 Ba sic 3-phase rec ti fier cir cuits: (a) half-wave wye, (b) full-wave bridge,

(

c) 6-phase star, (d) 3-phase dou ble-wye. 211

Fig ure 4.53 Volt age and cur rent waveshapes for an in duc tive-input fil ter driven by a

3-phase sup ply. 214

Fig ure 4.54 Char ac ter is tics of a ca pac i tive in put fil ter cir cuit: (a) sche matic di a gram,

(

b) volt age waveshape across the in put ca pac i tor, (c) waveshape of cur rent flow ing

through the di odes. 217

Fig ure 4.55 Plate volt age sam pling cir cuits: (a) sin gle-ended, (b) dif fer en tial. 218

Fig ure 4.56 Dif fer en tial am pli fier cir cuits: (a) ba sic cir cuit, (b) in stru men ta tion am -

pli fier. 219

Fig ure 4.57 Trans mit ter high-voltage power sup ply show ing re mote plate cur rent and

volt age sens ing cir cuits. 220

Ap plying Vac uum Tube De vices

Fig ure 5.1 The clas sic plate-modulated PA cir cuit for AM ap pli ca tions. 224
Fig ure 5.2 Ba sic meth ods of screen and plate mod u la tion: (a) plate mod u la tion, (b)

plate and screen mod u la tion, (

c) plate mod u la tion us ing sep a rate plate/screen sup -

plies. 225

Fig ure 5.3 Grid mod u la tion of a class C RF am pli fier. 226

Fig ure 5.4 Cath ode mod u la tion of a class C RF am pli fier. 227
Fig ure 5.5 Clas sic vac uum tube class C am pli fier: (a) rep re sen ta tive wave forms, (b)

sche matic di a gram. Con di tions are as fol lows: si nu soi dal grid drive, 120° an ode
cur rent, res o nant an ode load. 228

Fig ure 5.6 Class B push-pull mod u la tor stage for a high-level AM am pli fier. 229

Fig ure 5.7 The pulse du ra tion mod u la tion (PDM) method of pulse width mod u la tion.

230

Fig ure 5.8 The prin ci pal wave forms of the PDM sys tem. 231

Fig ure 5.9 Block di a gram of a PDM wave form gen er a tor. 232

Fig ure 5.10 Spec trum at the out put of a non lin ear de vice with an in put of two

equal-amplitude sine waves of

f

1

= 2.001 MHz and

f

2

= 2.003 MHz. 235

Fig ure 5.11 Ideal grid-plate trans fer curve for class AB op er a tion. 236

Fig ure 5.12 Sim plified sche matic di a gram of the Chireix outphasing mod u lated am -

pli fier: (

a) out put cir cuit, (b) drive sig nal gen er a tion cir cuit. 238

Fig ure 5.13 Doherty high-efficiency AM am pli fier: (a) op er at ing the ory, (b) sche -

matic di a gram of a typ i cal sys tem. 239

Fig ure 5.14 Screen-modulated Doherty-type cir cuit. 241
Fig ure 5.15 Prac ti cal im ple men ta tion of a screen-modulated Doherty-type 1 MW

power am pli fier (Con ti nen tal Elec tron ics). 242

Fig ure 5.16 Terman-Woodyard high-efficiency mod u lated am pli fier. 243
Fig ure 5.17 Dome high-efficiency mod u lated am pli fier. 244
Fig ure 5.18 Ba sic block di a gram of a TV trans mit ter. The three ma jor sub as sem blies

are the ex citer, IPA, and PA. The power sup ply pro vides op er at ing volt ages to all
sec tions, and high volt age to the PA stage. 247

Fig ure 5.19 Sche matic di a gram of a 60 kW kly stron-based TV trans mit ter. 249
Fig ure 5.20 Sche matic di a gram of the mod u la tor sec tion of a 60 kW TV trans mit ter.

250

Fig ure 5.21 Cath ode-driven tri ode power am pli fier. 251
Fig ure 5.22 Grid-driven grounded-screen tetrode power am pli fier. 254

Fig ure 5.23 In put match ing cir cuits: (a) in duc tive in put match ing, (b) ca pac i tive in put

match ing. 256

Fig ure 5.24 Interstage RF cou pling cir cuits: (a) vari able ca pac i tance match ing, (b)

vari able tank match ing, (

c) vari able pi match ing, (d) fixed broad band im ped -

ance-transformation net work. 257

Fig ure 5.25 Two com mon ap proaches to au to matic fre quency con trol of a mag ne tron

os cil la tor. 259

Fig ure 5.26 Dig i tal syn chro nizer sys tem for ra dar ap pli ca tions. 260

Fig ure 5.27 A line-type mod u la tor for ra dar. 261
Fig ure 5.28 Ac tive-switch mod u la tor cir cuits: (a) di rect-coupled sys tem, (b) ca pac i -

tor-coupled, (

c) trans former-coupled, (d) ca pac i tor- and trans former-coupled. 262

Fig ure 5.29 A mag netic mod u la tor cir cuit. 262

Mi cro wave Power Tubes

Fig ure 6.1 Mi cro wave power tube type as a func tion of fre quency and out put power.

267

Fig ure 6.2 Types of lin ear-beam mi cro wave tubes. 268
Fig ure 6.3 Sche matic di a gram of a lin ear-beam tube. 268

Fig ure 6.4 Types of crossed-field mi cro wave tubes. 269

Fig ure 6.5 Mag ne tron elec tron path look ing down into the cav ity with the mag netic

field ap plied. 269

Fig ure 6.6 Fam ily tree of the dis trib uted emis sion crossed-field am pli fier (CFA). 270

Fig ure 6.7 Cross sec tion of a 7289 pla nar tri ode. 272
Fig ure 6.8 Grounded-grid equiv a lent cir cuits: (a) low-frequency op er a tion, (b) mi cro -

wave-frequency op er a tion. The cath ode-heating and grid-bias cir cuits are not
shown. 273

Fig ure 6.9 Cutaway view of the tetrode (left) and the Diacrode (right). Note that the RF

cur rent peaks above and be low the Diacrode cen ter while on the tetrode there is only
one peak at the bot tom. 274

Fig ure 6.10 The el e ments of the Diacrode, in clud ing the up per cav ity. Dou ble cur rent,

and con se quently dou ble power, is achieved with the de vice be cause of the cur rent
peaks at the top and bot tom of the tube, as shown. 275

Fig ure 6.11 Sche matic rep re sen ta tion of a re flex kly stron. 276
Fig ure 6.12 Po si tion-time curves of elec trons in the an ode-repeller space, show ing the

ten dency of the elec trons to bunch around the elec tron pass ing through the an ode at
the time when the al ter nat ing gap volt age is zero and be com ing neg a tive. 277

Fig ure 6.13 Sche matic cross sec tion of a re flex kly stron os cil la tor. 279

Fig ure 6.14 Cross sec tion of a clas sic two-cavity kly stron os cil la tor. 280

Fig ure 6.15 Prin ci pal el e ments of a kly stron. 282
Fig ure 6.16 Kly stron cav ity: (a) phys i cal ar range ment, (b) equiv a lent cir cuit. 283
Fig ure 6.17 Bunching ef fect of a multicavity kly stron. 284

Fig ure 6.18 Di ode sec tion of a kly stron elec tron gun. 285

Fig ure 6.19 Beam form ing in the di ode sec tion of a kly stron elec tron gun. 285
Fig ure 6.20 Kly stron beam-current vari a tion as a func tion of emit ter tem per a ture. 286
Fig ure 6.21 Mod u lating-anode elec trode in a multicavity kly stron. 287

Fig ure 6.22 Beam-current vari a tion as a func tion of mod u lat ing-anode volt age. 288

Fig ure 6.23 The ef fect of a mag netic field on the elec tron beam of a kly stron. 288
Fig ure 6.24 Field pat tern of a kly stron elec tro mag net. 289

Fig ure 6.25 Field pat tern and beam shape in a prop erly ad justed mag netic field. 289

Fig ure 6.26 Dis tor tion of the field pat tern and beam shape due to mag netic ma te rial in

the mag netic field. 290

Fig ure 6.27 Kly stron cav ity: (a) phys i cal el e ment, (b) equiv a lent elec tric cir cuit. 290

Fig ure 6.28 Ca pac i tance-tuned kly stron cav ity: (a) phys i cal el e ment, (b) equiv a lent

cir cuit. 291

Fig ure 6.29 Sche matic equiv a lent cir cuit of a four-cavity kly stron. 292

Fig ure 6.30 Kly stron loop cou pling: (a) me chan i cal ar range ment, (b) equiv a lent elec -

tric cir cuit. 292

Fig ure 6.31 Plot of the rel a tive phase of the ref er ence elec trons as a func tion of ax ial

dis tance in a high-efficiency kly stron. 293

Fig ure 6.32 Plot of the nor mal ized RF beam cur rents as a func tion of dis tance along

the length of a high-efficiency kly stron. 293

Fig ure 6.33 Typ i cal gain, out put power, and drive re quire ments for a kly stron. 294

Fig ure 6.34 Kly stron beam cur rent as a func tion of an nu lar con trol elec trode volt age.

297

Fig ure 6.35 Block di a gram of a switch ing mod-anode pulser. 297

Fig ure 6.36 Me chan i cal de sign of the mul ti stage de pressed col lec tor as sem bly. Note

the "V" shape of the 4-element sys tem. 301

Fig ure 6.37 A par tially as sem bled MSDC col lec tor. 302
Fig ure 6.38 Beam cur rent as a func tion of an nu lar con trol elec trode volt age for an

MSDC kly stron. 302

Fig ure 6.39 Col lec tor elec tron tra jec to ries for the car rier-only con di tion. Note that

nearly all elec trons travel to the last elec trode (4), pro duc ing elec trode cur rent

I

4

.

303

Fig ure 6.40 Col lec tor tra jec to ries at 25 per cent sat u ra tion. With the ap pli ca tion of

mod u la tion, the elec trons be gin to sort them selves out. 303

Fig ure 6.41 Col lec tor elec tron tra jec to ries at 50 per cent sat u ra tion, ap prox i mately the

blank ing level. The last three elec trodes (2, 3, and 4) share elec trons in a pre dict able
man ner, pro duc ing cur rents

I

2

,

I

3

, and

I

4

. 304

Fig ure 6.42 Col lec tor tra jec to ries at 90 per cent sat u ra tion, the sync level. Note the sig -

nif i cant in crease in the num ber of elec trons at tracted to the first elec trode, pro duc -
ing

I

1

. 304

Fig ure 6.43 The dis tri bu tion of elec trode cur rent as a func tion of drive power. Note the

sig nif i cant drop in

I

4

as drive power is in creased.

I

5

is the elec trode at cath ode po ten -

tial. 305

Fig ure 6.44 The MSDC col lec tor as sem bly with the pro tec tive shield par tially re -

moved. 306

Fig ure 6.45 Me chan i cal con struc tion of an ex ter nal-cavity MSDC tube: ( a) de vice, (b)

de vice in cav ity as sem bly. 306

Fig ure 6.46 Par al lel con fig u ra tion of the MSDC power sup ply. Each sup ply sec tion

has an out put volt age that is an in te gral mul ti ple of 6.125 kV. 307

Fig ure 6.47 Se ries con fig u ra tion of the MSDC power sup ply, which uses four iden ti -

cal 6.125 kV power sup plies con nected in se ries to achieve the needed volt ages. 308

Fig ure 6.48 De vice band width as a func tion of fre quency and drive power. Beam volt -

age is 24.5 kV, and beam cur rent is 5.04 A for an out put power of 64 kW. These
traces rep re sent the full-power test of the MSDC de vice. 309

Fig ure 6.49 De vice band width in the beam-pulsing mode as a func tion of fre quency

and drive power. Beam volt age is 24.5 kV, and beam cur rent is 3.56 A with an out -
put power of 34.8 kW. 309

Fig ure 6.50 Sim plified block di a gram of a 60 kW (TVT) MSDC trans mit ter. The au ral

kly stron may uti lize a con ven tional or MSDC tube at the dis cre tion of the user. 311

Fig ure 6.51 Sim plified sche matic di a gram of the power sup ply for an MSDC kly stron

trans mit ter. A sin gle high-voltage trans former with mul ti ple taps is used to provide
the needed col lec tor volt age po ten tials. 311

Fig ure 6.52 Sim plified sche matic di a gram of the Klystrode tube. 313
Fig ure 6.53 A 60 kW Klystrode tube mounted in its sup port stand with the out put cav -

ity at tached. 313

Fig ure 6.54 A close-up view of the dou ble-tuned out put cav ity of a 60 kW Klystrode

tube. 314

Fig ure 6.55 Over all struc ture of the IOT gun and out put cav ity. 315

Fig ure 6.56 Pyrolytic graph ite grid: (a) shell, (b) grid blank, (c) grid cut. 316

Fig ure 6.57 IOT in put cav ity. 317
Fig ure 6.58 IOT out put cav ity. 318

Fig ure 6.59 Sim plified block di a gram of the crow bar cir cuit de signed into the

Klystrode SK-series (Comark) trans mit ters. The cir cuit is in tended to pro tect the
tube from po ten tially dam ag ing fault cur rents. 319

Fig ure 6.60 Sche matic over view of the MSDC IOT or con stant ef fi ciency am pli fier.

320

Fig ure 6.61 Cal cu lated ef fi ciency of an MSDC IOT com pared to an un mod i fied IOT.

322

Fig ure 6.62 Ba sic el e ments of a trav el ing wave tube. 323

Fig ure 6.63 Mag netic fo cus ing for a TWT: (a) so le noid-type, (b) per ma -

nent-magnet-type, (

c) pe ri odic per ma nent-magnet struc ture. 324

Fig ure 6.64 He lix struc tures for a TWT: (a) ring-loop cir cuit, (b) ring-bar cir cuit. 325
Fig ure 6.65 Cou pled-cavity in ter ac tion struc tures: (a) for ward fun da men tal cir cuit or

"cloverleaf," (

b) sin gle-slot space har monic cir cuit. 326

Fig ure 6.66 Ge neric TWT elec tron gun struc ture. 327
Fig ure 6.67 Cross-sectional view of a TWT elec tron gun with a curved hot cath ode.

328

Fig ure 6.68 Ge neric con fig u ra tion of a trav el ing wave tube. 328

Fig ure 6.69 Cross-sectional view of var i ous col lec tor con fig u ra tions for a TWT. 329
Fig ure 6.70 Power sup ply con fig u ra tion for a mul ti stage de pressed col lec tor TWT.

330

Fig ure 6.71 Lin ear in jected-beam mi cro wave tube. 332
Fig ure 6.72 Reentrant emit ting-sole crossed-field am pli fier tube. 333
Fig ure 6.73 Con ven tional mag ne tron struc ture. 334

Fig ure 6.74 Cav ity mag ne tron os cil la tor: (a) cutaway view, (b) cross sec tion view per -

pen dic u lar to the axis of the cath ode. 335

Fig ure 6.75 Cav ity mag ne tron os cil la tor an ode: (a) hole-and-slot type, (b) slot type,

(

c) vane type. 335

Fig ure 6.76 Struc ture of a co ax ial mag ne tron. 336

Fig ure 6.77 Struc ture of a tun able co ax ial mag ne tron. 336
Fig ure 6.78 Cross-sectional view of a lin ear mag ne tron. 337

Fig ure 6.79 Func tional sche matic of the M-type ra dial BWO. 338

Fig ure 6.80 Strap-fed de vices: (a) strapped ra dial mag ne tron, (b) nonreentrant strap -

ping of a BWO. 339

Fig ure 6.81 Ba sic struc ture of a he li cal beam tube. 341

Fig ure 6.82 Func tional sche matic di a gram of the gyrotron: (a) sin gle coiled he li cal

beam gyrotron, (

b) elec tron tra jec tory of dou ble-coil he li cal beam gyrotron. 342

Fig ure 6.83 Ba sic struc ture of the gyroklystron am pli fier. 343

Fig ure 6.84 Ba sic struc ture of the gyroklystron trav el ing wave tube am pli fier. 343

Fig ure 6.85 Ba sic struc ture of the gyrotron back ward os cil la tor. 344
Fig ure 6.86 Ba sic struc ture of the gyrotwystron am pli fier. 344

Fig ure 6.87 Ba sic struc ture of the ubitron. 346

Fig ure 6.88 Func tional sche matic di a grams of mod i fied kly strons: (a) laddertron, (b)

twystron. 346

Fig ure 6.89 Pro tec tion and me ter ing sys tem for a kly stron am pli fier. 349

RF In ter con nec tion and Switching

Fig ure 7.1 Skin ef fect on an iso lated round con duc tor car ry ing a mod er ately high fre -

quency sig nal. 356

Fig ure 7.2 Semiflexible co ax ial ca ble: (a) a sec tion of ca ble show ing the ba sic con -

struc tion, (

b) ca ble with var i ous ter mi na tions. 358

Fig ure 7.3 Mag ni tude and power fac tor of line im ped ance with in creas ing dis tance

from the load for the case of a short-circuited re ceiver and a line with mod er ate at -
ten u a tion: (

a) volt age dis tri bu tion, (b) im ped ance mag ni tude, (c) im ped ance phase.

361

Fig ure 7.4 Power rat ing data for a va ri ety of co ax ial trans mis sion lines: (a) 50

line,

(

b) 75

line. 364

Fig ure 7.5 Ef fects of trans mis sion line pres sur iza tion on peak power rat ing. Note that

P

= the rat ing of the line at the in creased pres sure, and P = the rat ing of the line at at -

mo spheric pres sure. 366

Fig ure 7.6 At ten u a tion char ac ter is tics for a se lec tion of co ax ial ca bles: (a) 50

line,

(

b) 75

line. 367

Fig ure 7.7 Con ver sion chart show ing the re la tion ship be tween deci bel loss and ef fi -

ciency of a trans mis sion line: (

a) high-loss line, (b) low-loss line. 369

Fig ure 7.8 The vari a tion of co ax ial ca ble at ten u a tion as a func tion of am bi ent tem per a -

ture. 370

Fig ure 7.9 The ef fect of load VSWR on trans mis sion line loss. 370
Fig ure 7.10 Field con fig u ra tion of the dom i nant or TE

1,0

mode in a rect an gu lar wave -

guide: (

a) side view, ( b) end view, (c) top view. 374

Fig ure 7.11 Field con fig u ra tion of the dom i nant mode in cir cu lar wave guide. 375
Fig ure 7.12 Ridged wave guide: (a) sin gle-ridged, (b) dou ble-ridged. 376
Fig ure 7.13 The ef fects of par a sitic en ergy in cir cu lar wave guide: (a) trapped

cross-polarization en ergy, (

b) de layed trans mis sion of the trapped en ergy. 377

Fig ure 7.14 Phys i cal con struc tion of dou bly trun cated wave guide. 378
Fig ure 7.15 A probe con fig ured to in tro duce a re flec tion in a wave guide that is ad just -

able in mag ni tude and phase. 379

Fig ure 7.16 Wave guide ob struc tions used to in tro duce com pen sat ing re flec tions: (a)

in duc tive win dow, (

b) ca pac i tive win dow, (c) post (in duc tive) el e ment. 380

Fig ure 7.17 Wave guide stub el e ments used to in tro duce com pen sat ing re flec tions: (a)

se ries

tee el e ment, (b) shunt tee el e ment. 381

Fig ure 7.18 Cav ity res o na tor cou pling: (a) cou pling loop, (b) equiv a lent cir cuit. 383
Fig ure 7.19 Fil ter char ac ter is tics by type: (a) low-pass, (b) high-pass, (c) bandpass, (d)

bandstop. 385

Fig ure 7.20 Fil ter char ac ter is tics by align ment, third-order, all-pole fil ters: (a) mag ni -

tude, (

b) mag ni tude in deci bels. 386

Fig ure 7.21 The ef fects of fil ter or der on rolloff (Butterworth align ment). 387

Fig ure 7.22 Phys i cal model of a 90° hy brid combiner. 388

Fig ure 7.23 Op er ating prin ci ples of a hy brid combiner. This cir cuit is used to add two

iden ti cal sig nals at in puts A and B. 389

Fig ure 7.24 The ef fects of load VSWR on in put VSWR and iso la tion: (a) re spec tive

curves, (

b) cou pler sche matic. 391

Fig ure 7.25 The ef fects of power im bal ance at the in puts of a hy brid cou pler. 392
Fig ure 7.26 Phase sen si tiv ity of a hy brid cou pler. 392

Fig ure 7.27 Nonconstant-impedance branch di plex er. In this con fig u ra tion, two banks

of fil ters feed into a co ax ial tee. 393

Fig ure 7.28 Band-stop (notch) con stant-impedance di plex er mod ule. This de sign in -

cor po rates two 3 dB hy brids and fil ters, with a ter mi nat ing load on the iso lated port.
394

Fig ure 7.29 Bandpass con stant-impedance di plex er. 395
Fig ure 7.30 Sche matic di a gram of a six-module bandpass multi plexer. This con fig u -

ra tion ac com mo dates a split an tenna de sign and in cor po rates patch pan els for by -
pass pur poses. 397

Fig ure 7.31 Re verse-coupled

1

4

-wave hy brid combiner. 400

Fig ure 7.32 Com mon RF switch ing con fig u ra tions. 401

Fig ure 7.33 Hy brid switch ing con fig u ra tions: (a) phase set so that the com bined en -

ergy is de liv ered to port 4, (

b) phase set so that the com bined en ergy is de liv ered to

port 3. 401

Fig ure 7.34 Ad di tional switch ing and com bin ing func tions en abled by add ing a sec -

ond hy brid and an other phase shifter to a hot switch ing combiner. 402

Fig ure 7.35 The di elec tric vane switcher, which con sists of a long di elec tric sheet

mounted within a sec tion of rect an gu lar wave guide. 404

Fig ure 7.36 Di elec tric post wave guide phase shifter. 405

Fig ure 7.37 Vari able-phase hy brid phase shifter. 406

Fig ure 7.38 Op ti cal an tenna feed sys tems: (a) lens, (b) re flec tor. 408
Fig ure 7.39 Se ries-feed net works: (a) end feed, (b) cen ter feed, (c) sep a rate op ti mi za -

tion, (

d) equal path length, (e) se ries phase shift ers. 409

Fig ure 7.40 Types of par al lel-feed net works: (a) matched cor po rate feed, (b) re ac tive

cor po rate feed, (

c) re ac tive stripline, (d) mul ti ple re ac tive di vider. 410

Fig ure 7.41 The But ler beam-forming net work. 410

Fig ure 7.42 Ba sic con cept of a Reggia-Spencer phase shifter. 411

Fig ure 7.43 Switched-line phase shifter us ing pin di odes. 412
Fig ure 7.44 Typ i cal con struc tion of a TR tube. 412
Fig ure 7.45 Bal anced du plex er cir cuit us ing dual TR tubes and two short-slot hy brid

junc tions: (

a) trans mit mode, (b) re ceive mode. 413

Fig ure 7.46 Con struc tion of a dissipative wave guide fil ter. 414
Fig ure 7.47 Ba sic char ac ter is tics of a circulator: (a) op er a tional sche matic, (b) dis trib -

uted con stant circulator, (

c) lump con stant circulator. 415

Fig ure 7.48 Out put of a kly stron op er at ing into dif fer ent loads through a high-power

iso la tor: (

a) re sis tive load, (b) an an tenna sys tem. 417

Fig ure 7.49 Use of a circulator as a di plex er in TV ap pli ca tions. 418

Fig ure 7.50 Using mul ti ple circulators to form a multi plexer. 418

Cooling Con sid er ations

Fig ure 8.1 Blackbody en ergy dis tri bu tion char ac ter is tics. 425

Fig ure 8.2 Nukiyama heat-transfer curves: (a) log a rith mic, (b) lin ear. 426
Fig ure 8.3 Cooling sys tem de sign for a power grid tube. 429

Fig ure 8.4 The use of a chim ney to im prove cool ing of a power grid tube. 429

Fig ure 8.5 A poorly de signed cool ing sys tem in which cir cu lat ing air in the out put

com part ment re duces the ef fec tive ness of the heat-removal sys tem. 430

Fig ure 8.6 A ma nom e ter, used to mea sure air pres sure. 430

Fig ure 8.7 Cooling air flow re quire ments for a power grid tube. 431
Fig ure 8.8 Com bined cor rec tion fac tors for land-based tube ap pli ca tions. 434
Fig ure 8.9 Con ver sion of mass air flow rate to vol u met ric air flow rate. 435
Fig ure 8.10 Func tional sche matic of a wa ter cool ing sys tem for a high-power am pli -

fier or os cil la tor. 436

Fig ure 8.11 Typ i cal con fig u ra tion of a wa ter pu ri fi ca tion loop. 437
Fig ure 8.12 The ef fect of tem per a ture on the re sis tiv ity of ultrapure wa ter. 438

Fig ure 8.13 Va por-phase-cooled tubes re moved from their com pan ion boil ers. 441

Fig ure 8.14 Va por-phase cool ing sys tem for a power tube. 442
Fig ure 8.15 Cross sec tion of a holed an ode de sign for va por-phase cool ing. 444

Fig ure 8.16 A 4CV35,000A tetrode mounted in its boiler as sem bly. 444

Fig ure 8.17 Con trol box for a va por-phase cool ing sys tem. 445

Fig ure 8.18 Cutaway view of a boiler and tube com bi na tion. 446

Fig ure 8.19 Cutaway view of a con trol box. 447

Fig ure 8.20 Typ i cal four-tube va por cool ing sys tem with a com mon wa ter sup ply. 448
Fig ure 8.21 Cutaway view of a pres sure equal izer fit ting. 449

Fig ure 8.22 Typ i cal four-tube va por cool ing sys tem us ing "steam-out-the-bottom"

boil ers. 450

Fig ure 8.23 Va por cool ing sys tem in cor po rat ing a res er voir of dis tilled wa ter. 451
Fig ure 8.24 Cutaway view of a "steam-out-the-bottom" boiler. 452

Fig ure 8.25 Air flow sys tem for an air-cooled power tube. 457

Fig ure 8.26 A ma nom e ter de vice used for mea sur ing back pres sure in the PA com part -

ment of an RF gen er a tor. 458

Fig ure 8.27 A typ i cal heat ing and cool ing ar range ment for a high-power trans mit ter

in stal la tion. Ducting of PA ex haust air should be ar ranged so that it of fers min i mum
re sis tance to air flow. 459

Fig ure 8.28 Case study in which ex ces sive sum mer time heat ing was elim i nated

through the ad di tion of a 1 hp ex haust blower to the build ing. 460

Fig ure 8.29 Case study in which ex ces sive back pres sure to the PA cav ity was ex pe ri -

enced dur ing win ter pe ri ods, when the roof top damper was closed. The prob lem was
elim i nated by re po si tion ing the damper as shown. 461

Fig ure 8.30 Case study in which air tur bu lence at the ex haust duct re sulted in re duced

air flow through the PA com part ment. The prob lem was elim i nated by add ing a 4-ft
ex ten sion to the out put duct. 462

Fig ure 8.31 Closed site ven ti la tion de sign, with a backup in let/out let sys tem. 463

Fig ure 8.32 Open site ven ti la tion de sign, us ing no air con di tioner. 465
Fig ure 8.33 Trans mit ter ex haust-collection hood. 466

Fig ure 8.34 Hy brid site ven ti la tion de sign, us ing an air con di tioner in ad di tion to fil -

tered out side air. 467

Fig ure 8.35 Re sults of the "foam ing test" for wa ter pu rity: (a) pure sam ple, (b) sam ple

that is ac cept able on a short-term ba sis, (

c) un ac cept able sam ple. 470

Re li abil ity Con sid er ations

Fig ure 9.1 Ex am ple fault tree anal y sis di a gram: (a) pro cess steps, (b) fault tree sym -

bols, (

c) ex am ple di a gram. 479

Fig ure 9.2 The sta tis ti cal dis tri bu tion of equip ment or com po nent fail ures vs. time for

elec tronic sys tems and de vices. 481

Fig ure 9.3 Dis tri bu tion of com po nent fail ures iden ti fied through burn-in test ing: (a)

steady-state high-temperature burn-in, (

b) tem per a ture cy cling. 482

Fig ure 9.4 The fail ure his tory of a piece of avi on ics equip ment vs. time. Note that 60

per cent of the fail ures oc curred within the first 200 hours of ser vice. 483

Fig ure 9.5 The roller-coaster haz ard rate curve for elec tronic sys tems. 484

Fig ure 9.6 The ef fects of en vi ron men tal con di tions on the roller-coaster haz ard rate

curve. 484

Fig ure 9.7 The ef fects of en vi ron men tal stress screen ing on the re li abil ity bath tub

curve. 485

Fig ure 9.8 Venn di a gram rep re sen ta tion of the re la tion ship be tween flaw pre cip i ta tion

and ap plied en vi ron men tal stress. 487

Fig ure 9.9 Es ti ma tion of the prob a ble time to fail ure from an ab nor mal sol der joint.

488

Fig ure 9.10 Proper fingerstock scor ing of fil a ment and con trol grid con tacts on a

power tube. 493

Fig ure 9.11 Dam age to the grid con tact of a power tube caused by in suf fi cient

fingerstock pres sure. 493

Fig ure 9.12 The ef fects of fil a ment volt age man age ment on a thoriated-tungsten fil a -

ment power tube: (

a) use ful life as a func tion of fil a ment volt age, (b) avail able

power as a func tion of fil a ment volt age. Note the dra matic in crease in emis sion
hours when fil a ment volt age man age ment is prac ticed. 500

Fig ure 9.13 The ef fect of kly stron emit ter tem per a ture on beam cur rent. 501

Fig ure 9.14 Arc pro tec tion cir cuit uti liz ing an en ergy-diverting el e ment. 503
Fig ure 9.15 A new, un used 4CX15,000A tube. Con trast the ap pear ance of this de vice

with the tubes shown in the fol low ing fig ures. 506

Fig ure 9.16 An ode dis si pa tion pat terns on two 4CX15,000A tubes: (a) ex ces sive heat -

ing, (

b) nor mal wear. 507

Fig ure 9.17 Base heat ing pat terns on two 4CX15,000A tubes: (a) ex ces sive heat ing,

(

b) nor mal wear. 507

Fig ure 9.18 A 4CX5000A tube that ap pears to have suf fered sock et ing prob lems. 508
Fig ure 9.19 The in te rior el e ments of a 4CX15,000A tube that had gone to air while the

fil a ment was lit. 508

Fig ure 9.20 A 4CX15,000A tube show ing signs of ex ter nal arc ing. 509

Fig ure 9.21 Elec tri cal con nec tions for re con di tion ing the gun el e ments of a kly stron.

515

De vice Per for mance Cri te ria

Fig ure 10.1 Root-mean-square (rms) volt age mea sure ments: (a) the re la tion ship of

rms and av er age val ues, (

b) the rms mea sure ment cir cuit. 525

Fig ure 10.2 Av er age volt age mea sure ments: (a) il lus tra tion of av er age de tec tion, (b)

av er age mea sure ment cir cuit. 526

Fig ure 10.3 Com par i son of rms and av er age volt age char ac ter is tics. 527

Fig ure 10.4 Peak volt age mea sure ments: (a) il lus tra tion of peak de tec tion, (b) peak

mea sure ment cir cuit. 528

Fig ure 10.5 Il lus tra tion of the crest fac tor in volt age mea sure ments. 529

Fig ure 10.6 The ef fects of in stru ment band width on volt age mea sure ments. 530

Fig ure 10.7 Ex am ple of the equiv a lence of volt age and power deci bels. 531
Fig ure 10.8 Il lus tra tion of the mea sure ment of a phase shift be tween two sig nals. 534

Fig ure 10.9 Il lus tra tion of to tal har monic dis tor tion (THD) mea sure ment of an am pli -

fier trans fer char ac ter is tic. 535

Fig ure 10.10 Ex am ple of read ing THD from a spec trum an a lyzer. 536
Fig ure 10.11 Com mon test setup to mea sure har monic dis tor tion with a spec trum an a -

lyzer. 537

Fig ure 10.12 Sim plified block di a gram of a har monic dis tor tion an a lyzer. 537

Fig ure 10.13 Typ i cal test setup for mea sur ing the har monic and spu ri ous out put of a

trans mit ter. The notch fil ter is used to re move the fun da men tal fre quency to pre vent
overdriving the spec trum an a lyzer in put and to aid in eval u a tion of dis tor tion com -
po nents. 538

Fig ure 10.14 Con ver sion graph for in di cated dis tor tion and true dis tor tion. 538

Fig ure 10.15 Ex am ple of in ter fer ence sources in dis tor tion and noise mea sure ments.

539

Fig ure 10.16 Trans fer-function mon i tor ing con fig u ra tion us ing an os cil lo scope and

dis tor tion an a lyzer. 539

Fig ure 10.17 Il lus tra tion of prob lems that oc cur when mea sur ing har monic dis tor tion

in band-limited sys tems. 540

Fig ure 10.18 Fre quency-domain re la tion ships of the de sired sig nals and low-order

IMD prod ucts. 541

Fig ure 10.19 Test setup for mea sur ing the intermodulation dis tor tion of an RF am pli -

fier us ing a spec trum an a lyzer. 542

Fig ure 10.20 Typ i cal two-tone IMD mea sure ment, which eval u ates the third-order

prod ucts within the bandpass of the orig i nal tones. 543

Fig ure 10.21 Il lus tra tion of the ad di tion of dis tor tion com po nents: (a) ad di tion of

trans fer-function nonlinearities, (

b) ad di tion of dis tor tion com po nents. 543

Fig ure 10.22 Il lus tra tion of the can cel la tion of dis tor tion com po nents: (a) can cel la tion

of trans fer-characteristic nonlinearities, (

b) can cel la tion of the dis tor tion wave form.

544

Fig ure 10.23 Ba sic model of an intermodulation cor rec tion cir cuit. 545

Fig ure 10.24 Dig i tal de cod ing pro cess: (a) 16-QAM, (b) 8-VSB. 546
Fig ure 10.25 Digitally mod u lated RF sig nal spec trum be fore and af ter am pli fi ca tion.

547

Fig ure 10.26 RF out put power vari a tion as a func tion of dc beam in put power for a kly -

stron. 554

Fig ure 10.27 RF out put power of a kly stron as a func tion of RF drive power. 555

Fig ure 10.28 Out put power vari a tion of a kly stron as a func tion of drive power un der

dif fer ent tun ing con di tions. 556

Fig ure 10.29 Ideal grid-plate trans fer curve for class AB op er a tion. 557

Fig ure 10.30 Fre quency spec trum of third-order IM with the in ter fer ing level equal to

the car rier level. 558

Fig ure 10.31 Il lus tra tion of how the VSWR on a trans mis sion line af fects the at ten u a -

tion per for mance of the line. Scale

A rep re sents the VSWR at the trans mit ter; scale B

rep re sents the cor re spond ing re flected power in watts for an ex am ple sys tem. 561

Fig ure 10.32 Il lus tra tion of how VSWR causes ad di tional loss in a length of feedline.

562

Fig ure 10.33 The gen er a tion of syn chro nous AM in a band width-limited FM sys tem.

Note that min i mum syn chro nous AM oc curs when the sys tem op er ates in the cen ter
of its passband. 564

Fig ure 10.34 IPA plate-to-PA-grid match ing net work de signed for wide band width

op er a tion. 565

Fig ure 10.35 Syn chro nous AM con tent of an ex am ple FM sys tem as a func tion of sys -

tem band width. 566

Fig ure 10.36 To tal har monic dis tor tion (THD) con tent of an ex am ple de mod u lated

FM sig nal as a func tion of trans mis sion-channel band width. 567

Fig ure 10.37 The spec trum of an ideal AM trans mit ter. THD mea sures 0 per cent; there

are no har monic side bands. 578

Fig ure 10.38 The spec trum of an ideal trans mit ter with the ad di tion of IPM. 569

Fig ure 10.39 Car rier am pli tude reg u la tion (car rier level shift) de fined. 570
Fig ure 10.40 The me chan ics of intermod gen er a tion at a two-transmitter in stal la tion.

571

Safe Han dling of Vac uum Tube De vices

Fig ure 11.1 Elec tro car dio gram of a hu man heart beat: (a) healthy rhythm, (b) ven tric u -

lar fi bril la tion. 577

Fig ure 11.2 Ef fects of elec tric cur rent and time on the hu man body. Note the "let-go"

range. 578

Fig ure 11.3 Ba sic de sign of a ground-fault in ter rupter (GFI). 579

Fig ure 11.4 Ex am ple of how high volt ages can be gen er ated in an RF load match ing

unit. 580

Fig ure 11.5 Ba sic first aid treat ment for elec tric shock. 581
Fig ure 11.6 AC ser vice en trance bond ing re quire ments: (a) 120 V phase-to-neutral

(240 V phase-to-phase), (

b) 3-phase 208 V wye (120 V phase-to-neutral), (c)

3-phase 240 V (or 480 V) delta. Note that the main bond ing jumper is re quired in
only two of the de signs. 585

Fig ure 11.7 The power den sity lim its for nonionizing ra di a tion ex po sure for hu mans.

588

Fig ure 11.8 ANSI/IEEE ex po sure guide lines for mi cro wave fre quen cies. 590

Whitaker, Jerry C. "

Index of Tables"

Power Vacuum Tubes Handbook 2

nd

Edition.

Ed. Jerry C. Whitaker
Boca Raton: CRC Press LLC, 2000

Index of Ta bles

Ta ble 1.1 Chan nel Des ig na tions for VHF- and UHF-TV Sta tions in the United States

17

Ta ble 1.2 Typ i cal Ra dar Ap pli ca tions 22

Ta ble 1.3 Stan dard Ra dar Fre quency Bands 22

Ta ble 1.4 Ra dio Fre quencies Used for Elec tronic Nav i ga tion 25
Ta ble 1.5 Com mon-Carrier Mi cro wave Fre quencies Used in the United States 29

Ta ble 1.6 Fre quency Band Des ig na tions 32

Ta ble 2.1 Typ i cal Equiv a lent Cir cuit Pa ram e ter Values for Quartz Crys tals 47

Ta ble 2.2 Com par i son of Am pli tude Mod u la tion Tech niques 64
Ta ble 3.1 Min i mum Cooling Air flow Re quire ments for the 4CX15000A Power

Tetrode at Sea Level 102

Ta ble 3.2 Gen eral Char ac ter is tics of the 4CX15000A Power Tetrode 104
Ta ble 3.3 Gen eral Char ac ter is tics of the 5CX1500B Power Pentode 106
Ta ble 3.4 Min i mum Cooling Air flow Re quire ments for the 5CX1500B Power Tetrode

107

Ta ble 3.5 Char ac ter is tics of Com mon Thermonic Emit ters 117
Ta ble 4.1 Pro tec tion Guide lines for Tetrode and Pentode De vices 170

Ta ble 4.2 Pro tec tion Guide lines for a Tri ode 171

Ta ble 4.3 Op er ating Pa ram e ters of 3-Phase Rec ti fier Con fig u ra tions 212
Ta ble 5.1 Ap prox i mate In put Ca pac i tance of Com mon Power Grid Tubes Used in

Grounded-Grid and Grounded-Cathode Ser vice 253

Ta ble 6.1 Typ i cal Op er ating Pa ram e ters for an In te gral-Cavity Kly stron for UHF-TV

Ser vice 295

Ta ble 6.2 Typ i cal Op er ating Pa ram e ters for an Ex ter nal-Cavity MSDC Kly stron for

UHF-TV ser vice 310

Ta ble 6.3 Typ i cal Op er ating Pa ram e ters of the IOT in TV Ser vice 318

Ta ble 7.1 Rep re sen ta tive Spec i fi ca tions for Var i ous Types of Flex i ble Air-Dielectric

Co ax ial Ca ble 363

Ta ble 7.2 Elec tri cal Char ac ter is tics of Com mon RF Con nec tors 366

Ta ble 7.3 Sum mary of Stan dard Fil ter Align ments 387
Ta ble 7.4 Sin gle 90° Hy brid Sys tem Op er ating Modes 390

Ta ble 7.5 Op er ating Modes of the Dual 90° Hy brid/Sin gle-Phase Shifter Combiner

Sys tem 403

Ta ble 7.6 Typ i cal Per for mance of a Di elec tric Vane Phase Shifter 405
Ta ble 7.7 Typ i cal Per for mance of a Vari able-Phase Hy brid Phase Shifter 407

Ta ble 8.1 Ther mal Con duc tiv ity of Com mon Ma te rials 423

Ta ble 8.2 Rel a tive Ther mal Con duc tiv ity of Var i ous Ma te rials As a Per cent age of the

Ther mal Con duc tiv ity of Cop per 423

Ta ble 8.3 Emissivity Char ac ter is tics of Com mon Ma te rials at 80°F 425

Ta ble 8.4 Vari a tion of Elec tri cal and Ther mal Prop erties of Com mon In su la tors As a

Func tion of Tem per a ture 428

Ta ble 8.5 Cor rec tion Fac tors for Q and

P 433

Ta ble 9.1 Com par i son of Con ven tional Re li abil ity Testing and En vi ron men tal Stress

Screening 486

Ta ble 10.1 Com mon Deci bel Values and Con ver sion Ra tios 532

Ta ble 10.2 Three-Halves Cur rent Values of Com monly Used Fac tors 550

Ta ble 11.1 The Ef fects of Cur rent on the Hu man Body 576
Ta ble 11.2 Re quired Safety Prac tices for En gi neers Working Around High-Voltage

Equip ment 579

Ta ble 11.3 Ba sic First Aid Pro ce dures for Burns (More de tailed in for ma tion can be ob -

tained from any Red Cross of fice.) 583

Ta ble 11.4 Six teen Com mon OSHA Vi o la tions 584

Ta ble 11.5 Com monly Used Names for PCB In su lating Ma te rial 592

Ta ble 11.6 Def i ni tion of PCB Terms as Iden tified by the EPA 594

Whitaker, Jerry C. "

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Document Outline

- Power Vacuum Tubes Handbook Second Edition

- Disclaimer
- Preface
- About the Author
- Acknowledgement
- Contents
- 1. Power Vacuum Tube Applications

- 1.1 Introduction

- 1.1.1 Vacuum Tube Development
- 1.1.2 Standardization
- 1.1.3 Transmission Systems

- 1.2 Vacuum Tube Applications

- 1.2.1 Market Overview
- 1.2.2 AM Radio Broadcasting
- 1.2.3 Shortwave Broadcasting
- 1.2.4 FM Radio Broadcasting
- 1.2.5 TV Broadcasting
- 1.2.6 Satellite Transmission
- 1.2.7 Radar
- 1.2.8 Electronic Navigation
- 1.2.9 Microwave Radio
- 1.2.10 Induction Heating
- 1.2.11 Electromagnetic Radiation Spectrum

- 1.3 Bibliography

- 2. Modulation Systems and Characteristics

- 2.1 Introduction

- 2.1.1 Modulation Systems
- 2.1.2 Principles of Resonance
- 2.1.3 Frequency Source
- 2.1.4 Operating Class
- 2.1.5 Broadband Amplifier Design
- 2.1.6 Thermal and Circuit Noise

- 2.2 Amplitude Modulation

- 2.2.1 High-Level AM Modulation
- 2.2.2 Vestigial-Sideband Amplitude Modulation
- 2.2.3 Single-Sideband Amplitude Modulation
- 2.2.4 Quadrature Amplitude Modulation (QAM)

- 2.3 Frequency Modulation

- 2.3.1 Modulation Index
- 2.3.2 Phase Modulation
- 2.3.3 Modifying FM Waves
- 2.3.4 Preemphasis and Deemphasis
- 2.3.5 Modulation Circuits

- 2.4 Pulse Modulation

- 2.4.1 Digital Modulation Systems
- 2.4.2 Pulse Amplitude Modulation
- 2.4.3 Pulse Time Modulation (PTM)
- 2.4.4 Pulse Code Modulation
- 2.4.5 Delta Modulation
- 2.4.6 Digital Coding Systems
- 2.4.7 Baseband Digital Pulse Modulation
- 2.4.8 Spread Spectrum Systems

- 2.5 References
- 2.6 Bibliography

- 3. Vacuum Tube Principles

- 3.1 Introduction
- 3.2 Characteristics of Electrons

- 3.2.1 Electron Optics
- 3.2.2 Thermal Emission From Metals
- 3.2.3 Secondary Emission
- 3.2.4 Diode
- 3.2.5 Triode
- 3.2.6 Tetrode
- 3.2.7 Pentode
- 3.2.8 High-Frequency Operating Limits

- 3.3 Vacuum Tube Design

- 3.3.1 Device Cooling
- 3.3.2 Cathode Assembly
- 3.3.3 Grid Structures
- 3.3.4 Plate Assembly
- 3.3.5 Ceramic Elements
- 3.3.6 Tube Construction
- 3.3.7 Connection Points
- 3.3.8 Tube Sockets

- 3.4 Neutralization

- 3.4.1 Circuit Analysis
- 3.4.2 Circuit Design
- 3.4.3 Grounded-Grid Amplifier Neutralization
- 3.4.4 Self-Neutralizing Frequency
- 3.4.5 Neutralization Adjustment

- 3.5 References
- 3.6 Bibliography

- 4. Designing Vacuum Tube Circuits

- 4.1 Introduction

- 4.1.1 Class A Amplifier
- 4.1.2 Class B and AB Amplifiers
- 4.1.3 Class C Amplifier

- 4.2 Principles of RF Power Amplification

- 4.2.1 Drive Power Requirements
- 4.2.2 Mechanical and Electrical Considerations
- 4.2.3 Bypassing Tube Elements
- 4.2.4 Parasitic Oscillations
- 4.2.5 Shielding
- 4.2.6 Protection Measures

- 4.3 Cavity Amplifier Systems

- 4.3.1 Bandwidth and Efficiency
- 4.3.2 Current Paths
- 4.3.3 The 1/4-Wavelength Cavity
- 4.3.4 The 1/2-Wavelength Cavity
- 4.3.5 Folded 1/2-Wavelength Cavity
- 4.3.6 Wideband Cavity
- 4.3.7 Output Coupling
- 4.3.8 Mechanical Design

- 4.4 High-Voltage Power Supplies

- 4.4.1 Silicon Rectifiers
- 4.4.2 Operating Rectifiers in Series
- 4.4.3 Operating Rectifiers in Parallel
- 4.4.4 Silicon Avalanche Rectifiers
- 4.4.5 Thyristor Servo Systems
- 4.4.6 Polyphase Rectifier Circuits
- 4.4.7 Power Supply Filter Circuits

- 4.5 Parameter Sampling Circuits
- 4.6 References
- 4.7 Bibliography

- 5. Applying Vacuum Tube Devices

- 5.1 Introduction
- 5.2 AM Power Amplification Systems

- 5.2.1 Control Grid Modulation
- 5.2.2 Suppressor Grid Modulation
- 5.2.3 Cathode Modulation
- 5.2.4 High-Level AM Amplification
- 5.2.5 Pulse Width Modulation

- 5.3 Linear Amplification

- 5.3.1 Device Selection
- 5.3.2 Grid-Driven Linear Amplifier
- 5.3.3 Cathode-Driven Linear Amplifier
- 5.3.4 Intermodulation Distortion

- 5.4 High-Efficiency Linear Amplification

- 5.4.1 Chireix Outphasing Modulated Amplifier
- 5.4.2 Doherty Amplifier
- 5.4.3 Screen-Modulated Doherty-Type Amplifier
- 5.4.4 Terman-Woodyard Modulated Amplifier
- 5.4.5 Dome Modulated Amplifier

- 5.5 Television Power Amplifier Systems

- 5.5.1 System Considerations
- 5.5.2 Power Amplifier

- 5.6 FM Power Amplifier Systems

- 5.6.1 Cathode-Driven Triode Amplifier
- 5.6.2 Grounded-Grid vs. Grid-Driven Tetrode
- 5.6.3 Grid-Driven Tetrode/Pentode Amplifiers
- 5.6.4 Impedance Matching into the Grid
- 5.6.5 Neutralization

- 5.7 Special-Application Amplifiers

- 5.7.1 Distributed Amplification
- 5.7.2 Radar

- 5.8 References
- 5.9 Bibliography

- 6. Microwave Power Tubes

- 6.1 Introduction

- 6.1.1 Linear-Beam Tubes
- 6.1.2 Crossed-Field Tubes

- 6.2 Grid Vacuum Tubes

- 6.2.1 Planar Triode
- 6.2.2 High-Power UHF Tetrode
- 6.2.3 Diacrode

- 6.3 Klystron

- 6.3.1 Reflex Klystron
- 6.3.2 The Two-Cavity Klystron
- 6.3.3 The Multicavity Klystron
- 6.3.4 Beam Pulsing
- 6.3.5 Integral vs. External Cavity
- 6.3.6 MSDC Klystron

- 6.4 Klystrode/Inductive Output Tube (IOT)

- 6.4.1 Theory of Operation
- 6.4.2 Electron Gun
- 6.4.3 Grid Structure
- 6.4.4 Input Cavity
- 6.4.5 Output Cavity
- 6.4.6 Application Considerations
- 6.4.7 Continuing Research Efforts

- 6.5 Constant Efficiency Amplifier

- 6.5.1 Theory of Operation

- 6.6 Traveling Wave Tube

- 6.6.1 Theory of Operation
- 6.6.2 Operating Efficiency
- 6.6.3 Operational Considerations

- 6.7 Crossed-Field Tubes

- 6.7.1 Magnetron
- 6.7.2 Backward Wave Oscillator
- 6.7.3 Strap-Fed Devices
- 6.7.4 Gyrotron

- 6.8 Other Microwave Devices

- 6.8.1 Quasiquantum Devices
- 6.8.2 Variations on the Klystron

- 6.9 Microwave Tube Life

- 6.9.1 Life-Support System
- 6.9.2 Protection Measures
- 6.9.3 Filament Voltage Control
- 6.9.4 Cooling System
- 6.9.5 Reliability Statistics

- 6.10 References
- 6.11 Bibliography

- 7. RF Interconnection and Switching

- 7.1 Introduction

- 7.1.1 Skin Effect

- 7.2 Coaxial Transmission Line

- 7.2.1 Electrical Parameters
- 7.2.2 Electrical Considerations
- 7.2.3 Coaxial Cable Ratings
- 7.2.4 Mechanical Parameters

- 7.3 Waveguide

- 7.3.1 Propagation Modes
- 7.3.2 Ridged Waveguide
- 7.3.3 Circular Waveguide
- 7.3.4 Doubly Truncated Waveguide
- 7.3.5 Impedance Matching
- 7.3.6 Installation Considerations
- 7.3.7 Cavity Resonators

- 7.4 RF Combiner and Diplexer Systems

- 7.4.1 Passive Filters A filter is a multiport-network designed specifically to respond
- 7.4.2 Four-Port Hybrid Combiner
- 7.4.3 Non-Constant-Impedance Diplexer
- 7.4.4 Constant-Impedance Diplexer
- 7.4.5 Microwave Combiners
- 7.4.6 Hot Switching Combiners
- 7.4.7 Phased-Array Antenna Systems

- 7.5 High-Power Isolators

- 7.5.1 Theory of Operation
- 7.5.2 Applications

- 7.6 References
- 7.7 Bibliography

- 8. Cooling Considerations

- 8.1 Introduction

- 8.1.1 Thermal Properties
- 8.1.2 Heal Transfer Mechanisms
- 8.1.3 The Physics of Boiling Water

- 8.2 Application of Cooling Principles

- 8.2.1 Forced-Air Cooling Systems
- 8.2.2 Water Cooling
- 8.2.3 Vapor-Phase Cooling
- 8.2.4 Temperature Measurements
- 8.2.5 Air-Handling System

- 8.3 Operating Environment

- 8.3.1 Air-Handling System
- 8.3.2 Air Cooling System Design
- 8.3.3 Site Design Guidelines
- 8.3.4 Water/Vapor Cooling System Maintenance

- 8.4 References
- 8.5 Bibliography

- 9. Reliability Considerations

- 9.1 Introduction

- 9.1.1 Terminology

- 9.2 Quality Assurance

- 9.2.1 Inspection Process
- 9.2.2 Reliability Evaluation
- 9.2.3 Failure Analysis
- 9.2.4 Standardization

- 9.3 Reliability Analysis

- 9.3.1 Statistical Reliability
- 9.3.2 Environmental Stress Screening
- 9.3.3 Latent Defects
- 9.3.4 Operating Environment
- 9.3.5 Failure Modes
- 9.3.6 Maintenance Considerations

- 9.4 Vacuum Tube Reliability

- 9.4.1 Thermal Cycling
- 9.4.2 Tube-Changing Procedure
- 9.4.3 Power Tube Conditioning
- 9.4.4 Filament Voltage
- 9.4.5 Filament Voltage Management
- 9.4.6 PA Stage Tuning
- 9.4.7 Fault Protection
- 9.4.8 Vacuum Tube Life
- 9.4.9 Examining Tube Performance
- 9.4.10 Shipping and Handling Vacuum Tubes

- 9.5 Klystron Reliability

- 9.5.1 Cleaning and Flushing the Cooling System
- 9.5.2 Cleaning Ceramic Elements
- 9.5.3 Reconditioning Klystron Gun Elements
- 9.5.4 Focusing Electromagnet Maintenance
- 9.5.5 Power Control Considerations

- 9.6 References
- 9.7 Bibliography

- 10. Device Performance Criteria

- 10.1 Introduction
- 10.2 Measurement Parameters

- 10.2.1 Power Measurements
- 10.2.2 Decibel Measurement
- 10.2.3 Noise Measurement
- 10.2.4 Phase Measurement
- 10.2.5 Nonlinear Distortion

- 10.3 Vacuum Tube Operating Parameters

- 10.3.1 Stage Tuning
- 10.3.2 Amplifier Balance
- 10.3.3 Parallel Tube Amplifiers
- 10.3.4 Harmonic Energy
- 10.3.5 Klystron Tuning Considerations
- 10.3.6 Intermodulation Distortion
- 10.3.7 VSWR

- 10.4 RF System Performance

- 10.4.1 Key System Measurements
- 10.4.2 Synchronous AM in FM Systems 3
- 10.4.3 Incidental Phase Modulation
- 10.4.4 Carrier Amplitude Regulation
- 10.4.5 Site-Related Intermodulation Products

- 10.5 References
- 10.6 Bibliography

- 11. Safe Handling of Vacuum Tube Devices

- 11.1 Introduction
- 11.2 Electric Shock

- 11.2.1 Effects on the Human Body
- 11.2.2 Circuit Protection Hardware
- 11.2.3 Working with High Voltage
- 11.2.4 First Aid Procedures

- 11.3 Operating Hazards

- 11.3.1 OSHA Safety Considerations
- 11.3.2 Beryllium Oxide Ceramics
- 11.3.3 Corrosive and Poisonous Compounds
- 11.3.4 FC-75 Toxic Vapor
- 11.3.5 Nonionizing Radiation
- 11.3.6 X-Ray Radiation Hazard
- 11.3.7 Implosion Hazard
- 11.3.8 Hot Coolant and Surfaces
- 11.3.9 Polychlorinated Biphenyls

- 11.4 References
- 11.5 Bibliography

- Reference Data

- Standard Units
- Standard Prefixes
- Common Standard Units for Electronics Work
- Power Conversion Factors (Decibels to Watts)
- Voltage Standing Wave Ratio and Key Parameters
- Specifications of Standard Copper Wire
- Celsius-to-Fahrenheit Conversion
- Inch-to-Millimeter Conversion
- Millimeters-to-Decimal Inches Conversion
- Conversion of Common Fractions to Decimal and Millimeter Units
- Conversion Ratios for Length
- Conversion Ratios for Area
- Conversion Ratios for Mass
- Conversion Ratios for Volume
- Conversion Ratios for Cubic Measure
- Conversion Ratios for Electrical Quantities
- Basic Electrical Units

- Glossary

- A
- B
- C
- D
- E
- F
- G
- H
- I
- J
- K
- L
- M
- N
- O
- P
- Q
- R
- S
- T
- U
- V
- W
- X
- Y

- Index of Figures

- Power Vac uum Tube Ap pli ca tions
- Modulation Sys tems and Char ac ter is tics
- Vacuum Tube Prin ci ples
- De signing Vac uum Tube Cir cuits
- Ap plying Vac uum Tube De vices
- Microwave Power Tubes
- RF In terconnection and Switching
- Cooling Con sid er ations
- Re li abil ity Con siderations
- Device Per formance Cri te ria
- Safe Han dling of Vac uum Tube De vices

- Index of Tables
- Cited References

- (ebook) CRC Press - Power Vacuum Tube Handbook (Whitaker).pdf